AN-9729_技术文档

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AN-9729

LED Application Design Guide Using Half-Bridge LLC

Resonant Converter for 100W Street Lighting

Among various kinds of resonant converters, the simplest

and most popular is the LC series resonant converter, where

Introduction

This application note describes the LED driving system

using a half-bridge LLC resonant converter for high

power LED lighting applications, such as outdoor or

street lighting. Due to the existence of the non-isolation

DC-DC converter to control the LED current and the light

intensity, the conventional PWM DC-DC converter has

the problem of low-power conversion efficiency. The

half-bridge LLC converter can perform the LED current

control and the efficiency can be significantly improved.

Moreover, the cost and the volume of the whole LED

driving system can be reduced.

the rectifier-load network is placed in series with the L-C

resonant network, as depicted in Figure 1^[2-4].In this

configuration, the resonant network and the load act as a

voltage divider. By changing the frequency of driving

voltage V_d, the impedance of the resonant network changes.

The input voltage is split between this impedance and the

reflected load. Since it is a voltage divider, the DC gain of a

LC series resonant converter is always <1. At light-load

condition, the impedance of the load is large compared to

the impedance of the resonant network; all the input voltage

is imposed on the load. This makes it difficult to regulate

the output at light load. Theoretically, frequency should be

infinite to regulate the output at no load.

Consideration of LED Drive

LED lighting is rapidly replacing conventional lighting

sources like incandescent bulbs, fluorescent tubes, and

halogens because LED lighting reduces energy

consumption. LED lighting has greater longevity,

contains no toxic materials, and emits no harmful UV

rays, which are 5 ~ 20 times longer than fluorescent tubes

and incandescent bulbs. All metal halide and fluorescent

lamps, including CFLs, n contain mercury.

The amount of current through an LED determines the

light it emits. The LED characteristics determine the

forward voltage necessary to achieve the required level of

current. Due to the variation in LED voltage versus

current characteristics, controlling only the voltage across

the LED leads to variability in light output. Therefore,

most LED drivers use current regulation to support

brightness control. Brightness can be controlled directly

by changing the LED current.

Figure 1. Half-Bridge, LC Series Resonant Converter

To overcome the limitation of series resonant converters,

the LLC resonant converter has been proposed^[8-12]. The

LLC resonant converter is a modified LC series resonant

converter implemented by placing a shunt inductor across

the transformer primary winding, as depicted in Figure 2.

When this topology was first presented, it did not receive

much attention due to the counterintuitive concept that

increasing the circulating current in the primary side with

a shunt inductor can be beneficial to circuit operation.

However, it can be very effective in improving efficiency

for high-input voltage applications where the switching

loss is more dominant than the conduction loss.

Consideration of LLC Resonant

Converter

The attempt to obtain ever-increasing power density of

switched-mode power supplies has been limited by the

size of passive components. Operation at higher

frequencies considerably reduces the size of passive

components, such as transformers and filters; however,

switching losses have been an obstacle to high-frequency

operation. To reduce switching losses and allow high-

frequency operation, resonant switching techniques have

been developed. These techniques process power in a

sinusoidal manner and the switching devices are softly

commutated. Therefore, the switching losses and noise

In most practical designs, this shunt inductor is realized

using the magnetizing inductance of the transformer. The

circuit diagram of LLC resonant converter looks much the

same as the LC series resonant converter: the only

difference is the value of the magnetizing inductor. While

the series resonant converter has

a magnetizing

inductance larger than the LC series resonant inductor

(L_r), the magnetizing inductance in an LLC resonant

converter is just 3~8 times L_r, which is usually

implemented by introducing an air gap in the transformer.

can be dramatically reduced^[1-7]

.

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AN-9729

APPLICATION NOTE

network even though a square-wave voltage is

applied to the resonant network. The current (I_p) lags

the voltage applied to the resonant network (that is,

the fundamental component of the square-wave

voltage (V_d) applied to the half-bridge totem pole),

which allows the MOSFETs to be turned on with

zero voltage. As shown in Figure 4, the MOSFET

turns on while the voltage across the MOSFET is

zero by flowing current through the anti-parallel

diode.

The rectifier network produces DC voltage by

rectifying the AC current with rectifier diodes and a

capacitor. The rectifier network can be implemented

as a full-wave bridge or center-tapped configuration

with capacitive output filter.

Figure 2. Half-Bridge LLC Resonant Converter

.

An LLC resonant converter has many advantages over a

series resonant converter. It can regulate the output over

wide line and load variations with a relatively small

variation of switching frequency. It can achieve zero

voltage switching (ZVS) over the entire operating range.

All essential parasitic elements, including junction

capacitances of all semiconductor devices and the leakage

inductance and magnetizing inductance of the

transformer, are utilized to achieve soft switching.

Square-Wave Generator

Q₁

Rectifier

Network

Resonant

Network

I_DS1

This application note presents design considerations of an

LLC resonant half-bridge converter employing

Fairchild’s FLS-XS series. It includes explanation of the

LLC resonant converter operation principles, designing

the transformer and resonant network, and selecting the

components. The step-by-step design procedure,

explained with a design example, helps design the LLC

resonant converter.

I_p

I_o

+

n:1

I_D

L_r

I_m

L_m

+

V_d

-

V_IN

Q₂

R_o

V_O

-

C_r

Figure 3. Schematic of Half-Bridge LLC

Resonant Converter

LLC Resonant Converter and

Fundamental Approximation

I_p

I_m

Figure 3 shows a simplified schematic of a half-bridge

LLC resonant converter, where L_mis the magnetizing

inductance that acts as a shunt inductor, L_ris the series

resonant inductor, and C_ris the resonant capacitor.

Figure 4 illustrates the typical waveforms of the LLC

resonant converter. It is assumed that the operation

frequency is same as the resonance frequency, determined

by the resonance between L_rand C_r. Since the

magnetizing inductor is relatively small, a considerable

amount of magnetizing current (I_m) exists, which

freewheels in the primary side without being involved in

the power transfer. The primary-side current (I_p) is sum of

the magnetizing current and the secondary-side current

referred to the primary.

I_DS1

I_D

V_IN

V_d

V_gs1

V_gs2

In general, the LLC resonant topology consists of three

stages shown in Figure 3; square-wave generator,

resonant network, and rectifier network.

.

The square-wave generator produces a square-wave

voltage, V_d, by driving switches Q₁and Q₂alternately

with 50% duty cycle for each switch. A small dead

time is usually introduced between the consecutive

transitions. The square-wave generator stage can be

built as a full-bridge or half-bridge type.

The resonant network consists of a capacitor, leakage

inductances, and the magnetizing inductance of the

transformer. The resonant network filters the higher

harmonic currents. Essentially, only sinusoidal

current is allowed to flow through the resonant

Figure 4. Typical Waveforms of Half-Bridge LLC

Resonant Converter

The filtering action of the resonant network allows use of

the fundamental approximation to obtain the voltage gain

of the resonant converter, which assumes that only the

fundamental component of the square-wave voltage input

to the resonant network contributes to the power transfer

to the output. Because the rectifier circuit in the

secondary side acts as an impedance transformer, the

equivalent load resistance is different from actual load

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AN-9729

APPLICATION NOTE

resistance. Figure 5 shows how this equivalent load

resistance is derived. The primary-side circuit is replaced

by a sinusoidal current source, I_ac, and a square wave of

voltage, V_RI, appears at the input to the rectifier. Since the

average of |I_ac| is the output current, I_o, I_ac, is obtained as:

C_r

L_r

V_d

+

V_O

+

V_IN

V_RI

L_m

R_o

-

N_p:N_s

  I

2

I_ac



^osin(t)

(1)

8n²

 ²

R_ac



R_o

n=N_p/N_s

and V_RIis given as:

V_RI V_oif sin(t)  0

F

L_r

C_r

(2)

V_Ro

V_RI V_oif sin(t)  0

where V_ois the output voltage.

F

R_ac

V_d

F

(nV_RI

)

L_m

The fundamental component of V_RIis given as:

Figure 6. AC Equivalent Circuit for LLC

Resonant Converter

4V

F

V_RI



^osin(t)

(3)



With the equivalent load resistance obtained in Equation

5, the characteristics of the LLC resonant converter can

be derived. Using the AC equivalent circuit of Figure 6,

the voltage gain, M, is obtained as:

Since harmonic components of V_RIare not involved in the

power transfer, AC equivalent load resistance can be

calculated by dividing V_RI^Fby I_acas:

F

V_RI

I_ac

8 V_o

8

4n V



^osin(t)

R_ac



R_o

(4)

F

 ²I_o ²

V_RO

n V_RI

V_d

2n V_o

M 



F

4 V

V_d

V

ⁱⁿsin(t)

in

Considering the transformer turns ratio (n=N_p/N_s), the

equivalent load resistance shown in the primary side is

obtained as:

 2

)²(m 1)

(6)



_o

(

8n²

 ²



²



²

(5)

R_ac



R_o

(

1)  j

(

1)(m 1)Q

2

_p

_o_o

By using the equivalent load resistance, the AC

equivalent circuit is obtained, as illustrated in Figure 6,

where V_dand V_ROare the fundamental components of

the driving voltage, V_d, and reflected output voltage,

where:

L_p L_m L_r, R_ac

F

8n²

 ²

L_p

R_o, m 

L_r



V

_RO(nV_RI), respectively.

L_r

1

Q 

, _o

, _p

C_rR_ac

L_rC_r

L_pC_r

pk

I_ac

As can be seen in Equation (6), there are two resonant

frequencies. One is determined by L_rand C_r, while the

other is determined by L_pand C_r.

Equation (6) shows the gain is unity at resonant frequency

(ω_o), regardless of the load variation, which is given as:

2

(m 1)_p

2nV_o

M 



1 at  _o

(7)

2

V

_o²_p

in

The gain of Equation (6) is plotted in Figure 7 for

different Q values with m=3, f_o=100kHz, and f_p=57kHz.

As observed in Figure 7, the LLC resonant converter

shows gain characteristics that are almost independent of

the load when the switching frequency is around the

resonant frequency, f_o. This is a distinct advantage of

LLC-type resonant converter over the conventional series

resonant converter. Therefore, it is natural to operate the

converter around the resonant frequency to minimize the

switching frequency variation.

  I

2

I_ac



^osin(wt)

4V

F

V_RI





Figure 5. Derivation of Equivalent Load Resistance R_ac

The operating range of the LLC resonant converter is

limited by the peak gain (attainable maximum gain),

which is indicated with ‘*’ in Figure 7. Note that the peak

voltage gain does not occur at f_oor f_p. The peak gain

frequency where the peak gain is obtained exists between

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APPLICATION NOTE

f_pand f_o, as shown in Figure 7. As Q decreases (as load

decreases), the peak gain frequency moves to f_pand

higher peak gain is obtained. Meanwhile, as Q increases

(as load increases), the peak gain frequency moves to f_o

and the peak gain drops; the full load condition should be

worst case for the resonant network design.

In Figure 8, the effective series inductor (L_p) and shunt

inductor (L_p-L_r) are obtained by assuming n²L_lks=L_lkpand

referring the secondary-side leakage inductance to the

primary side as:

L_p L_m L

lkp

(8)

L_r L  L_m//(n²L ) L  L_m// L

1

f_o



lkp

lks

lkp

f_p



2 L_rC_r

2 L_pC_r

When handling an actual transformer, equivalent circuit

with L_pand L_ris preferred since these values can be

measured with a given transformer. In an actual

L_r/ C_r

Q 

R_ac

transformer, L_pand L_rcan be measured in the primary

side with the secondary-side winding open circuited and

short circuited, respectively.

In Figure 9, notice that a virtual gain M_Vis introduced,

which is caused by the secondary-side leakage

inductance. By adjusting the gain equation of Equation

(6) using the modified equivalent circuit of Figure 9, the

gain equation for integrated transformer is obtained by:

M_{@ f}1

o



_o

(

)²(m 1)  M_V

2n V_O

M 



²

_p



²

V

1)(m 1)Q^e

1) (m 1)Q^e

in

(

1)  j( ) (

2

_o_o

Figure 7. Typical Gain Curves of LLC Resonant

²

_o



Converter (m=3)

(

₂) m(m 1)



²

_p

²

Consideration for Integrated

Transformer

(

1)  j( )(

2

_o_o

where:

(9)

For practical design, it is common to implement the

magnetic components (series inductor and shunt inductor)

using an integrated transformer; where the leakage

inductance is used as a series inductor, while the

magnetizing inductor is used as a shunt inductor. When

building the magnetizing components in this way, the

equivalent circuit in Figure 6 should be modified as

shown in Figure 8 because leakage inductance exists, not

only in the primary side, but also in the secondary side.

Not considering the leakage inductance in the transformer

secondary side generally results in an ineffective design.

8n²

R_o

 ²M_V

L_p

L_r

e

R_ac



₂, m 

, _o

L_r

1

Q^e

, _p

e

C_rR_ac

L_rC_r

L_pC_r

The gain at the resonant frequency (ω_o) is fixed

regardless of the load variation, which is given as:

L_p

m

M  M_V



at  _o

(10)

L_p L_r

m 1

The gain at the resonant frequency (ω_o) is unity when

using individual core for series inductor, as shown in

Equation 7. However, when implementing the magnetic

components with integrated transformer, the gain at the

resonant frequency (ω_o) is larger than unity due to the

virtual gain caused by the leakage inductance in the

transformer secondary side.

L_r L_lkp L_m//(n²L

)

lks

 L_lkp L_m// L

lkp

L_p

(M_V



)

L_p L_lkp L_m

L_p L_r

The gain of Equation (9) is plotted in Figure 10 for

different Q^evalues with m=3, f_o=100kHz, and f_p=57kHz.

As observed in Figure 9, the LLC resonant converter shows

gain characteristics almost independent of the load when

the switching frequency is around the resonant frequency,

f_o.

1: M_V

R_ac

Figure 8. Modified Equivalent Circuit to

Accommodate the Secondary-Side Leakage

Inductance

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AN-9729

APPLICATION NOTE

1

f_o



f_p



2 L_rC_r

Gain (M)

B

2 L_pC_r

A

L_r/ C_r

Q^e



e

R_ac

Load Increase

I

II

Above

Resonance

(f_s>f_o)

Below

Resonance

(f_s<f_o)

M_{@ f} M_V

o

f_o

f_s

Figure 10. Operation Modes According to the

Operation Frequency

1

Figure 9. Typical Gain Curves of LLC Resonant

2 f_o

Converter (m=3) Using an Integrated Transformer

1

2 f_S

Consideration of Operation Mode

and Attainable Maximum Gain

Operation Mode

The LLC resonant converter can operate at frequency

below or above the resonance frequency (f_o), as illustrated

in Figure 10. Figure 11 shows the waveforms of the

currents in the transformer primary side and secondary

side for each operation mode. Operation below the

resonant frequency (case I) allows the soft commutation

of the rectifier diodes in the secondary side, while the

circulating current is relatively large. The circulating

current increases more as the operation frequency moves

downward from the resonant frequency. Meanwhile,

operation above the resonant frequency (case II) allows

the circulating current to be minimized, but the rectifier

diodes are not softly commutated. Below-resonance

operation is preferred for high output voltage

applications, such as street LED lighting systems where

the reverse-recovery loss in the rectifier diode is severe.

Below-resonance operation has a narrow frequency range

with respect to the load variation since the frequency is

limited below the resonance frequency even at no-load

condition.

Figure 11. Waveforms of Each Operation Mode

Required Maximum Gain and Peak Gain

Above the peak gain frequency, the input impedance of

the resonant network is inductive and the input current of

the resonant network (I_p) lags the voltage applied to the

resonant network (V_d). This permits the MOSFETs to

turn on with zero voltage (ZVS), as illustrated in Figure

12. Meanwhile, the input impedance of the resonant

network becomes capacitive and I_pleads V_dbelow the

peak gain frequency. When operating in capacitive

region, the MOSFET body diode is reverse recovered

during the switching transition, which results in severe

noise. Another problem of entering the capacitive region

is that the output voltage becomes out of control since the

slope of the gain is reversed. The minimum switching

frequency should be limited above the peak gain

frequency.

On the other hand, above-resonance operation has less

conduction loss than the below-resonance operation. It

can show better efficiency for low output voltage

applications, such as Liquid Crystal Display (LCD) TV or

laptop adaptor, where Schottky diodes are available for

the secondary-side rectifiers and reverse-recovery

problems are insignificant. However, operation above the

resonant frequency may cause too much frequency

increase at light-load condition. Above-frequency

operation requires frequency skipping to prevent too

much increase of the switching frequency.

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AN-9729

APPLICATION NOTE

Even though the peak gain at a given condition can be

obtained using the gain in Equation (6), it is difficult to

express the peak gain in explicit form. To simplify the

analysis and design, the peak gains are obtained using

simulation tools and depicted in Figure 14, which shows

how the peak gain (attainable maximum gain) varies with

Q for different m values. It appears that higher peak gain

can be obtained by reducing m or Q values. With a given

resonant frequency (f_o) and Q value, decreasing m means

reducing the magnetizing inductance, which results in

increased circulating current. There is a trade-off between

the available gain range and conduction loss.

M

Inductive

Region

Capacitive

Region

Peak Gain

f_s

V_d

I_p

V_d

I_p

2.2

2.1

2

I_DS1

1.9

1.8

1.7

1.6

1.5

Reverse Recovery

ZVS

Figure 12. Operation Waveforms for Capacitive

and Inductive Regions

The available input voltage range of the LLC resonant

converter is determined by the peak voltage gain. Thus,

the resonant network should be designed so that the gain

curve has an enough peak gain to cover the input voltage

range. However, ZVS condition is lost below the peak

gain point, as depicted in Figure 12. Therefore, some

margin is required when determining the maximum gain

to guarantee stable ZVS operation during the load

transient and startup. Typically 10~20% of the maximum

gain is used as a margin, as shown in Figure 13.

m=2.25

1.4

m=2.5

1.3

m=3.0

m=3.5

m=4.0

1.2

m=4.5

m=5.0

m=6.0

m=8.0 m=7.0

1.1

m=9.0

0.5

1

Gain (M)

0.3

0.2

0.4

0.6

0.7

0.8

0.9

1

1.1

1.2

1.3

1.4

Q

Peak Gain

10~20% of M^max

Maximum Operation

Figure 14. Peak Gain (Attainable Maximum Gain)

Gain

vs. Q for Different m Values

(M^max

)

f_s

f_o

Figure 13. Determining the Maximum Gain

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APPLICATION NOTE

Features of FLS-XS Series

Table 1. Pin Description

FLS-XS series is an integrated Pulse Frequency

Modulation (PFM) controller and MOSFETs specifically

designed for Zero Voltage Switching (ZVS) half-bridge

converters with minimal external components. The

internal controller includes an under-voltage lockout,

optimized high-side / low-side gate driver, temperature-

compensated precise current controlled oscillator, and

self-protection circuitry. Compared with discrete

MOSFET and PWM controller solutions, FLS-XS series

can reduce total cost, component count, size, and weight;

while simultaneously increasing efficiency, productivity,

and system reliability.

Pin#

Name

Description

This pin is the drain of the high-side

MOSFET, typically connected to the

input DC link voltage.

1

V_DL

This pin is for discharging the external

soft-start capacitor when any

protections are triggered. When the

voltage of this pin drops to 0.2V, all

protections are reset and the controller

starts to operate again.

2

AR

This pin is to program the switching

frequency. Typically, opto-coupler and

resistor are connected to this pin to

regulate the output voltage.

3

4

R_T

This pin is to sense the current flowing

through the low-side MOSFET.

Typically negative voltage is applied

on this pin.

CS

5

6

SG

PG

This pin is the control ground.

This pin is the power ground. This pin

is connected to the source of the low-

side MOSFET.

This pin is the supply voltage of the

control IC.

7

8

9

LV_CC

NC

No connection.

Figure 15. Package Diagram

This pin is the supply voltage of the

high-side drive circuit.

HV_CC

This pin is the drain of the low-side

MOSFET. Typically transformer is

connected to this pin.

10

V_CTR

LV_CC

7

V_DL

1

V_REF

9

HV_CC

I_RT

LV_CCGood

Internal

Bias

V_REF

3V

1V

S

R

I_RT

Q

2I_RT

LUV+ / LUV-

HUV+ / HUV-

2V

Level

Shifter

High-Side

Gate Driver

Time

Delay

R_T

3

2

350ns

10

V_CTR

Divider

Balancing

Delay

Low-Side

Gate Driver

Time

Delay

AR

V_CssH/ V_CssL

5k

350ns

S

Q

R

Shutdown

LV_CCgood

TSD

V_AOCP

LV_CC

V_OVP

Delay

50ns

6

5

PG

SG

V_OCP

Delay

-1

1.5

s

4

CS

Figure 16. Functional Block Diagram of FSFR-Series

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APPLICATION NOTE

Figure 17. Reference Circuit for Design Example of LLC Resonant Half-Bridge Converter

Design Procedure

In this section, a design procedure is presented using the

schematic in Figure 17 as a reference. An integrated

transformer with center tap, secondary side is used and

input is supplied from Power Factor Correction (PFC) pre-

regulator. A DC-DC converter with 100W/100V output

has been selected as a design example. The design

specifications are as follows:

Even though the input voltage is regulated as constant by

PFC pre-regulator, it drops during the hold-up time. The

minimum input voltage considering the hold-up time

requirement is given as:

2P T_HU

min

(13

)

V_in V_O.PFC²

in

C_DL

.

Nominal input voltage: 400V_DC(output of PFC

stage)

where V_O.PFCis the nominal PFC output voltage, T_HUis

a hold-up time, and C_DLis the DC link bulk capacitor.

.

Output: 100V/1A (100W)

Hold-up time requirement: 30ms (50Hz line freq.)

DC link capacitor of PFC output: 240µF

(Design Example) Assuming the efficiency is 92%,

P

100

o

P 



 109W

in

E _ff

0.92

[STEP-1] Define System Specifications

max

V

V_O.PFC 400V

in

Estimated Efficiency (E_ff): The power conversion

efficiency must be estimated to calculate the maximum

input power with a given maximum output power. If no

reference data is available, use E_ff= 0.88~0.92 for low-

voltage output applications and E_ff= 0.92~0.96 for high-

voltage output applications. With the estimated efficiency,

the maximum input power is given as:

2P T_HU

min

2

in

V_in V_O.PFC



C_DL

3

21093010

2



400



 364V

6

24010

[STEP-2] Determine Maximum and Minimum

Voltage Gains of the Resonant Network

P

o

(11

)

P 

in

As discussed in the previous section, it is typical to operate

the LLC resonant converter around the resonant frequency

(f_o) to minimize switching frequency variation. Since the

input of the LLC resonant converter is supplied from PFC

output voltage, the converter should be designed to operate

at f_ofor the nominal PFC output voltage.

E_ff

min

Input Voltage Range (V_inand V_in^max): The maximum

input voltage would be the nominal PFC output voltage as:

max

(12

)

V

V_O.PFC

in

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

8

AN-9729

APPLICATION NOTE

As observed in Equation (10), the gain at f_ois a function of

m (m=L_p/L_r). The gain at f_ois determined by choosing that

value of m. While a higher peak gain can be obtained with

a small m value, too small m value results in poor coupling

of the transformer and deteriorates the efficiency. It is

typical to set m to be 3~7, which results in a voltage gain

of 1.1~1.2 at the resonant frequency (f_o).

[STEP-4] Calculate Equivalent Load

Resistance

With the transformer turns ratio obtained from Equation

(16), the equivalent load resistance is obtained as:

8n²V_o²

R_ac

(17)

 ²

P

o

With the chosen m value, the voltage gain for the nominal

PFC output voltage is obtained as:

(Design Example)

(V_oV_F)²

8n²

 ²

8 2.22²100.9²

 ²100

R_ac



 405

P

m

o

M ^min



@f=f_o

(14)

m 1

[STEP-5] Design the Resonant Network

which would be the minimum gain because the nominal

With m value chosen in STEP-2, read proper Q value from

the peak gain curves in Figure 14 that allows enough peak

gain. Considering the load transient and stable zero-

voltage-switching (ZVS) operation, 10~20% margin

should be introduced on the maximum gain when

determining the peak gain. Once the Q value is

determined, the resonant parameters are obtained as:

1

PFC output voltage is the maximum input voltage (V_in^max).

The maximum voltage gain is given as:

max

V

M ^max



M ^min

in

(15)

min

V

in

C_r

(18)

(Design Example) The ratio (m) between L_pand L_ris

chosen as 5. The minimum and maximum gains are

obtained as:

2Q f_o R_ac

1

L_r

(19)

(20)

(2 f_o)²C_r

V

m

5

min

RO

max

M



1.12

m 1

51

V

in

L_p m L_r

2

max

min

V

400

max

min

in

M



m



1.12  1.23

364

V

in

(Design Example)

From STEP-2, the maximum voltage gain (M ^max) for the

minimum input voltage (V_in^min) is 1.23. With 15% margin, a

peak gain of 1.41 is required. m has been chosen as 5 in

STEP-2 and Q is obtained as 0.42 from the peak gain curves

in Figure 19. By selecting the resonant frequency as 100kHz,

the resonant components are determined as:

Gain (M)

Peak Gain

(Available Maximum Gain)

1.23

min

M^max

for V_IN

for

V_IN

1

max

C



 9.35nF

1.12

r

M^min

3

2Q  f  R

2 0.42 10010  405

o

ac

( V_O.PFC

)

1

L_r



 271H

(2f_o)²C_r(2 10010³)²9.3510^9

m

M 

1.12

m 1

L_p m L_r1355H

f_s

f_o

Figure 18. Maximum Gain / Minimum Gain

[STEP-3] Determine the Transformer Turns

Ratio (n=N_p/N_s)

With the minimum gain (M^min) obtained in STEP-2, the

transformer turns ratio is given as:

max

N_p

V

n 



 M ^min

in

(16)

N_s2(V_oV_F)

where V_Fis the secondary-side rectifier diode voltage

drop.

(Design Example) assuming V_Fis 0.9V,

Figure 19. Resonant Network Design Using the Peak Gain

max

N_p

V_in

400

(Attainable Maximum Gain) Curve for m=5

n 



 M_min



1.12  2.22

N_s2(V_OV_F)

2(100  0.9)

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

9

AN-9729

APPLICATION NOTE

[STEP-6] Design the Transformer

The worst case for the transformer design is the minimum

switching frequency condition, which occurs at the

minimum input voltage and full-load condition. To obtain

the minimum switching frequency, plot the gain curve

using gain Equation 9 and read the minimum switching

frequency. The minimum number of turns for the

transformer primary-side is obtained as:

n(V_oV_F)

min

N_p



(21)

2 f_s^min M_V B  A

e

where A_eis the cross-sectional area of the transformer

core in m²and B is the maximum flux density swing in

Tesla, as shown in Figure 20. If there is no reference

data, use B =0.3~0.4 T.

Figure 21. Gain Curve

n (V_o+V_F)/M_V

V

^RI1/(2f_s)

[STEP-7] Transformer Construction

Parameters L_pand L_rof the transformer were determined

in STEP-5. L_pand L_rcan be measured in the primary side

with the secondary-side winding open circuited and short

circuited, respectively. Since LLC converter design

requires a relatively large L_r, a sectional bobbin is typically

used, as shown in Figure 22, to obtain the desired L_rvalue.

For a sectional bobbin, the number of turns and winding

configuration are the major factors determining the value

of L_r, while the gap length of the core does not affect L_r

much. L_pcan be controlled by adjusting the gap length.

Table 2 shows measured L_pand L_rvalues with different

gap lengths. A gap length of 0.05mm obtains values for L_p

and L_rclosest to the designed parameters.

-n (V_o+V_F)/M_V

B

B

Figure 20. Flux Density Swing

Choose the proper number of turns for the secondary side

that results in primary-side turns larger than N_p^minas:

min

N_p n N_s N_p

(22)

N p

N _s2

N _s1

(Design Example) EER3542 core (A_e=107mm²) is

selected for the transformer. From the gain curve of Figure

21, the minimum switching frequency is obtained as

70KHz. The minimum primary-side turns of the

transformer is given as:

n(V_oV_F)

2 f_s^minB 1.11 A_e

min

N _p



2.22100.9

28010³0.41.1110710^6

Figure 22. Sectional Bobbin



 30 turns

min

Choose N_sso that the resultant N_pis larger than N_p

:

min

Table 2. Measured L_pand L_rwith Different Gap Lengths

N _p n N_s 2.2213  29  N _p

min

N _p n N_s 2.2214  31 N _p

Gap Length

Lp

Lr

min

0.0mm

0.05mm

0.10mm

0.15mm

0.20mm

0.25mm

2,295μH

943μH

630μH

488μH

419μH

366μH

123μH

122μH

118μH

117μH

115μH

114μH

N _p n N_s 2.2215  33  N _p

min

N _p n N_s 2.2216  36  N _p

min

N _p n N_s 2.2217  38  N _p

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

10

AN-9729

APPLICATION NOTE

The nominal voltage of the resonant capacitor in normal

operation is given as:

(Design Example)

Final Resonant Network Design

max

RMS

V

2  I_Cr

2  f_oC_r

nom

in

Even though the integrated transformer approach in LLC

resonant converter design can implement the magnetic

components in a single core and save one magnetic

component, the value of L_ris not easy to control in real

transformer design. Resonant network design sometimes

requires iteration with a resultant L_rvalue after the

transformer is built. The resonant capacitor value is also

changed since it should be selected among off-the-shelf

capacitors. The final resonant network design is

summarized in Table 3 and the new gain curves are shown

in Figure 23.

V_C





(24)

r

2

However, the resonant capacitor voltage increases much

higher at overload condition or load transient. Actual

capacitor selection should be based on the Over-Current

Protection (OCP) trip point. With the OCP level, IOCP, the

maximum resonant capacitor voltage is obtained as:

max

V

I_OCP

nom

in

V_C





(25)

r

2

2  f_oC_r

(Design Example)

Table 3. Final Resonant Network Design Parameters

I_O

n(V_oV_F)

1

RMS

2

]²[

]

Parameters Initial Design Final Design

I_C



[

r

E _ff

2 2n

4 2 f_oM_v(L_p L_r)

L_p

L_r

1365µH

273H

9.3nF

100kHz

5

850µH

170µH

15nF

99.7kHz

5

1

 1

2 2  2.22

2.22(100  0.9)

2

]



[

]

[

C_r

3

 6

0.92

4 2 9910 1.1268010

f_o

=0.78A

m

Q

0.42

0.26

The peak current in the primary side in normal operation is:

peak

rms

M@f_o

1.12

I_C



2  I_C

 1.103A

r

Minimum

Frequency

80kHz

peak

OCP level is set to 1.75A with 50% margin on I_Cr

:

RMS

max

2  I_C

2.0

1.8

V_in

nom

r

100% load

V_C





r

80% load

60% load

2

2  f_oC_r

40% load

20% load

400

2

2 0.78





 318V

2 9910³1510^9

1.6

1.4

1.2

1.0

0.8

0.6

f ^min

f ^normal

max

V_in

I_OCP

max

V_C





r

2

2  f_oC_r

400

2

1.75

2  9910³1510^9





 387.7V

M ^max

M ^min

A 630V rated low-ESR film capacitor is selected for the

resonant capacitor.

[STEP-9] Rectifier Network Design

When the center tap winding is used in the transformer

secondary side, the diode voltage stress is twice of the

output voltage expressed as:

60

70

80

90

100

110

120

130

140

40

50

Frequency (KHz)

Figure 23. Gain Curve of the Final Resonant

Network Design

V_D 2(V_oV_F)

(26)

[STEP-8] Select the Resonant Capacitor

The RMS value of the current flowing through each

rectifier diode is given as:

When choosing the resonant capacitor, the current rating

should be considered because a considerable amount of

current flows through the capacitor. The RMS current

through the resonant capacitor is given as:



4

RMS

I_D



I_o

(27)

Meanwhile, the ripple current flowing through output

capacitor is given as:

1

 I_o

2 2n

n(V_oV_F)

4 2 f_oM_V(L_p L_r)

RMS

2

]

I_C



[

]²[

(23)

r

 I_o

 ²8

E_ff

RMS

2

I_Co

 (

)² I_o



I_o

(28)

8

2 2

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

11

AN-9729

APPLICATION NOTE

The voltage ripple of the output capacitor is:



2

(29

)

V_o I_o R_C

where R_Cis the effective series resistance (ESR) of the

output capacitor and the power dissipation is the output

capacitor is:

 (I_Co^RMS)² R_C

(30

)

P

Loss.Co

(Design Example) The voltage stress and current stress of

the rectifier diode are:

V_D 2(V_oV_F)  2(100  0.9)  201.8V

Figure 24. Typical Circuit Configuration for RT Pin



4

RMS

I_D



I_o 0.785A

Soft-Start To prevent excessive inrush current and

overshoot of output voltage during startup, increase the

voltage gain of the resonant converter progressively. Since

the voltage gain of the resonant converter is reversely

proportional to the switching frequency, soft-start is

implemented by sweeping down the switching frequency

from an initial high frequency (f ^ISS) until the output

voltage is established, as illustrated in Figure 25. The soft-

start circuit is made by connecting RC series network on

the RT pin as shown in Figure 24. FLS-XS series also has

an internal soft-start for 3ms to reduce the current

overshoot during the initial cycles, which adds 40KHz to

the initial frequency of the external soft-start circuit, as

shown in Figure 25. The actual initial frequency of the

soft-start is given as:

The 600V/8A Ultra fast recovery diode is selected for the

rectifier, considering the voltage overshoot caused by the

stray inductance.

The RMS current of the output capacitor is:

 ²8

8

I_o

RMS

2

I_C



(

)

 I_o



I_o 0.48A

o

2 2

When two electrolytic capacitors with ESR of 100m are

used in parallel, the output voltage ripple is given as:



2



2

0.1

V_o



I_o R_C



1( )  0.079V

2

The loss in electrolytic capacitors is:

P_Loss,C (I_C^RMS)² R_C 0.48²0.05  0.01W

o

5.2k 5.2k

f ^ISS (



)100 40 (kHz)

(33

)

[STEP-10] Control Circuit Configuration

R_min

R_SS

It is typical to set the initial frequency of soft-start (f ^ISS) as

2~3 times of the resonant frequency (f_o).

Figure 24 shows the typical circuit configuration for the RT

pin of FLS-XS series, where the opto-coupler transistor is

connected to the RT pin to control the switching frequency.

The minimum switching frequency occurs when the opto-

coupler transistor is fully tuned off, which is given as:

The soft-start time is determined by the RC time constant:

(34

)

T_SS 3 ~ 4 timesof R_SSC_SS

5.2k

R_min

f_min



100(kHz)

(31)

Assuming the saturation voltage of the opto-coupler

transistor is 0.2V, the maximum switching frequency is

determined as:

5.2k 4.68k

f_max (



)100(kHz)

(32)

R_min

R_max

Figure 25. Frequency Sweep of the Soft-Start

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

12

AN-9729

APPLICATION NOTE

(Design Example) The minimum frequency is 80kHz in

STEP-6. R_minis determined as:

(Design Example) Since the OCP level is determined as

1.75A in STEP-8 and the OCP threshold voltage is -0.6V,

a sensing resistor of 0.33 is used. The RC time constant

is set to 100ns (1/100 of switching period) with

1kresistor and 100pF capacitor.

100KHz

R_min



 5.2K  6.5K

f_min

Considering the output voltage overshoot during transient

(10%) and the controllability of the feedback loop, the

maximum frequency is set as 140kHz. R_maxis determined as:

[STEP-12] Voltage and Current Feedback

Power supplies for LED lighting must be controlled by

Constant Current (CC) Mode as well as a Constant

Voltage (CV) Mode. Because the forward-voltage drop of

LED varies with the junction temperature and the current

also increases greatly consequently, devices can be

damaged.

Figure 28 shows an example of a CC and CV Mode

feedback circuit for single output LED power supply.

During normal operation, CC Mode is dominant and CV

control circuit does not activate as long as the feedback

voltage is lower than reference voltage, which means that

CV control circuit only acts as OVP for abnormal modes.

4.68K

R_max



f_o1.40

100KHz

4.68K

5.2K

R_min

(



)



 7.8K

99KHz 1.40 5.2K

(



)

100KHz

6.5K

Setting the initial frequency of soft-start as 250kHz (2.5

times of the resonant frequency), the soft-start resistor R_SS

is given as:

5.2K

R_SS



f_ISS 40KHz

100KHz

5.2K

R_min

(



)

(Design Example) The output voltage (V_O) is 100V in

design target. V_Ois determined as:

5.2K



 4K

250KHz  40KHz 5.2K

(



)

R_FU

100KHz

6.5K

V_o 2.5(1

)

R_FL

Set the upper-side feedback resistance (R_FU) as 330K.

R_FLis determined as:

[STEP-11] Current Sensing and Protection

FLS-XS series senses low-side MOSFET drain current as

a negative voltage, as shown in Figure 26 and Figure 27.

Half-wave sensing allows low-power dissipation in the

sensing resistor, while full-wave sensing has less

switching noise in the sensing signal. Typically, RC low-

pass filter is used to filter out the switching noise in the

sensing signal. The RC time constant of the low-pass filter

should be 1/100~1/20 of the switching period.

2.5 R_FU

(V_o 2.5)

2.5 330K

(100  2.5)

R_FL



 8.46K

The output voltage of op-amp is given as:

V_senseV_REF



 sC201V_OC

 sC201

R201 R203

0 

1



R201 R203

V_senseV_REF

1

V_OC 

(



)

sC201 R201 R203

Actually, the V_sensehas a negative value and assume all

resistors have the same value for simplification;

1

V_OC 

(V_senseV_REF)

sC201 R

The output voltage of the op-amp for CC control keeps

zero voltage as long as the sensing voltages are lower than

the reference voltage.

Figure 26. Half-Wave Sensing

Figure 27. Full-Wave Sensing

Figure 28. Example of CC and CV Feedback Circuit

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

13

AN-9729

APPLICATION NOTE

Figure 29 shows another example of a CC and Over-

Voltage Regulation (OVR) Mode feedback circuit for

multi-output LED power supply. The FAN7346 is a LED

current-balance controller that controls four LED arrays to

maintain equal LED current. To prevent LED driving

voltage being over the withstanding voltage of component,

the FAN7346 controls LED driving voltage. The OVR

control circuit activates when the ENA pin is in HIGH

state. If OVR pin voltage is lower than 1.5V, the Feedback

Control (FB) pin voltage follows headroom control to

maintain minimum voltage of drain voltages as 1V. If

OVR pin voltage is higher than 1.5V, the FAN7346

controls FB (FB is pulled LOW) through FB regulation so

the OVR pin voltage is not over 1.5V.

thermal stress problems in the other channel. OLP function

has auto-recovery: As soon as drain voltage is higher than

0.3V, OLP is finished and drain voltage feedback system

is restored.

To sense over-current condition, the FAN7346 monitors

FBx pin voltage. If FBx voltage is higher than 1V for

20µs, CHx is considered in over-current condition. After

sensing OCP condition, individual channel switch is

latched off. So, even if a channel is in OCP condition,

other channels keep operating. Any OCP channel is

restarted after UVLO is reset.

(Design Example) The output voltage (V_O) is 100V in

design target. V_Ois determined as:

LED current is controlled by FBx pin voltage. The

external current balance switch is operating in linear

region to control LED current. Sensed voltage at the FBx

pin is compared with internal reference voltage and

controller signals the gate (or base) for external current

balance switch. Internal reference voltage is made from

ADIM voltage. The LED current is determined as:

R8

V_o 1.5(1

)

R10

Set the upper-side feedback resistance (R8) as 1M. R10

is determined as:

1.5 R8

(V_o1.5) (100 1.5)

1.51M

R10 



 15.23K

V_ADIM

The output channel current (I_LED) is 250mA in design target.

Setting the V^ADIMis above 4V, the current sense R_SENSEis

determined as:

I_LED



(35)

10 R_SENSE

ADIM voltage is clamped internally from 0.5V to 4V. The

protections; such as open LED Protection (OLP), Short

LED Protection (SLP), and Over-Current Protection

(OCP); which increase system reliability, are applied in

individual string protection method.

V_ADIM

4V

R_SENSE



 1.6

10 I_LED10 250mA

Choose the sense resistor (R29, R30, R31, and R32) is

To sense a short LED condition, the FAN7346 senses

drain voltage level. If LEDs are shorted, the LED forward

voltage is lower than other LED strings, so its drain

voltage of external balance switch is higher than other

drain voltage. The SLP condition detection threshold

voltage can be programmed by SLPR voltage. The internal

short LED protection reference is determined as:

1.5  , the OCP level is determined as:

V_{OCP _TH}

1V

I_OCP



 666mA

R_SENSE

1.5

V_{SLP _TH} 10V_SLPR

(36)

Minimum SLP threshold voltage is 0V and maximum SLP

threshold voltage is 45V. If any string is in SLP condition,

SLP string is turned off and other string is operated

normally. If the sensed drain voltage (CHx voltage) is

higher than the programmed threshold voltage for 20µs,

CHx goes to short LED protection. As soon as

encountering SLP, the corresponding channel is forced off.

To sense an open LED condition, the FAN7346 senses

drain voltage level. If LED string is opened, its drain

voltage of external balance switch is grounded, so the

FAN7346 detects the open-LED condition. The detection

threshold voltage is 0.3V. If CHx voltage is lower than

0.3V for 20µs, its drain voltage feedback is pulled up to

5V. This means the opened LED string is eliminated from

drain feedback loop. Without OLP, minimum drain

voltage is 0V, so drain voltage feedback forces the FB

signal to increase output power. This can cause SLP or

Figure 29. Example of CC and OVR Feedback Circuit

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

14

AN-9729

APPLICATION NOTE

Design Summary

design example. EER3543 core with sectional bobbin is

used for the transformer. Efficiency at full-load is around

94%.

Figure 30 and Figure 31 show the final schematic of the

LLC resonant half-bridge converter for LED lighting

Figure 30. Final Schematic of Half-Bridge LLC Resonant Converter for Single Channel

Figure 31. Final Schematic of Half-Bridge LLC Resonant Converter for Multi Channel

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

15

AN-9729

APPLICATION NOTE

V_CR[200V/div]

300V

Experimental Verification

To show the validity of the design procedure presented in

this application note, the converter of the design example

was built and tested. All the circuit components are used as

designed in the design example.

V_DS[200V/div]

Figure 32 and Figure 33 show the operation waveforms at

full-load and no-load conditions for nominal input voltage.

As observed, the MOSFET drain-to-source voltage (V_DS)

drops to zero by resonance before the MOSFET is turned

on and zero voltage switching is achieved.

1.1A

I_P[1A/div]

Time (5µs/div)

Figure 34. Resonant Capacitor Voltage and Primary-

Side Current Waveforms at Full-Load Condition

Figure 34 shows the waveforms of the resonant capacitor

voltage and primary-side current at full-load condition.

The peak values of the resonant capacitor voltage and

primary-side current are 300V and 1.1A, respectively,

which are well matched with the calculated values in

STEP-8 of design procedure section.

I_P[1A/div]

Figure 35 shows the rectifier diode voltage and current

waveforms at full-load condition. Due to the voltage

overshoot caused by stray inductance, the voltage stress is

a little bit higher than the value calculated in STEP-9.

I_D[1A/div]

257V

Figure 36 shows the output load current and output voltage

of op-amp waveforms for constant-current control when

output load is step changed from 140mA to 1000mA at t₀.

V_D[100V/div]

Figure 37 shows the operation waveform when LED string

is opened and restored condition

Time (5µs/div)

Figure 35. Rectifier Diode Voltage and Current

Waveforms at Full-Load Condition

I_P[1A/div]

V_{OP_CC}[2V/div]

I_DS[1A/div]

V_DS[200V/div]

1000mA

CC Mode

Transient Mode

140mA

Time (5µs/div)

t₁

t₀

I_LOAD[0.5A/div]

Time (50ms/div)

Figure 32. Operation Waveforms at Full-Load Condition

Figure 36. Soft-Start Waveforms

I_P[2A/div]

V_LED

[100V/div]

Open LED

Restore LED

I_{LED_OPEN}

[200mA/div]

I_DS[1A/div]

V_{DS_NORMAL_LED}

[500mV/div]

V_DS[200V/div]

V_{DSD_OPEN_LED}

[500mV/div]

Time (5µs/div)

Figure 33. Operation Waveforms at No-Load Condition

Figure 37. Open LED Protection Operation

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

16

AN-9729

APPLICATION NOTE

References

[1] Robert L. Steigerwald, “A Comparison of Half-bridge

resonant converter topologies,” IEEE Transactions on

Power Electronics, Vol. 3, No. 2, April 1988.

[7] M. Emsermann, “An Approximate Steady State and Small

Signal Analysis of the Parallel Resonant Converter Running

Above Resonance,” Proc. Power Electronics and Variable

Speed Drives ’91, 1991, pp. 9-14.

[2] A. F. Witulski and R. W. Erickson, “Design of the series

resonant converter for minimum stress,” IEEE Transactions

on Aerosp. Electron. Syst., Vol. AES-22, pp. 356-363,

July 1986.

[8] Yan Liang, Wenduo Liu, Bing Lu, van Wyk, J.D, “Design

of integrated passive component for a 1MHz 1kW half-

bridge LLC resonant converter,” IAS 2005, pp. 2223-2228.

[9] B. Yang, F.C. Lee, M. Concannon, “Over-current protection

[3] R. Oruganti, J. Yang, and F.C. Lee, “Implementation of

Optimal Trajectory Control of Series Resonant Converters,”

Proc. IEEE PESC ’87, 1987.

methods for LLC resonant converter” APEC 2003, pp. 605 - 609.

[10] Yilei Gu, Zhengyu Lu, Lijun Hang, Zhaoming Qian,

Guisong Huang, “Three-level LLC series resonant DC/DC

converter,” IEEE Transactions on Power Electronics

Vol.20, July 2005, pp.781 – 789.

[4] V. Vorperian and S. Cuk, “A Complete DC Analysis of the

Series Resonant Converter,” Proc. IEEE PESC’82, 1982.

[5] Y. G. Kang, A. K. Upadhyay, D. L. Stephens, “Analysis and

design of a half-bridge parallel resonant converter operating

above resonance,” IEEE Transactions on Industry

[11] Bo Yang, Lee, F.C, A.J Zhang, Guisong Huang, “LLC

resonant converter for front-end DC/DC conversion,” APEC

2002. pp.1108 – 1112.

Applications, Vol. 27, March-April 1991, pp. 386 – 395.

[12] Bing Lu, Wenduo Liu, Yan Liang, Fred C. Lee, Jacobus D.

Van Wyk, “Optimal design methodology for LLC Resonant

Converter,” APEC, 2006, pp.533-538.

[6] R. Oruganti, J. Yang, and F.C. Lee, “State Plane Analysis of

Parallel Resonant Converters,” Proc. IEEE PESC ’85, 1985.

This application note based on Fairchild Semiconductor Application Note AN-4137.

Related Datasheets

FLS1800XS — Half-Bridge LLC Resonant Control IC for Lighting

FAN7346 — 4-Channel LED Current Balance Control IC

DISCLAIMER

FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY PRODUCTS

HEREIN TO IMPROVE RELIABILITY, FUNCTION, OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY LIABILITY ARISING OUT OF

THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN; NEITHER DOES IT CONVEY ANY LICENSE

UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS.

LIFE SUPPORT POLICY

FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR

SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR CORPORATION.

As used herein:

1. Life support devices or systems are devices or systems

which, (a) are intended for surgical implant into the body, or

(b) support or sustain life, or (c) whose failure to perform

when properly used in accordance with instructions for use

provided in the labeling, can be reasonably expected to

result in significant injury to the user.

2. A critical component is any component of a life support

device or system whose failure to perform can be

reasonably expected to cause the failure of the life support

device or system, or to affect its safety or effectiveness.

Rev. 1.0.1 • 11/16/12

www.fairchildsemi.com

17


型号：	AN-9729
厂家：	FAIRCHILD SEMICONDUCTOR
描述：	LED Application Design Guide Using Half-Bridge LLC Resonant Converter for 100W Street Lighting LED应用设计指南使用半桥LLC谐振转换器为100W路灯转换器
文件：	总17页 (文件大小：1283K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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