LTC3890EGN-1#PBF [Linear]
暂无描述;型号: | LTC3890EGN-1#PBF |
厂家: | Linear |
描述: | 暂无描述 |
文件: | 总36页 (文件大小:876K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LTC3890-1
60V Low I , Dual, 2-Phase
Q
Synchronous Step-Down
DC/DC Controller
DESCRIPTION
FEATURES
The LTC®3890-1 is a high performance dual step-down
switching regulator DC/DC controller that drives all
N-channelsynchronouspowerMOSFETstages.Aconstant
frequency current mode architecture allows a phase-
lockablefrequencyofupto850kHz.Powerlossandsupply
noiseareminimizedbyoperatingthetwocontrolleroutput
stages out-of-phase.
n
Wide V Range: 4V to 60V (65V Abs Max)
IN
n
Low Operating I : 50µA (One Channel On)
Q
n
n
n
Wide Output Voltage Range: 0.8V ≤ V
≤ 24V
OUT
R
SENSE
or DCR Current Sensing
Out-of-Phase Controllers Reduce Required Input
Capacitance and Power Supply Induced Noise
Phase-Lockable Frequency (75kHz to 850kHz)
Programmable Fixed Frequency (50kHz to 900kHz)
Selectable Continuous, Pulse-Skipping or Low Ripple
Burst Mode® Operation at Light Loads
Very Low Dropout Operation: 99% Duty Cycle
Adjustable Output Voltage Soft-Start or Tracking
Power Good Output Voltage Monitor
Output Overvoltage Protection
Low Shutdown I : <14µA
Internal LDO Powers Gate Drive from V or EXTV
No Current Foldback During Start-Up
Narrow SSOP Package
n
n
n
The50μAno-loadquiescentcurrentextendsoperatinglife
in battery-powered systems. OPTI-LOOP® compensation
allows the transient response to be optimized over a wide
range of output capacitance and ESR values. A wide 4V
to 60V input supply range encompasses a wide range of
intermediate bus voltages and battery chemistries.
n
n
n
n
n
n
n
n
Independent TRACK/SS pins for each controller ramp the
output voltages during start-up. Current foldback limits
MOSFET heat dissipation during short-circuit conditions.
ThePLLIN/MODEpinselectsamongBurstModeoperation,
pulse-skipping mode, or continuous conduction mode at
light loads.
Q
IN
CC
APPLICATIONS
Foraleadless32-pinQFNpackagewithadditionalfeatures
of adjustable current limit, clock out, phase modulation
and two PGOOD outputs, see the LTC3890 data sheet.
L, LT, LTC, LTM, Linear Technology, Burst Mode, OPTI-LOOP and the Linear logo are
registered trademarks of Linear Technology Corporation. All other trademarks are the property
of their respective owners. Protected by U.S. Patents including 5481178, 5705919, 5929620,
6100678, 6144194, 6177787, 6304066, 6580258, 7230497.
n
Automotive Always-On Systems
n
Battery Operated Digital Devices
n
Distributed DC Power Systems
TYPICAL APPLICATION
High Efficiency Dual 8.5V/3.3V Output Step-Down Converter
Efficiency and Power Loss
V
vs Output Current
IN
9V TO 60V
22µF
10000
1000
100
100
90
80
70
60
50
40
30
20
4.7µF
V
V
= 12V
IN
OUT
V
IN
INTV
CC
= 3.3V
TG1
TG2
0.1µF
0.1µF
BOOST1
SW1
BOOST2
SW2
4.7µH
8µH
BG1
BG2
LTC3890-1
PGND
10
+
+
SENSE1
SENSE1
SENSE2
0.01Ω
0.008Ω
1
–
–
V
8.5V
3A
SENSE2
10
0
OUT2
V
OUT1
3.3V
5A
V
V
FB1
FB2
0.1
10
100k
470µF
100k
10.5k
ITH1
ITH2
0.0001 0.001
0.01
0.1
1
1000pF
34.8k
1000pF
OUTPUT CURRENT (A)
330µF
TRACK/SS1 SGND TRACK/SS2
0.1µF
38901 TA01b
31.6k
34.8k
0.1µF
38901 TA01a
38901fb
1
LTC3890-1
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
TOP VIEW
Input Supply Voltage (V )......................... –0.3V to 65V
IN
Topside Driver Voltages
1
2
TRACK/SS1
PGOOD1
TG1
28
27
26
25
24
23
22
21
20
19
18
17
16
15
ITH1
BOOST1, BOOST2 ..................................–0.3V to 71V
Switch Voltage (SW1, SW2) ......................... –5V to 65V
(BOOST1-SW1), (BOOST2-SW2) ................ –0.3V to 6V
RUN1, RUN2 ............................................... –0.3V to 8V
Maximum Current Sourced into Pin
V
FB1
+
3
SENSE1
SENSE1
–
4
SW1
5
BOOST1
BG1
FREQ
PLLIN/MODE
SGND
6
7
V
IN
from Source >8V...................................................100µA
8
PGND
RUN1
+
–
+
–
SENSE1 , SENSE2 , SENSE1
9
EXTV
CC
RUN2
SENSE2 Voltages ..................................... –0.3V to 28V
–
10
11
12
13
14
INTV
CC
SENSE2
PLLIN/MODE, INTV Voltages ................... –0.3V to 6V
CC
+
BG2
SENSE2
FREQ Voltage........................................–0.3V to INTV
CC
BOOST2
SW2
V
FB2
EXTV ..................................................... –0.3V to 14V
CC
ITH2
ITH1, ITH2, V , V Voltages................... –0.3V to 6V
TG2
FB1 FB2
TRACK/SS2
PGOOD1 Voltage ......................................... –0.3V to 6V
TRACK/SS1, TRACK/SS2 Voltages ............. –0.3V to 6V
Operating Junction Temperature Range (Notes 2, 3)
LTC3890E-1, LTC3890I-1 ................... –40°C to 125°C
LTC3890H-1....................................... –40°C to 150°C
LTC3890MP-1 .................................... –55°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
GN PACKAGE
28-LEAD PLASTIC SSOP
T
= 150°C, θ = 90°C/W
JA
JMAX
ORDER INFORMATION
LEAD FREE FINISH
LTC3890EGN-1#PBF
LTC3890IGN-1#PBF
LTC3890HGN-1#PBF
LTC3890MPGN-1#PBF
TAPE AND REEL
PART MARKING*
LTC3890GN-1
LTC3890GN-1
LTC3890GN-1
LTC3890GN-1
PACKAGE DESCRIPTION
28-Lead Plastic SSOP
28-Lead Plastic SSOP
28-Lead Plastic SSOP
28-Lead Plastic SSOP
TEMPERATURE RANGE
–40°C to 125°C
–40°C to 125°C
–40°C to 150°C
–55°C to 150°C
LTC3890EGN-1#TRPBF
LTC3890IGN-1#TRPBF
LTC3890HGN-1#TRPBF
LTC3890MPGN-1#TRPBF
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
38901fb
2
LTC3890-1
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted. (Note 2)
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
V
V
Input Supply Operating Voltage Range
Regulated Feedback Voltage
4
60
V
IN
I
Voltage = 1.2V (Note 4)
TH1,2
FB1,2
–40°C to 85°C, All Grades
LTC3890E-1, LTC3890I-1,
LTC3890H-1, LTC3890MP-1
0.792
0.788
0.786
0.800
0.800
0.800
0.808
0.812
0.812
V
V
V
l
l
I
Feedback Current
(Note 4)
5
50
nA
FB1,2
V
V
Reference Voltage Line Regulation
Output Voltage Load Regulation
V
= 4.5V to 60V (Note 4)
IN
0.002
0.02
%/V
REFLNREG
LOADREG
(Note 4)
l
l
Measured in Servo Loop,
0.01
0.1
%
∆
Voltage = 1.2V to 0.7V
ITH
(Note 4)
Measured in Servo Loop,
–0.01
2
–0.1
%
∆
Voltage = 1.2V to 2V
ITH
g
m1,2
Transconductance Amplifier g
Input DC Supply Current
I
= 1.2V, Sink/Source = 5µA (Note 4)
mmho
m
TH1,2
I
Q
(Note 5)
Pulse-Skipping or Forced Continuous
Mode (One Channel On)
RUN1 = 5V and RUN2 = 0V, V = 0.83V or
2
2
mA
mA
µA
FB1
RUN1 = 0V and RUN2 = 5V, V = 0.83V
FB2
Pulse-Skipping or Forced Continuous
Mode (Both Channels On)
RUN1,2 = 5V, V
= 0.83V (No Load)
FB1,2
Sleep Mode (One Channel On)
RUN1 = 5V and RUN2 = 0V, V = 0.83V or
50
75
FB1
RUN1 = 0V and RUN2 = 5V, V = 0.83V
FB2
Sleep Mode (Both Channels On)
Shutdown
RUN1,2 = 5V, V
RUN1,2 = 0V
= 0.83V (No Load)
60
14
100
25
µA
µA
FB1,2
l
l
UVLO
Undervoltage Lockout
INTV Ramping Up
3.92
3.80
4.2
4.0
V
V
CC
INTV Ramping Down
3.6
7
CC
V
OVL
Feedback Overvoltage Protection
Measured at V
Each Channel
, Relative to Regulated V
FB1,2 FB1,2
10
13
1
%
+
–
+
I
I
SENSE Pin Current
µA
SENSE
SENSE
–
SENSE Pins Current
Each Channel
–
–
V
V
< INTV – 0.5V
1
µA
µA
SENSE
SENSE
CC
> INTV + 0.5V
700
99
CC
DF
MAX
Maximum Duty Factor
In Dropout
98
%
I
Soft-Start Charge Current
V
= 0V
TRACK1,2
0.7
1.0
1.4
µA
TRACK/SS1,2
l
l
V
RUN1
V
RUN2
On
On
RUN1 Pin On Threshold
RUN2 Pin On Threshold
V
RUN1
V
RUN2
Rising
Rising
1.15
1.20
1.21
1.25
1.27
1.30
V
V
V
V
Hyst RUN Pin Hysteresis
50
75
mV
mV
RUN1,2
l
Maximum Current Sense Threshold
V
= 0.7V, V
–, – = 3.3V, I = 0
64
85
SENSE(MAX)
FB1,2
SENSE1
2
LIM
Gate Driver
TG1,2
Pull-Up On-Resistance
Pull-Down On-Resistance
2.5
1.5
Ω
Ω
BG1,2
Pull-Up On-Resistance
Pull-Down On-Resistance
2.4
1.1
Ω
Ω
TG Transition Time:
Rise Time
(Note 6)
TG1,2 t
TG1,2 t
C
C
= 3300pF
25
25
ns
ns
r
f
LOAD
LOAD
Fall Time
= 3300pF
BG Transition Time:
Rise Time
(Note 6)
LOAD
LOAD
BG1,2 t
BG1,2 t
C
C
= 3300pF
= 3300pF
25
25
ns
ns
r
f
Fall Time
38901fb
3
LTC3890-1
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted. (Note 2)
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
TG/BG t
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
C
= 3300pF Each Driver
30
ns
1D
LOAD
BG/TG t
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
C
= 3300pF Each Driver
30
95
ns
ns
1D
LOAD
t
Minimum On-Time
(Note 7)
ON(MIN)
INTV Linear Regulator
CC
V
V
V
V
V
V
Internal V Voltage
6V < V < 60V, V = 0V
EXTVCC
4.85
4.85
4.5
5.1
0.7
5.1
0.6
4.7
250
5.35
1.1
V
%
V
INTVCCVIN
LDOVIN
CC
IN
INTV Load Regulation
I
CC
= 0mA to 50mA, V
= 0V
CC
EXTVCC
Internal V Voltage
6V < V < 13V
EXTVCC
5.35
1.1
INTVCCEXT
LDOEXT
CC
INTV Load Regulation
I
CC
= 0mA to 50mA, V
= 8.5V
%
V
CC
EXTVCC
EXTV Switchover Voltage
EXTV Ramping Positive
4.9
EXTVCC
CC
CC
EXTV Hysteresis
mV
LDOHYS
CC
Oscillator and Phase-Locked Loop
f
f
f
f
f
f
Programmable Frequency
Programmable Frequency
Programmable Frequency
Low Fixed Frequency
R
R
R
= 25k, PLLIN/MODE = DC Voltage
= 65k, PLLIN/MODE = DC Voltage
= 105k, PLLIN/MODE = DC Voltage
= 0V, PLLIN/MODE = DC Voltage
105
440
835
350
535
kHz
kHz
kHz
kHz
kHz
kHz
25kΩ
65kΩ
105kΩ
LOW
FREQ
FREQ
FREQ
FREQ
FREQ
375
505
V
V
320
485
75
380
585
850
High Fixed Frequency
= INTV , PLLIN/MODE = DC Voltage
CC
HIGH
SYNC
l
Synchronizable Frequency
PLLIN/MODE = External Clock
PGOOD1 Output
V
PGOOD1 Voltage Low
PGOOD1 Leakage Current
PGOOD1 Trip Level
I
= 2mA
= 5V
0.2
0.4
1
V
PGL
PGOOD
I
V
V
µA
PGOOD
PGOOD
V
PG
with Respect to Set Regulated Voltage
FB
V
Ramping Negative
–13
7
–10
2.5
–7
13
%
%
FB
Hysteresis
V
with Respect to Set Regulated Voltage
FB
V
Ramping Positive
10
2.5
%
%
FB
Hysteresis
t
PG
Delay for Reporting a Fault
25
µs
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Ratings for extended periods may affect device reliability and
lifetime.
Note 3: T is calculated from the ambient temperature T and power
J A
dissipation P according to the following formula:
D
T = T + (P • 90°C/W)
J
A
D
Note 4: The LTC3890-1 is tested in a feedback loop that servos V
to
ITH1,2
Note 2: The LTC3890-1 is tested under pulsed load condition such that
a specified voltage and measures the resultant V . The specification at
FB
T ≈ T .The LTC3890E-1 is guaranteed to meet performance specifications
85°C is not tested in production and is assured by design, characterization
and correlation to production testing at other temperatures (125°C for the
LTC3890E-1/LTC3890I-1, 150°C for the LTC3890H-1/LTC3890MP-1). For
the LTC3890MP-1, the specification at –40°C is not tested in production
and is assured by design, characterization and correlation to production
testing at –55°C.
J
A
from 0°C to 85°C. Specifications over the –40°C to 125°C operating
junction temperature range are assured by design, characterization and
correlation with statistical process controls. The LTC3890I-1 is guaranteed
over the –40°C to 125°C operating junction temperature range, the
LTC3890H-1 is guaranteed over the –40°C to 150°C operating junction
temperature range and the LTC3890MP-1 is tested and guaranteed over
the –55°C to 150°C operating junction temperature range.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See the Applications information section.
High junction temperatures degrade operating lifetimes; operating lifetime
is derated for junction temperatures greater than 125°C. Note that the
maximum ambient temperature consistent with these specifications is
determined by specific operating conditions in conjunction with board layout,
the rated package thermal impedance and other environmental factors.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 7: The minimum on-time condition is specified for an inductor peak-
to-peak ripple current ≥ of I
(See Minimum On-Time Considerations in
MAX
the Applications Information section).
38901fb
4
LTC3890-1
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency and Power Loss
vs Output Current
Efficiency vs Output Current
Efficiency vs Input Voltage
10000
1000
100
100
90
80
70
60
50
40
30
20
100
90
80
70
60
50
40
30
20
100
98
96
94
92
90
88
86
84
V
V
= 12V
IN
OUT
BURST EFFICIENCY
V
= 8.5V
OUT
= 3.3V
V
= 3.3V
OUT
V
= 8.5V
OUT2
CCM LOSS
BURST LOSS
PULSE-SKIPPING
LOSS
10
V
= 3.3V
OUT1
CCM EFFICIENCY
1
Burst Mode OPERATION
PULSE-SKIPPING
EFFICIENCY
10
0
10
0
82
80
V
IN
= 12V
I
= 2A
LOAD
5
0.1
10
0.0001 0.001
0.01
0.1
1
0.0001 0.001
0.01
0.1
1
10
0
10 15 20 25 30 35 40 45 50 55 60
INPUT VOLTAGE (V)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
38901 G01
38901 G02
38901 G03
FIGURE 13 CIRCUIT
FIGURE 13 CIRCUIT
FIGURE 13 CIRCUIT
Load Step
Burst Mode Operation
Load Step
Pulse-Skipping Mode
Load Step
Forced Continuous Mode
V
V
V
OUT
OUT
OUT
100mV/DIV
AC-
100mV/DIV
AC-
100mV/DIV
AC-
COUPLED
COUPLED
COUPLED
I
L
I
I
L
L
2A/DIV
2A/DIV
2A/DIV
38901 G04
38901 G05
38901 G06
50µs/DIV
50µs/DIV
50µs/DIV
V
V
= 12V
V
V
= 12V
V
V
= 12V
IN
OUT
IN
OUT
IN
OUT
= 3.3V
= 3.3V
= 3.3V
FIGURE 13 CIRCUIT
FIGURE 13 CIRCUIT
FIGURE 13 CIRCUIT
Inductor Current at Light Load
Soft Start-Up
Tracking Start-Up
FORCED
CONTINUOUS
MODE
V
V
OUT2
OUT2
2V/DIV
2V/DIV
Burst Mode
OPERATION
1A/DIV
V
V
OUT1
OUT1
2V/DIV
2V/DIV
PULSE-SKIPPING
MODE
38901 G09
38901 G08
38901 G07
2ms/DIV
FIGURE 13 CIRCUIT
2ms/DIV
FIGURE 13 CIRCUIT
5µs/DIV
V
V
LOAD
= 12V
IN
= 3.3V
OUT
I
= 200µA
38901fb
5
LTC3890-1
TYPICAL PERFORMANCE CHARACTERISTICS
Total Input Supply Current
vs Input Voltage
EXTVCC Switchover and INTVCC
Voltages vs Temperature
INTVCC Line Regulation
300
250
200
150
100
6.0
5.8
5.6
5.4
5.2
5.0
4.8
4.6
4.4
4.2
4.0
5.5
5.0
V
= 3.3V
OUT
FIGURE 13 CIRCUIT
INTV
CC
300µA LOAD
4.5
4.0
3.5
3.0
EXTV RISING
CC
EXTV FALLING
CC
NO LOAD
50
0
I
= 10mA
LOAD
–75 –50 –25
0
25 50 75 100 125 150
0 5
10 15 20 25 30 35 40 45 50 55 60 65
INPUT VOLTAGE (V)
5
10 15 20 25 30 35 40 45 50 55 60 65
INPUT VOLTAGE (V)
TEMPERATURE (°C)
38901 G10
38901 G12
38901 G11
Maximum Current Sense Voltage
vs ITH Voltage
Maximum Current Sense
Threshold vs Duty Cycle
SENSE– Pin Input Bias Current
800
700
600
500
400
300
200
100
0
80
60
40
20
90
85
80
75
70
65
60
5% DUTY CYCLE
PULSE-SKIPPING MODE
Burst Mode
OPERATION
0
–20
–40
FORCED CONTINUOUS MODE
–100
5
10
15
25
0.8
(V)
1.2
1.4
0
20
0
0.2
0.4 0.6
V
1.0
0
10 20 30 40 50 60 70 80 90 100
V
COMMON MODE VOLTAGE (V)
DUTY CYCLE (%)
SENSE
ITH
38901 G14
38901 G13
38901 G15
Foldback Current Limit
Quiescent Current vs Temperature
INTVCC vs Load Current
5.50
5.25
5.00
80
70
60
50
40
30
20
10
0
80
75
70
65
60
55
50
45
40
35
30
V
= 12V
V
= 12V
IN
IN
EXTV = 0V
CC
EXTV = 8.5V
CC
4.75
4.50
4.25
4.00
EXTV = 5V
CC
20
60
LOAD CURRENT (mA)
80
100
0
40
0
100 200 300 400 500 600 700 800
FEEDBACK VOLTAGE (MV)
38901 G16
–75
–50 –25
0
25 50
75 100 125 150
TEMPERATURE (°C)
38901 G18
38901 G17
38901fb
6
LTC3890-1
TYPICAL PERFORMANCE CHARACTERISTICS
Regulated Feedback Voltage
vs Temperature
TRACK/SS Pull-Up Current
vs Temperature
Shutdown (RUN) Threshold
vs Temperature
1.40
1.35
808
1.10
1.05
1.00
0.95
806
804
RUN1 RISING
1.30
1.25
1.20
1.15
1.10
1.05
1.00
RUN2 RISING
802
800
798
796
794
RUN1 FALLING
RUN2 FALLING
100
0.90
792
25 50 75
125 150
–50 –25
75 100 125 150
TEMPERATURE (°C)
–75 –50 –25
0
25 50 75 100 125 150
–75 –50 –25
0
–75
0
25 50
TEMPERATURE (°C)
TEMPERATURE (°C)
38901 G19
38901 G20
38901 G21
SENSE– Pin Total Input Bias Current
vs Temperature
Shutdown Current
vs Input Voltage
Oscillator Frequency
vs Temperature
30
25
20
15
10
5
800
700
600
500
400
300
200
100
0
600
550
500
450
FREQ = INTV
CC
V
> INTV + 0.5V
CC
OUT
400
350
300
FREQ = GND
25 50
V
< INTV – 0.5V
CC
OUT
0
0
–100
5
10 15 20 25 30 35 40 45 50 55 60 65
–50 –25
75 100 125 150
75 125 150
100
–75
25 50
–75 –50 –25
0
INPUT VOLTAGE (V)
TEMPERATURE (°C)
TEMPERATURE (°C)
38901 G23
38901 G22
38901 G24
Undervoltage Lockout Threshold
vs Temperature
Oscillator Frequency
vs Input Voltage
Shutdown Current vs Temperature
22
20
18
4.2
4.1
4.0
3.9
3.8
3.7
3.6
356
354
352
350
348
V
= 12V
FREQ = GND
IN
RISING
16
14
12
10
8
FALLING
346
344
–75
–25
0
25 50 75 100 125 150
–75 –50 –25
0
25 50 75 100 125 150
–50
5
10 15 20 25 30 35 40 45 50 55 60 65
TEMPERATURE (°C)
TEMPERATURE (°C)
INPUT VOLTAGE (V)
38901 G27
38901 G26
38901 G25
38901fb
7
LTC3890-1
PIN FUNCTIONS
ITH1, ITH2 (Pin 1, Pin 13): Error Amplifier Outputs and
Switching Regulator Compensation Points. Each associ-
ated channel’s current comparator trip point increases
with this control voltage.
both controllers, determines how the LTC3890-1 operates
atlightloads.PullingthispintogroundselectsBurstMode
operation.Aninternal100kresistortogroundalsoinvokes
Burst Mode Operation when the pin is floated. Tying this
pintoINTV forcescontinuousinductorcurrentoperation.
CC
V
, V (Pin 2, Pin 12): Receives the remotely sensed
FB1 FB2
Tying this pin to a voltage greater than 1.2V and less than
feedback voltage for each controller from an external
INTV – 1.3V selects pulse-skipping operation.
CC
resistive divider across the output.
SGND (Pin 7): Small-signal ground common to both
+
+
SENSE1 , SENSE2 (Pin 3, Pin 11): The (+) input to the
differential current comparators are normally connected
to DCR sensing networks or current sensing resistors.
The ITH pin voltage and controlled offsets between the
controllers, must be routed separately from high current
groundstothecommon(–)terminalsoftheC capacitors.
IN
RUN1, RUN2 (Pin 8, Pin 9): Digital Run Control Inputs
for Each Controller. Forcing RUN1 below 1.16V or RUN2
below 1.20V shuts down that controller. Forcing both of
these pins below 0.7V shuts down the entire LTC3890-1,
reducing quiescent current to approximately 14µA.
–
+
SENSE and SENSE pins in conjunction with R
the current trip threshold.
set
SENSE
–
–
SENSE1 , SENSE2 (Pin 4, Pin 10): The (–) Input to
the Differential Current Comparators. When greater than
–
INTV – 0.5V, the SENSE pin supplies current to the
INTV (Pin19):OutputoftheInternalLinearLowDropout
CC
CC
current comparator.
Regulator.Thedriverandcontrolcircuitsarepoweredfrom
this voltage source. Must be decoupled to power ground
with a minimum of 4.7µF ceramic or other low ESR ca-
FREQ (Pin 5): The Frequency Control Pin for the Internal
VCO. Connecting the pin to GND forces the VCO to a fixed
pacitor. Do not use the INTV pin for any other purpose.
CC
low frequency of 350kHz. Connecting the pin to INTV
CC
forces the VCO to a fixed high frequency of 535kHz.
Other frequencies between 50kHz and 900kHz can be
programmed using a resistor between FREQ and GND.
An internal 20µA pull-up current develops the voltage to
be used by the VCO to control the frequency.
EXTV (Pin 20): External Power Input to an Internal LDO
CC
Connected to INTV . This LDO supplies INTV power,
CC
CC
bypassing the internal LDO powered from V whenever
IN
EXTV is higher than 4.7V. See EXTV Connection in the
CC
CC
Applications Information section. Do not float or exceed
14V on this pin.
PLLIN/MODE (Pin 6): External Synchronization Input to
PhaseDetectorandForcedContinuousModeInput. When
an external clock is applied to this pin, the phase-locked
loop will force the rising TG1 signal to be synchronized
with the rising edge of the external clock. When not syn-
chronizing to an external clock, this input, which acts on
PGND (Pin 21): Driver Power Ground. Connects to the
sources of bottom (synchronous) N-channel MOSFETs
and the (–) terminal(s) of C .
IN
V (Pin 22): Main Supply Pin. A bypass capacitor should
IN
be tied between this pin and the signal ground pin.
38901fb
8
LTC3890-1
PIN FUNCTIONS
BG1, BG2 (Pin 23, Pin 18): High Current Gate Drives
PGOOD1 (Pin 27): Open-Drain Logic Output. PGOOD1 is
for Bottom (Synchronous) N-Channel MOSFETs. Voltage
pulled to ground when the voltage on the V pin is not
FB1
swing at these pins is from ground to INTV .
within 10% of its set point.
CC
BOOST1,BOOST2(Pin24,Pin17):BootstrappedSupplies
to the Topside Floating Drivers. Capacitors are connected
betweentheBOOSTandSWpinsandSchottkydiodesare
tied between the BOOST and INTVCC pins. Voltage swing
at the BOOST pins is from INTVCC to (VIN + INTVCC).
TRACK/SS1, TRACK/SS2(Pin28, Pin14):ExternalTrack-
ing and Soft-Start Input. The LTC3890-1 regulates the
V
FB1,2
voltage to the smaller of 0.8V or the voltage on the
TRACK/SS1,2 pin. An internal 1µA pull-up current source
is connected to this pin. A capacitor to ground at this
pin sets the ramp time to final regulated output voltage.
Alternatively, a resistor divider on another voltage supply
connected to this pin allows theLTC3890-1outputto track
the other supply during start-up.
SW1, SW2 (Pin 25, Pin 16): Switch Node Connections
to Inductors.
TG1, TG2 (Pin 26, Pin 15): High Current Gate Drives for
Top N-Channel MOSFETs. These are the outputs of float-
ing drivers with a voltage swing equal to INTV – 0.5V
CC
superimposed on the switch node voltage SW.
38901fb
9
LTC3890-1
FUNCTIONAL DIAGRAM
INTV
V
IN
CC
DUPLICATE FOR SECOND
CONTROLLER CHANNEL
BOOST
D
B
C
B
TG
DROP
OUT
DET
TOP
BOT
+
C
PGOOD1
0.88V
IN
D
BOT
–
SW
TOP ON
V
S
R
Q
FB1
+
INTV
CC
Q
–
SWITCH
LOGIC
0.72V
BG
SHDN
C
OUT
PGND
20µA
FREQ
V
OUT
VCO
CLK2
CLK1
+
–
R
SENSE
0.425V
SLEEP
L
ICMP
IR
–
+
+
–
PFD
C
LP
+
+
–
–
+
3mV
SENSE
SENSE
SYNC
DET
2.7V
0.65V
PLLIN/MODE
–
100k
SLOPE COMP
V
FB
R
B
+
V
IN
0.80V
TRACK/SS
EA
–
R
A
EXTV
CC
+
–
OV
C
C
0.88V
ITH
5.1V
LDO
EN
5.1V
LDO
EN
7µA (RUN1)
0.5µA (RUN2)
SHDN
RST
FB
C
R
C
C2
FOLDBACK
1µA
TRACK/SS
+
–
2(V
)
4.7V
11V
C
SHDN
SS
SGND
INTV
RUN
CC
38901 FD
38901fb
10
LTC3890-1
OPERATION (Refer to the Functional Diagram)
Main Control Loop
Shutdown and Start-Up (RUN1, RUN2 and
TRACK/ SS1, TRACK/SS2 Pins)
The LTC3890-1 uses a constant frequency, current mode
step-down architecture with the two controller channels
operating 180 degrees out-of-phase. During normal op-
eration, each external top MOSFET is turned on when the
clock for that channel sets the RS latch, and is turned off
when the main current comparator, ICMP, resets the RS
latch. The peak inductor current at which ICMP trips and
resets the latch is controlled by the voltage on the ITH pin,
which is the output of the error amplifier, EA. The error
amplifier compares the output voltage feedback signal at
The two channels of the LTC3890-1 can be independently
shutdownusingtheRUN1andRUN2pins.Pullingeitherof
these pins below 1.15V shuts down the main control loop
for that controller. Pulling both pins below 0.7V disables
both controllers and most internal circuits, including the
INTV LDOs. In this state, the LTC3890-1 draws only
CC
14µA of quiescent current.
Releasing either RUN pin allows a small internal current to
pull up the pin to enable that controller. The RUN1 pin has a
7µApull-upcurrentwhiletheRUN2pinhasasmaller0.5µA.
The 7µA current on RUN1 is designed to be large enough
so that the RUN1 pin can be safely floated (to always en-
able the controller) without worry of condensation or other
small board leakage pulling the pin down. This is ideal for
always-on applications where one or both controllers are
enabled continuously and never shut down.
the V pin, (which is generated with an external resistor
FB
divider connected across the output voltage, V , to
OUT
ground)totheinternal0.800Vreferencevoltage.Whenthe
load current increases, it causes a slight decrease in V
FB
relative to the reference, which causes the EA to increase
the ITH voltage until the average inductor current matches
the new load current.
After the top MOSFET is turned off each cycle, the bottom
MOSFETisturnedonuntileithertheinductorcurrentstarts
to reverse, as indicated by the current comparator IR, or
the beginning of the next clock cycle.
The RUN pin may be externally pulled up or driven directly
by logic. When driving the RUN pin with a low impedance
source, do not exceed the absolute maximum rating of
8V. The RUN pin has an internal 11V voltage clamp that
allows the RUN pin to be connected through a resistor to a
INTV /EXTV Power
CC
CC
highervoltage(forexample,V ),solongasthemaximum
IN
Power for the top and bottom MOSFET drivers and most
current into the RUN pin does not exceed 100µA.
other internal circuitry is derived from the INTV pin.
CC
The start-up of each controller’s output voltage V
is
When the EXTV pin is tied to a voltage less than 4.7V,
OUT
CC
controlled by the voltage on the TRACK/SS pin for that
channel. When the voltage on the TRACK/SS pin is less
than the 0.8V internal reference, the LTC3890-1 regulates
the V LDO (low dropout linear regulator) supplies 5.1V
IN
from V to INTV . If EXTV is taken above 4.7V, the V
IN
CC
CC
IN
LDO is turned off and an EXTV LDO is turned on. Once
CC
the V voltage to the TRACK/SS pin voltage instead of the
enabled, the EXTV LDO supplies 5.1V from EXTV to
FB
CC
CC
0.8V reference. This allows the TRACK/SS pin to be used
toprogramasoft-startbyconnectinganexternalcapacitor
from the TRACK/SS pin to SGND. An internal 1µA pull-up
current charges this capacitor creating a voltage ramp on
the TRACK/SS pin. As the TRACK/SS voltage rises linearly
from 0V to 0.8V (and beyond up to 5V), the output voltage
INTV . Using the EXTV pin allows the INTV power
CC
CC
CC
to be derived from a high efficiency external source such
as one of the LTC3890-1 switching regulator outputs.
Each top MOSFET driver is biased from the floating boot-
strap capacitor C , which normally recharges during each
B
cycle through an external diode when the top MOSFET
V
OUT
rises smoothly from zero to its final value.
turns off. If the input voltage, V , decreases to a voltage
IN
Alternatively the TRACK/SS pin can be used to cause the
start-up of V to track that of another supply. Typically,
close to V , the loop may enter dropout and attempt
OUT
to turn on the top MOSFET continuously. The dropout
detector detects this and forces the top MOSFET off for
about one-twelfth of the clock period every tenth cycle to
OUT
this requires connecting to the TRACK/SS pin an external
resistor divider from the other supply to ground (see the
Applications Information section).
allow C to recharge.
B
38901fb
11
LTC3890-1
OPERATION (Refer to the Functional Diagram)
Light Load Current Operation (Burst Mode Operation,
Pulse-Skipping, or Forced Continuous Mode)
(PLLIN/MODE Pin)
just before the inductor current reaches zero, preventing
it from reversing and going negative. Thus, the controller
operates in discontinuous operation.
The LTC3890-1 can be enabled to enter high efficiency
BurstModeoperation, constantfrequency pulse-skipping
mode, or forced continuous conduction mode at low load
currents. To select Burst Mode operation, tie the PLLIN/
MODE pin to a DC voltage below 0.8V (e.g., SGND). To
select forced continuous operation, tie the PLLIN/MODE
In forced continuous operation or clocked by an external
clock source to use the phase-locked loop (see Frequency
Selection and Phase-Locked Loop section), the inductor
current is allowed to reverse at light loads or under large
transient conditions. The peak inductor current is deter-
mined by the voltage on the ITH pin, just as in normal
operation. In this mode, the efficiency at light loads is
lower than in Burst Mode operation. However, continuous
operation has the advantage of lower output voltage ripple
and less interference to audio circuitry. In forced continu-
ousmode,theoutputrippleisindependentofloadcurrent.
pin to INTV . To select pulse-skipping mode, tie the
CC
PLLIN/MODE pin to a DC voltage greater than 1.2V and
less than INTV – 1.3V.
CC
WhenacontrollerisenabledforBurstModeoperation, the
minimum peak current in the inductor is set to approxi-
mately 25% of the maximum sense voltage even though
the voltage on the ITH pin indicates a lower value. If the
average inductor current is higher than the load current,
the error amplifier, EA, will decrease the voltage on the
ITH pin. When the ITH voltage drops below 0.425V, the
internal sleep signal goes high (enabling sleep mode)
and both external MOSFETs are turned off. The ITH pin is
then disconnected from the output of the EA and parked
at 0.450V.
WhenthePLLIN/MODEpinisconnectedforpulse-skipping
mode, the LTC3890-1 operates in PWM pulse-skipping
mode at light loads. In this mode, constant frequency
operation is maintained down to approximately 1% of
designedmaximumoutputcurrent. Atverylightloads, the
current comparator, ICMP, may remain tripped for several
cycles and force the external top MOSFET to stay off for
the same number of cycles (i.e., skipping pulses). The
inductor current is not allowed to reverse (discontinuous
operation). This mode, like forced continuous operation,
exhibits low output ripple as well as low audio noise and
reduced RF interference as compared to Burst Mode
operation. It provides higher low current efficiency than
forced continuous mode, but not nearly as high as Burst
Mode operation.
In sleep mode, much of the internal circuitry is turned off,
reducing the quiescent current that the LTC3890-1 draws.
If one channel is shut down and the other channel is in
sleep mode, the LTC3890-1 draws only 50µA of quiescent
current. Ifbothchannelsareinsleepmode, theLTC3890-1
draws only 60µA of quiescent current. In sleep mode,
the load current is supplied by the output capacitor. As
the output voltage decreases, the EA’s output begins to
rise. When the output voltage drops enough, the ITH pin
is reconnected to the output of the EA, the sleep signal
goes low, and the controller resumes normal operation
by turning on the top external MOSFET on the next cycle
of the internal oscillator.
Frequency Selection and Phase-Locked Loop
(FREQ and PLLIN/MODE Pins)
Theselectionofswitchingfrequencyisatrade-offbetween
efficiency and component size. Low frequency opera-
tion increases efficiency by reducing MOSFET switching
losses, but requires larger inductance and/or capacitance
to maintain low output ripple voltage.
WhenacontrollerisenabledforBurstModeoperation, the
inductorcurrentisnotallowedtoreverse. Thereversecur-
rentcomparator,IR,turnsoffthebottomexternalMOSFET
The switching frequency of the LTC3890-1’s controllers
can be selected using the FREQ pin.
38901fb
12
LTC3890-1
OPERATION (Refer to the Functional Diagram)
If the PLLIN/MODE pin is not being driven by an external
clock source, the FREQ pin can be tied to SGND, tied to
INTVCC orprogrammedthroughanexternalresistor.Tying
FREQtoSGNDselects350kHzwhiletyingFREQtoINTVCC
selects 535kHz. Placing a resistor between FREQ and
SGND allows the frequency to be programmed between
50kHz and 900kHz, as shown in Figure 10.
MOSFET is turned off and the bottom MOSFET is turned
on until the overvoltage condition is cleared.
Power Good (PGOOD1 Pin)
ThePGOOD1pinisconnectedtoanopendrainofaninternal
N-channel MOSFET. The MOSFET turns on and pulls the
PGOOD1 pin low when the corresponding V pin volt-
FB1
age is not within 10% of the 0.8V reference voltage. The
A phase-locked loop (PLL) is available on the LTC3890-1
to synchronize the internal oscillator to an external clock
source that is connected to the PLLIN/MODE pin. The
LTC3890-1’s phase detector adjusts the voltage (through
an internal lowpass filter) of the VCO input to align the
turn-on of controller 1’s external top MOSFET to the ris-
ing edge of the synchronizing signal. Thus, the turn-on
of controller 2’s external top MOSFET is 180 degrees out
of phase to the rising edge of the external clock source.
PGOOD1 pin is also pulled low when the corresponding
RUN1 pin is low (shut down). When the V pin voltage
FB1
is within the 10% requirement, the MOSFET is turned
off and the pin is allowed to be pulled up by an external
resistor to a source no greater than 6V.
Foldback Current
When the output voltage falls to less than 70% of its
nominal level, foldback current limiting is activated, pro-
gressively lowering the peak current limit in proportion to
the severity of the overcurrent or short-circuit condition.
Foldback current limiting is disabled during the soft-start
The VCO input voltage is prebiased to the operating
frequency set by the FREQ pin before the external clock
is applied. A resistor connected between the FREQ pin
and SGND can prebias VCO’s input voltage to the desired
frequency. If prebiased near the external clock frequency,
the PLL loop only needs to make slight changes to the
VCO input in order to synchronize the rising edge of the
external clock’s to the rising edge of TG1. The ability to
prebias the loop filter allows the PLL to lock-in rapidly
without deviating far from the desired frequency.
interval (as long as the V voltage is keeping up with the
FB
TRACK/SS voltage).
Theory and Benefits of 2-Phase Operation
Why the need for 2-phase operation? Up until the 2-phase
family, constant-frequency dual switching regulators
operated both channels in phase (i.e., single phase
operation). This means that both switches turned on at
the same time, causing current pulses of up to twice the
amplitude of those for one regulator to be drawn from the
input capacitor and battery. These large amplitude current
pulses increased the total RMS current flowing from the
input capacitor, requiring the use of more expensive input
capacitorsandincreasingbothEMIandlossesintheinput
capacitor and battery.
The typical capture range of the phase-locked loop is
from approximately 55kHz to 1MHz, with a guarantee
to be between 75kHz and 850kHz. In other words, the
LTC3890-1’sPLL isguaranteed to lock to anexternal clock
source whose frequency is between 75kHz and 850kHz.
The typical input clock thresholds on the PLLIN/MODE
pin are 1.6V (rising) and 1.2V (falling).
Output Overvoltage Protection
With 2-phase operation, the two channels of the dual
switchingregulatorareoperated180degreesout-of-phase.
Thiseffectivelyinterleavesthecurrentpulsesdrawnbythe
switches,greatlyreducingtheoverlaptimewheretheyadd
An overvoltage comparator guards against transient over-
shoots as well as other more serious conditions that may
overvoltage the output. When the V pin rises by more
FB
than 10% above its regulation point of 0.800V, the top
38901fb
13
LTC3890-1
OPERATION (Refer to the Functional Diagram)
together. The result is a significant reduction in total RMS
input current, which in turn allows less expensive input
capacitors to be used, reduces shielding requirements for
EMI and improves real world operating efficiency.
the RMS input current varies for single phase and 2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.
It can readily be seen that the advantages of 2-phase op-
eration are not just limited to a narrow operating range,
for most applications is that 2-phase operation will reduce
the input capacitor requirement to that for just one chan-
nel operating at maximum current and 50% duty cycle.
Figure1comparestheinputwaveformsforarepresentative
single-phase dual switching regulator to the LTC3890-1
2-phase dual switching regulator. An actual measure-
ment of the RMS input current under these conditions
shows that 2-phase operation dropped the input current
3.0
from 2.53A
to 1.55A
. While this is an impressive
RMS
RMS
SINGLE PHASE
reduction in itself, remember that the power losses are
DUAL CONTROLLER
2.5
2.0
1.5
1.0
0.5
0
2
proportionaltoI
,meaningthattheactualpowerwasted
RMS
is reduced by a factor of 2.66. The reduced input ripple
voltage also means less power is lost in the input power
path, which could include batteries, switches, trace/con-
nectorresistancesandprotectioncircuitry.Improvements
in both conducted and radiated EMI also directly accrue
as a result of the reduced RMS input current and voltage.
2-PHASE
DUAL CONTROLLER
V
O1
V
O2
= 5V/3A
= 3.3V/3A
Of course, the improvement afforded by 2-phase opera-
tion is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
0
10
20
30
40
INPUT VOLTAGE (V)
38901 F02
Figure 2. RMS Input Current Comparison
voltage V (Duty Cycle = V /V ). Figure 2 shows how
IN
OUT IN
5V SWITCH
20V/DIV
3.3V SWITCH
20V/DIV
INPUT CURRENT
5A/DIV
INPUT VOLTAGE
500mV/DIV
38901 F01
I
= 2.53A
I = 1.55A
IN(MEAS) RMS
IN(MEAS)
RMS
Figure 1. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators
Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the 2-Phase Regulator Allows
Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency
38901fb
14
LTC3890-1
APPLICATIONS INFORMATION
The Typical Application on the first page is a basic
LTC3890-1applicationcircuit.LTC3890-1canbeconfigured
to use either DCR (inductor resistance) sensing or low
valueresistorsensing.Thechoicebetweenthetwocurrent
sensing schemes is largely a design trade-off between
cost, power consumption, and accuracy. DCR sensing
is becoming popular because it saves expensive current
sensing resistors and is more power efficient, especially
in high current applications. However, current sensing
resistors provide the most accurate current limits for the
controller. Other external component selection is driven
by the load requirement, and begins with the selection of
programmed current limit unpredictable. If inductor DCR
sensing is used (Figure 4b), sense resistor R1 should be
placed close to the switching node, to prevent noise from
coupling into sensitive small-signal nodes.
TO SENSE FILTER,
NEXT TO THE CONTROLLER
C
OUT
38901 F03
INDUCTOR OR R
SENSE
Figure 3. Sense Lines Placement with Inductor or Sense Resistor
R
(if R
is used) and inductor value. Next, the
SENSE
SENSE
V
powerMOSFETsandSchottkydiodesareselected. Finally,
input and output capacitors are selected.
V
IN
IN
INTV
CC
BOOST
TG
+
–
SENSE and SENSE Pins
R
SENSE
SW
V
OUT
+
–
The SENSE and SENSE pins are the inputs to the current
comparators. The common mode voltage range on these
pins is 0V to 28V (abs max), enabling the LTC3890-1 to
regulate output voltages up to a nominal 24V (allowing
margin for tolerances and transients).
LTC3890-1
BG
R1*
+
SENSE
PLACE CAPACITOR NEAR
SENSE PINS
C1*
–
SENSE
SGND
+
The SENSE pin is high impedance over the full common
*R1 AND C1 ARE OPTIONAL.
38901 F04a
mode range, drawing at most 1µA. This high impedance
allows the current comparators to be used in inductor
DCR sensing.
(4a) Using a Resistor to Sense Current
V
INTV
V
IN
IN
–
The impedance of the SENSE pin changes depending on
CC
–
the common mode voltage. When SENSE is less than
INDUCTOR
DCR
BOOST
TG
INTV – 0.5V, a small current of less than 1µA flows out
CC
L
–
of the pin. When SENSE is above INTV + 0.5V, a higher
SW
V
OUT
CC
LTC3890-1
current (~700µA) flows into the pin. Between INTV
–
CC
BG
0.5V and INTV + 0.5V, the current transitions from the
CC
R1
C1* R2
+
smaller current to the higher current.
SENSE
Filter components mutual to the sense lines should be
placed close to the LTC3890-1, and the sense lines should
run close together to a Kelvin connection underneath the
current sense element (shown in Figure 3). Sensing cur-
rent elsewhere can effectively add parasitic inductance
and capacitance to the current sense element, degrading
the information at the sense terminals and making the
–
SENSE
SGND
38901 F04b
R2
R1 + R2
L
DCR
||
(R1 R2) • C1 =
*PLACE C1 NEAR
SENSE PINS
R
= DCR
SENSE(EQ)
(4b) Using the Inductor DCR to Sense Current
Figure 4. Current Sensing Methods
38901fb
15
LTC3890-1
APPLICATIONS INFORMATION
Low Value Resistor Current Sensing
the DCR is greater than the target sense resistor value.
To properly dimension the external filter components, the
DCR of the inductor must be known. It can be measured
using a good RLC meter, but the DCR tolerance is not
always the same and varies with temperature; consult
the manufacturers’ data sheets for detailed information.
A typical sensing circuit using a discrete resistor is shown
in Figure 4a. R
output current.
is chosen based on the required
SENSE
The current comparator has a maximum threshold
. The current comparator threshold voltage
V
SENSE(MAX)
Using the inductor ripple current value from the Inductor
Value Calculation section, the target sense resistor value is:
sets the peak of the inductor current, yielding a maximum
average output current, I
, equal to the peak value less
MAX
VSENSE(MAX)
half the peak-to-peak ripple current, ∆I . To calculate the
L
RSENSE(EQUIV)
=
sense resistor value, use the equation:
∆IL
IMAX
+
2
VSENSE(MAX)
RSENSE
=
∆IL
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimumvaluefortheMaximumCurrentSenseThreshold
IMAX
+
2
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimumvaluefortheMaximumCurrentSenseThreshold
(V
).
SENSE(MAX)
Next, determine the DCR of the inductor. When provided,
use the manufacturer’s maximum value, usually given at
20°C. Increase this value to account for the temperature
coefficient of copper resistance, which is approximately
(V
).
SENSE(MAX)
When using the controller in very low dropout conditions,
the maximum output current level will be reduced due to
the internal compensation required to meet stability cri-
terion for buck regulators operating at greater than 50%
duty factor. A curve is provided in the Typical Performance
Characteristics section to estimate this reduction in peak
inductorcurrentdependingupontheoperatingdutyfactor.
0.4%/°C. A conservative value for T
is 100°C.
L(MAX)
To scale the maximum inductor DCR to the desired sense
resistor value (R ), use the divider ratio:
D
RSENSE(EQUIV)
RD =
DCRMAX atT
L(MAX)
Inductor DCR Sensing
C1 is usually selected to be in the range of 0.1µF to 0.47µF.
ThisforcesR1||R2toaround2k, reducingerrorthatmight
For applications requiring the highest possible efficiency
at high load currents, the LTC3890-1 is capable of sensing
the voltage drop across the inductor DCR, as shown in
Figure 4b. The DCR of the inductor represents the small
amount of DC resistance of the copper wire, which can be
lessthan1mΩfortoday’slowvalue,highcurrentinductors.
In a high current application requiring such an inductor,
power loss through a sense resistor would cost several
points of efficiency compared to inductor DCR sensing.
+
have been caused by the SENSE pin’s 1µA current.
The equivalent resistance R1|| R2 is scaled to the room
temperature inductance and maximum DCR:
L
R1||R2 =
DCR at 20°C • C1
(
)
The sense resistor values are:
If the external (R1||R2) • C1 time constant is chosen to be
exactly equal to the L/DCR time constant, the voltage drop
across the external capacitor is equal to the drop across
theinductorDCRmultipliedbyR2/(R1+R2).R2scalesthe
voltage across the sense terminals for applications where
R1||R2
RD
R1•RD
1– RD
R1=
; R2 =
38901fb
16
LTC3890-1
APPLICATIONS INFORMATION
The maximum power loss in R1 is related to duty cycle,
and will occur in continuous mode at the maximum input
voltage:
25% of the current limit determined by R
. Lower
SENSE
inductor values (higher ∆I ) will cause this to occur at
L
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
V
IN(MAX) – VOUT • V
(
)
OUT
PLOSS R1=
R1
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
sense resistors. Light load power loss can be modestly
higher with a DCR network than with a sense resistor, due
totheextraswitchinglossesincurredthroughR1.However,
DCR sensing eliminates a sense resistor, reduces conduc-
tion losses and provides higher efficiency at heavy loads.
Peak efficiency is about the same with either method.
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
affordthecorelossfoundinlowcostpowderedironcores,
forcingtheuseofmoreexpensiveferriteormolypermalloy
cores. Actual core loss is independent of core size for a
fixedinductorvalue,butitisverydependentoninductance
value selected. As inductance increases, core losses go
down. Unfortunately, increased inductance requires more
turns of wire and therefore copper losses will increase.
Inductor Value Calculation
Ferrite designs have very low core loss and are preferred
for high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates hard, which means that induc-
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFET switching and gate charge losses. In addition to
this basic trade-off, the effect of inductor value on ripple
currentandlowcurrentoperationmustalsobeconsidered.
Power MOSFET and Schottky Diode
(Optional) Selection
The inductor value has a direct effect on ripple current. The
inductor ripple current, ∆I , decreases with higher induc-
L
Two external power MOSFETs must be selected for each
controller in the LTC3890-1: one N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
tance or higher frequency and increases with higher V :
IN
VOUT
1
∆IL =
V
1–
OUT
f L
( ) ( )
V
IN
Thepeak-to-peakdrivelevelsaresetbytheINTV voltage.
CC
Accepting larger values of ∆I allows the use of low in-
This voltage is typically 5.1V during start-up (see EXTV
L
CC
ductances, but results in higher output voltage ripple and
Pin Connection). Consequently, logic-level threshold
greater core losses. A reasonable starting point for setting
MOSFETs must be used in most applications. Pay close
ripple current is ∆I =0.3(I
). The maximum ∆I occurs
L
attentiontotheBV specificationfortheMOSFETsaswell.
L
MAX
DSS
at the maximum input voltage.
Selection criteria for the power MOSFETs include the
The inductor value also has secondary effects. The tran-
sition to Burst Mode operation begins when the average
inductor current required results in a peak current below
on-resistance, R
, Miller capacitance, C
, input
DS(ON)
MILLER
voltage and maximum output current. Miller capacitance,
C
, can be approximated from the gate charge curve
MILLER
38901fb
17
LTC3890-1
APPLICATIONS INFORMATION
usually provided on the MOSFET manufacturers’ data
The term (1+ δ) is generally given for a MOSFET in the
sheet. C
is equal to the increase in gate charge
form of a normalized R
vs Temperature curve, but
MILLER
DS(ON)
along the horizontal axis while the curve is approximately
flat divided by the specified change in V . This result is
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
DS
then multiplied by the ratio of the application applied V
DS
The optional Schottky diodes D3 and D4 shown in
Figure 11 conduct during the dead-time between the
conduction of the two power MOSFETs. This prevents
the body diode of the bottom MOSFET from turning on,
storing charge during the dead-time and requiring a
reverse recovery period that could cost as much as 3%
to the Gate charge curve specified V . When the IC is
DS
operating in continuous mode the duty cycles for the top
and bottom MOSFETs are given by:
VOUT
Main Switch Duty Cycle =
V
IN
in efficiency at high V . A 1A to 3A Schottky is generally
IN
V − V
IN
OUT
a good compromise for both regions of operation due
to the relatively small average current. Larger diodes
result in additional transition losses due to their larger
junction capacitance.
Synchronous Switch Duty Cycle =
V
IN
The MOSFET power dissipations at maximum output
current are given by:
C and C
Selection
IN
OUT
VOUT
2
PMAIN
=
=
I
1+ δ R
+
)
(
MAX) (
)
DS(ON)
The selection of C is simplified by the 2-phase architec-
V
IN
IN
ture and its impact on the worst-case RMS current drawn
throughtheinputnetwork(battery/fuse/capacitor).Itcanbe
shown that the worst-case capacitor RMS current occurs
when only one controller is operating. The controller with
2
IMAX
2
V
R
(
C
)
(
•
(
)
IN
DR
MILLER
1
1
+
f
( )
the highest (V )(I ) product needs to be used in the
OUT OUT
VINTVCC – VTHMIN
V
THMIN
formula shown in Equation 1 to determine the maximum
RMS capacitor current requirement. Increasing the out-
put current drawn from the other controller will actually
decrease the input RMS ripple current from its maximum
value. The out-of-phase technique typically reduces the
input capacitor’s RMS ripple current by a factor of 30%
to 70% when compared to a single phase power supply
solution.
V – V
2
IN
OUT
P
I
1+ δ R
(
MAX) (
)
SYNC
DS(ON)
V
IN
where δ is the temperature dependency of R
and
DS(ON)
R
(approximately 2Ω) is the effective driver resistance
DR
at the MOSFET’s Miller threshold voltage. V
typical MOSFET minimum threshold voltage.
is the
THMIN
Incontinuousmode,thesourcecurrentofthetopMOSFET
is a square wave of duty cycle (V )/(V ). To prevent
large voltage transients, a low ESR capacitor sized for the
maximum RMS current of one channel must be used. The
maximum RMS capacitor current is given by:
2
BothMOSFETshaveI RlosseswhilethetopsideN-channel
equation includes an additional term for transition losses,
OUT
IN
which are highest at high input voltages. For V < 20V
IN
the high current efficiency generally improves with larger
MOSFETs, while for V > 20V the transition losses rapidly
IN
IMAX
increasetothepointthattheuseofahigherR
device
1/2
DS(ON)
(1)
CIN Required IRMS
≈
V
V – V
IN
OUT
(
)
(
)
OUT
withlowerC
actuallyprovideshigherefficiency.The
MILLER
VIN
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short-circuit when the synchronous switch is on close
to 100% of the period.
This formula has a maximum at V = 2V , where I
IN
OUT
RMS
= I /2. This simple worst-case condition is commonly
OUT
usedfordesignbecauseevensignificantdeviationsdonot
38901fb
18
LTC3890-1
APPLICATIONS INFORMATION
offermuchrelief.Notethatcapacitormanufacturers’ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet
size or height requirements in the design. Due to the high
operatingfrequencyoftheLTC3890-1, ceramiccapacitors
The output ripple is highest at maximum input voltage
since ∆I increases with input voltage.
L
Setting Output Voltage
The LTC3890-1 output voltages are each set by an exter-
nal feedback resistor divider carefully placed across the
output, as shown in Figure 5. The regulated output voltage
is determined by:
can also be used for C . Always consult the manufacturer
IN
if there is any question.
RB
RA
The benefit of the LTC3890-1 2-phase operation can be
calculatedbyusingEquation1forthehigherpowercontrol-
ler and then calculating the loss that would have resulted
if both controller channels switched on at the same time.
The total RMS power lost is lower when both controllers
are operating due to the reduced overlap of current pulses
required through the input capacitor’s ESR. This is why
the input capacitor’s requirement calculated above for the
worst-case controller is adequate for the dual controller
design. Also, the input protection fuse resistance, battery
resistance, and PC board trace resistance losses are also
reduced due to the reduced peak currents in a 2-phase
system. The overall benefit of a multiphase design will
only be fully realized when the source impedance of the
power supply/battery is included in the efficiency testing.
The drains of the top MOSFETs should be placed within
VOUT = 0.8V 1+
To improve the frequency response, a feedforward ca-
pacitor, C , may be used. Great care should be taken to
FF
route the V line away from noise sources, such as the
FB
inductor or the SW line.
V
OUT
R
B
C
FF
1/2 LTC3890-1
V
FB
R
A
38901 F05
Figure 5. Setting Output Voltage
Tracking and Soft-Start (TRACK/SS Pins)
1cmofeachotherandshareacommonC (s). Separating
IN
The start-up of each V
is controlled by the voltage on
OUT
the drains and C may produce undesirable voltage and
IN
the respective TRACK/SS pin. When the voltage on the
TRACK/SS pin is less than the internal 0.8V reference, the
current resonances at V .
IN
A small (0.1µF to 1µF) bypass capacitor between the chip
LTC3890-1 regulates the V pin voltage to the voltage on
FB
V
pin and ground, placed close to the LTC3890-1, is
IN
the TRACK/SS pin instead of 0.8V. The TRACK/SS pin can
also suggested. A 10Ω resistor placed between C (C1)
IN
be used to program an external soft-start function or to
and the V pin provides further isolation between the
two channels.
IN
allow V
to track another supply during start-up.
OUT
Soft-start is enabled by simply connecting a capacitor
from the TRACK/SS pin to ground, as shown in Figure 6.
An internal 1µA current source charges the capacitor,
providing a linear ramping voltage at the TRACK/SS pin.
The selection of C
is driven by the effective series
OUT
resistance (ESR). Typically, once the ESR requirement
is satisfied, the capacitance is adequate for filtering. The
output ripple (∆V ) is approximated by:
OUT
The LTC3890-1 will regulate the V pin (and hence V
)
FB
OUT
according to the voltage on the TRACK/SS pin, allowing
1
∆VOUT ≈ ∆I ESR+
L
V
OUT
to rise smoothly from 0V to its final regulated value.
8 • f • COUT
The total soft-start time will be approximately:
where f is the operating frequency, C
is the output
OUT
0.8V
1µA
tSS = CSS
•
capacitance and ∆I is the ripple current in the inductor.
L
38901fb
19
LTC3890-1
APPLICATIONS INFORMATION
1/2 LTC3890-1
TRACK/SS
V
V
X(MASTER)
C
SS
SGND
38901 F06
OUT(SLAVE)
Figure 6. Using the TRACK/SS Pin to Program Soft-Start
Alternatively, the TRACK/SS pin can be used to track two
(or more) supplies during start-up, as shown qualitatively
in Figures 7a and 7b. To do this, a resistor divider should
38901 F07a
TIME
be connected from the master supply (V ) to the TRACK/
X
(7a) Coincident Tracking
SS pin of the slave supply (V ), as shown in Figure 8.
OUT
During start-up V
will track V according to the ratio
OUT
X
set by the resistor divider:
V
V
X(MASTER)
OUT(SLAVE)
RTRACKA + RTRACKB
RA + RB
VX
RA
=
•
VOUT RTRACKA
For coincident tracking (V
= V during start-up):
X
OUT
R = R
A
TRACKA
TRACKB
R = R
B
38901 F07b
TIME
INTV Regulators
CC
(7b) Ratiometric Tracking
The LTC3890-1 features two separate internal P-channel
low dropout linear regulators (LDO) that supply power at
Figure 7. Two Different Modes of Output Voltage Tracking
theINTV pinfromeithertheV supplypinortheEXTV
CC
IN
CC
CC
pindependingontheconnectionoftheEXTV pin.INTV
CC
powers the gate drivers and much of the LTC3890-1’s
V
V
OUT
x
internalcircuitry.TheV LDOandtheEXTV LDOregulate
IN
CC
INTV to 5.1V. Each of these can supply a peak current of
CC
1/2 LTC3890-1
R
B
50mA and must be bypassed to ground with a minimum
of 4.7µF ceramic capacitor. No matter what type of bulk
capacitor is used, an additional 1µF ceramic capacitor
V
FB
R
A
R
R
TRACKB
placed directly adjacent to the INTV and PGND IC pins is
TRACK/SS
CC
38901 F08
highlyrecommended.Goodbypassingisneededtosupply
the high transient currents required by the MOSFET gate
drivers and to prevent interaction between the channels.
TRACKA
Figure 8. Using the TRACK/SS Pin for Tracking
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maxi-
mum junction temperature rating for the LTC3890-1 to be
exceeded. The INTV current, which is dominated by the
CC
gate charge current, may be supplied by either the V
IN
LDO or the EXTV LDO. When the voltage on the EXTV
CC
CC
38901fb
20
LTC3890-1
APPLICATIONS INFORMATION
pin is less than 4.7V, the V LDO is enabled. Power dis-
an 8.5V supply reduces the junction temperature in the
previous example from 125°C to:
IN
sipation for the IC in this case is highest and is equal to
V
• I
. The gate charge current is dependent on
IN
INTVCC
T = 70°C + (15mA)(8.5V)(90°C/W) = 82°C
J
operating frequency as discussed in the Efficiency Con-
siderations section. The junction temperature can be
estimated by using the equations given in Note 3 of the
Electrical Characteristics. For example, the LTC3890-1
However,for3.3Vandotherlowvoltageoutputs,additional
circuitryisrequiredtoderiveINTV powerfromtheoutput.
CC
The following list summarizes the four possible connec-
INTV current is limited to less than 15mA from a 40V
CC
tions for EXTV :
CC
supply when not using the EXTV supply at 70°C ambi-
CC
1. EXTV Grounded.ThiswillcauseINTV tobepowered
CC
CC
ent temperature:
fromtheinternal5.1Vregulatorresultinginanefficiency
T = 70°C + (15mA)(40V)(90°C/W) = 125°C
J
penalty of up to 10% at high input voltages.
To prevent the maximum junction temperature from be-
ing exceeded, the input supply current must be checked
while operating in forced continuous mode (PLLIN/MODE
2. EXTV Connected directly to V . This is the normal
CC
OUT
connection for a 5V to 14V regulator and provides the
highest efficiency.
= INTV ) at maximum V .
CC
IN
3. EXTV ConnectedtoanExternalsupply. Ifanexternal
CC
When the voltage applied to EXTV rises above 4.7V, the
CC
supply is available in the 5V to 14V range, it may be
V LDO is turned off and the EXTV LDO is enabled. The
IN
CC
used to power EXTV providing it is compatible with
CC
EXTV LDO remains on as long as the voltage applied to
CC
the MOSFET gate drive requirements. Ensure that
EXTV remains above 4.5V. The EXTV LDO attempts
CC
CC
EXTV < V .
CC
IN
to regulate the INTV voltage to 5.1V, so while EXTV
CC
CC
CC
CC
4. EXTV ConnectedtoanOutput-DerivedBoostNetwork.
CC
is less than 5.1V, the LDO is in dropout and the INTV
For 3.3V and other low voltage regulators, efficiency
voltage is approximately equal to EXTV . When EXTV
CC
gains can still be realized by connecting EXTV to an
CC
is greater than 5.1V, up to an absolute maximum of 14V,
output-derivedvoltagethathasbeenboostedtogreater
INTV is regulated to 5.1V.
CC
than 4.7V. This can be done with the capacitive charge
Using the EXTVCC LDO allows the MOSFET driver and
control power to be derived from one of the LTC3890-1’s
switching regulator outputs (4.7V ≤ VOUT ≤ 14V) during
normal operation and from the VIN LDO when the out-
put is out of regulation (e.g., start-up, short-circuit). If
more current is required through the EXTVCC LDO than
is specified, an external Schottky diode can be added
between the EXTVCC and INTVCC pins. In this case, do
not apply more than 6V to the EXTVCC pin and make sure
that EXTVCC ≤ VIN.
pump shown in Figure 9. Ensure that EXTV < V .
CC
IN
C
IN
BAT85
BAT85
BAT85
V
IN
MTOP
MBOT
NDS7002
TG1
1/2 LTC3890-1
L
R
SENSE
V
EXTV
SW
OUT
CC
Significant efficiency and thermal gains can be realized
C
D
BG1
OUT
by powering INTV from the output, since the V cur-
CC
IN
rent resulting from the driver and control currents will be
scaled by a factor of (Duty Cycle)/(Switcher Efficiency).
For 5V to 14V regulator outputs, this means connecting
38901 F09
PGND
Figure 9. Capacitive Charge Pump for EXTVCC
the EXTV pin directly to V . Tying the EXTV pin to
CC
OUT
CC
38901fb
21
LTC3890-1
APPLICATIONS INFORMATION
than this amount in shutdown, INTV can be loaded
Topside MOSFET Driver Supply (C , D )
CC
B
B
with an external resistor or clamped with a Zener diode.
Alternatively, the PGOOD resistor can be used to sink the
Externalbootstrapcapacitors,C ,connectedtotheBOOST
B
pinssupplythegatedrivevoltagesforthetopsideMOSFETs.
current (assuming the resistor pulls up to INTV ) since
CC
Capacitor C in the Functional Diagram is charged though
B
PGOOD is pulled low when the converter is shut down.
Nonetheless, using a low-leakage diode is the best choice
to maintain low quiescent current under all conditions.
external diode D from INTV when the SW pin is low.
B
CC
When one of the topside MOSFETs is to be turned on, the
driver places the C voltage across the gate-source of the
B
desired MOSFET. This enhances the top MOSFET switch
Fault Conditions: Current Limit and Current Foldback
and turns it on. The switch node voltage, SW, rises to V
IN
The LTC3890-1 includes current foldback to help limit
load current when the output is shorted to ground. If
the output voltage falls below 70% of its nominal output
level, then the maximum sense voltage is progressively
lowered from 100% to 45% of its maximum selected
value. Under short-circuit conditions with very low duty
cycles, the LTC3890-1 will begin cycle skipping in order to
limit the short-circuit current. In this situation the bottom
MOSFET will be dissipating most of the power but less
than in normal operation. The short-circuit ripple current
and the BOOST pin follows. With the topside MOSFET
on, the boost voltage is above the input supply: V
=
BOOST
V + V
. The value of the boost capacitor, C , needs
IN
INTVCC
B
to be 100 times that of the total input capacitance of the
topsideMOSFET(s).Thereversebreakdownoftheexternal
Schottky diode must be greater than V
.
IN(MAX)
The external diode D can be a Schottky diode or silicon
B
diode, but in either case it should have low-leakage and
fast recovery. Pay close attention to the reverse leakage
current specification for this diode, especially at high
temperatures where it generally increases substantially.
For applications with output voltages greater than ~5V
is determined by the minimum on-time. t
, of the
ON(MIN)
LTC3890-1 (≈90ns), the input voltage and inductor value:
V
L
IN
∆IL(SC) = t
ON(MIN)
that are switching infrequently, a leaky diode D can fully
B
discharge the bootstrap capacitor C , creating a current
B
path from the output voltage to the BOOST pin to INTV .
CC
The resulting average short-circuit current is:
Not only does this increase the quiescent current of the
1
2
converter, but it can cause INTV to rise to dangerous
CC
ISC = 45% •ILIM(MAX) – ∆IL(SC)
levels if the leakage exceeds the current consumption on
INTV .
CC
Fault Conditions: Overvoltage Protection (Crowbar)
Particularly, this is a concern in Burst Mode operation at
no load or very light loads, where the part is switching
The overvoltage crowbar is designed to blow a system
input fuse when the output voltage of the regulator rises
muchhigherthannominallevels.Thecrowbarcauseshuge
currents to flow, that blow the fuse to protect against a
shorted top MOSFET if the short occurs while the control-
ler is operating.
very infrequently and the current draw on INTV is very
CC
low (typically about 35µA). Generally, pulse-skipping and
forced continuous modes are less sensitive to leakage,
since the more frequent switching keeps the bootstrap
capacitor C charged, preventing a current path from the
B
output voltage to INTV .
CC
A comparator monitors the output for overvoltage condi-
tions. The comparator detects faults greater than 10%
above the nominal output voltage. When this condition
is sensed, the top MOSFET is turned off and the bottom
MOSFET is turned on until the overvoltage condition is
cleared. ThebottomMOSFETremainsoncontinuouslyfor
However, in cases where the converter has been operat-
ing (in any mode) and then is shut down, if the leakage
of diode D fully discharges the bootstrap capacitor C
B
B
before the output voltage discharges to below ~5V, then
the leakage current path can be created from the output
aslongastheovervoltageconditionpersists;ifV returns
OUT
voltage to INTV . In shutdown, the INTV pin is able to
CC
CC
to a safe level, normal operation automatically resumes.
sink about 30µA. To accommodate diode leakage greater
38901fb
22
LTC3890-1
APPLICATIONS INFORMATION
1000
900
800
700
600
500
400
300
200
100
0
AshortedtopMOSFETwillresultinahighcurrentcondition
which will open the system fuse. The switching regulator
will regulate properly with a leaky top MOSFET by altering
the duty cycle to accommodate the leakage.
Phase-Locked Loop and Frequency Synchronization
The LTC3890-1 has an internal phase-locked loop (PLL)
comprised of a phase frequency detector, a lowpass filter,
and a voltage-controlled oscillator (VCO). This allows the
turn-on of the top MOSFET of controller 1 to be locked to
the rising edge of an external clock signal applied to the
PLLIN/MODEpin.Theturn-onofcontroller2’stopMOSFET
is thus 180 degrees out of phase with the external clock.
The phase detector is an edge sensitive digital type that
provides zero degrees phase shift between the external
and internal oscillators. This type of phase detector does
not exhibit false lock to harmonics of the external clock.
15 25 35 45 55 65 75 85 95 105 115 125
FREQ PIN RESISTOR (kΩ)
38901 F10
Figure 10. Relationship Between Oscillator Frequency
and Resistor Value at the FREQ Pin
frequency be near external clock frequency, doing so will
prevent the operating frequency from passing through a
large range of frequencies as the PLL locks.
If the external clock frequency is greater than the internal
oscillator’sfrequency,f ,thencurrentissourcedcontinu-
OSC
Table 2 summarizes the different states in which the FREQ
pin can be used.
ously from the phase detector output, pulling up the VCO
input. When the external clock frequency is less than f
,
OSC
current is sunk continuously, pulling down the VCO input.
If the external and internal frequencies are the same but
exhibit a phase difference, the current sources turn on for
an amount of time corresponding to the phase difference.
The voltage at the VCO input is adjusted until the phase
and frequency of the internal and external oscillators are
identical. At the stable operating point, the phase detector
output is high impedance and the internal filter capacitor,
CLP, holds the voltage at the VCO input.
Table 2
FREQ PIN
PLLIN/MODE PIN
DC Voltage
FREQUENCY
350kHz
0V
INTV
DC Voltage
535kHz
CC
Resistor
DC Voltage
50kHz to 900kHz
Any of the Above
External Clock
Phase-Locked to
External Clock
Minimum On-Time Considerations
Minimum on-time, t , is the smallest time dura-
tion that the LTC3890-1 is capable of turning on the top
MOSFET. It is determined by internal timing delays and the
gate charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that:
Note that the LTC3890-1 can only be synchronized to an
external clock whose frequency is within range of the
LTC3890-1’s internal VCO, which is nominally 55kHz to
1MHz.Thisisguaranteedtobebetween75kHzand850kHz.
ON(MIN)
Typically,theexternalclock(onthePLLIN/MODEpin)input
highthresholdis1.6V,whiletheinputlowthresholdis1.1V.
VOUT
V f
IN ( )
RapidphaselockingcanbeachievedbyusingtheFREQpin
to set a free-running frequency near the desired synchro-
nization frequency. The VCO’s input voltage is prebiased
at a frequency corresponding to the frequency set by the
FREQ pin. Once prebiased, the PLL only needs to adjust
the frequency slightly to achieve phase lock and synchro-
nization. Although it is not required that the free-running
tON(MIN)
<
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
38901fb
23
LTC3890-1
APPLICATIONS INFORMATION
(Efficiency). For example, in a 20V to 5V application,
Theminimumon-timefortheLTC3890-1isapproximately
90ns. However, as the peak sense voltage decreases the
minimum on-time gradually increases up to about TBDns.
This is of particular concern in forced continuous applica-
tions with low ripple current at light loads. If the duty cycle
drops below the minimum on-time limit in this situation,
a significant amount of cycle skipping can occur with cor-
respondingly larger current and voltage ripple.
10mAofINTV currentresultsinapproximately2.5mA
CC
of V current. This reduces the midcurrent loss from
IN
10% or more (if the driver was powered directly from
V ) to only a few percent.
IN
2
3. I R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resis-
tor, and input and output capacitor ESR. In continuous
mode the average output current flows through L and
Efficiency Considerations
R
, but is chopped between the topside MOSFET
SENSE
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
andthesynchronousMOSFET.IfthetwoMOSFETshave
approximately the same R
, then the resistance
DS(ON)
of one MOSFET can simply be summed with the resis-
2
tances of L, R
and ESR to obtain I R losses. For
DS(ON)
SENSE
example, if each R
= 30mΩ, R = 50mΩ, R
L SENSE
= 10mΩ and R
= 40mΩ (sum of both input and
ESR
output capacitance losses), then the total resistance
is 130mΩ. This results in losses ranging from 3% to
13% as the output current increases from 1A to 5A for
a 5V output, or a 4% to 20% loss for a 3.3V output.
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percent-
age of input power.
Efficiency varies as the inverse square of V
for the
OUT
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3890-1 circuits: 1) IC VIN current, 2) INTVCC
regulator current, 3) I2R losses, 4) topside MOSFET
sameexternalcomponentsandoutputpowerlevel. The
combined effects of increasingly lower output voltages
andhighercurrentsrequiredbyhighperformancedigital
systemsisnotdoublingbutquadruplingtheimportance
of loss terms in the switching regulator system!
transition losses
.
1. The V current is the DC supply current given in the
IN
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high
ElectricalCharacteristicstable,whichexcludesMOSFET
driverandcontrolcurrents. V currenttypicallyresults
IN
input voltages (t
ypically 15V or greater). Transition
in a small (<0.1%) loss.
losses can be estimated from:
2. INTV current is the sum of the MOSFET driver and
CC
Transition Loss = (1.7) • V • 2 • I
• C
• f
IN
O(MAX)
RSS
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge, dQ, moves
Other hidden losses such as copper trace and internal
battery resistances can account for an additional 5%
to 10% efficiency degradation in portable systems. It
is very important to include these system level losses
during the design phase. The internal battery and fuse
resistancelossescanbeminimizedbymakingsurethat
from INTV to ground. The resulting dQ/dt is a current
CC
out of INTV that is typically much larger than the
CC
control circuit current. In continuous mode, I
GATECHG
= f(Q + Q ), where Q and Q are the gate charges of
T
B
T
B
C has adequate charge storage and very low ESR at
IN
the topside and bottom side MOSFETs.
the switching frequency. A 25W supply will typically
require a minimum of 20µF to 40µF of capacitance
having a maximum of 20mΩ to 50mΩ of ESR. The
LTC3890-1 2-phase architecture typically halves this
SupplyingINTV fromanoutput-derivedsourcepower
CC
through EXTV will scale the V current required for
CC
IN
thedriverandcontrolcircuitsbyafactorof(DutyCycle)/
38901fb
24
LTC3890-1
APPLICATIONS INFORMATION
input capacitance requirement over competing solu-
tions.OtherlossesincludingSchottkyconductionlosses
during dead-time and inductor core losses generally
account for less than 2% total additional loss.
Placing a power MOSFET directly across the output ca-
pacitor and driving the gate with an appropriate signal
generator is a practical wayto producea realistic load step
condition. The initial output voltage step resulting from
the step change in output current may not be within the
bandwidth of the feedback loop, so this signal cannot be
used to determine phase margin. This is why it is better
to look at the ITH pin signal which is in the feedback loop
andisthefilteredandcompensatedcontrolloopresponse.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to ∆ILOAD (ESR), where ESR is the effective
seriesresistanceofCOUT. ∆ILOAD alsobeginstochargeor
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. OPTI-
LOOP compensation allows the transient response to be
optimized over a wide range of output capacitance and
ESR values. The availability of the ITH pin not only allows
optimization of control loop behavior, but it also provides
a DC coupled and AC filtered closed-loop response test
point. The DC step, rise time and settling at this test
point truly reflects the closed-loop response. Assuming
apredominantlysecondordersystem,phasemarginand/
or damping factor can be estimated using the percentage
of overshoot seen at this pin. The bandwidth can also
be estimated by examining the rise time at the pin. The
ITH external components shown in Figure 13 circuit will
provide an adequate starting point for most applications.
The gain of the loop will be increased by increasing R
C
and the bandwidth of the loop will be increased by de-
creasing C . If R is increased by the same factor that C
C
C
C
is decreased, the zero frequency will be kept the same,
thereby keeping the phase shift the same in the most
critical frequency range of the feedback loop. The output
voltage settling behavior is related to the stability of the
closed-loopsystemandwilldemonstratetheactualoverall
supply performance.
A second, more severe transient is caused by switching
in loads with large (>1µF) supply bypass capacitors. The
dischargedbypasscapacitorsareeffectivelyputinparallel
with C , causing a rapid drop in V . No regulator can
OUT
OUT
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
C
LOAD
to C
is greater than 1:50, the switch rise time
OUT
should be controlled so that the load rise time is limited
to approximately 25 • C . Thus a 10µF capacitor would
LOAD
require a 250µs rise time, limiting the charging current
to about 200mA.
The ITH series R -C filter sets the dominant pole-zero
C
C
Design Example
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1µs to 10µs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop.
As a design example for one channel, assume V
=
IN
= 5A,
12V(nominal), V = 22V (max), V
= 3.3V, I
IN
OUT
MAX
V
= 75mV and f = 350kHz.
SENSE(MAX)
Theinductancevalueischosenfirstbasedona30%ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the FREQ pin
to GND, generating 350kHz operation. The minimum
inductance for 30% ripple current is:
VOUT
f L
( ) ( )
VOUT
∆IL =
1–
IN(NOM)
V
38901fb
25
LTC3890-1
APPLICATIONS INFORMATION
A 4.7µH inductor will produce 29% ripple current. The
peak inductor current will be the maximum DC value plus
one half the ripple current, or 5.73A. Increasing the ripple
current will also help ensure that the minimum on-time
of 95ns is not violated. The minimum on-time occurs at
C is chosen for an RMS current rating of at least 3A at
IN
temperature assuming only this channel is on. C
is
OUT
chosen with an ESR of 0.02Ω for low output ripple. The
output ripple in continuous mode will be highest at the
maximum input voltage. The output voltage ripple due to
ESR is approximately:
maximum V :
IN
VOUT
3.3V
V
= R
(∆I ) = 0.02Ω(1.45A) = 29mV
ORIPPLE
ESR L P-P
tON(MIN)
=
=
= 429ns
V
f
22V 350kHz
IN(MAX) ( )
(
)
PC Board Layout Checklist
The equivalent R
using the minimum value for the maximum current sense
threshold (43mV):
resistor value can be calculated by
SENSE
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the IC. These items are also illustrated graphically in the
layoutdiagramofFigure11.Figure12illustratesthecurrent
waveforms present in the various branches of the 2-phase
synchronousregulatorsoperatinginthecontinuousmode.
Check the following in your layout:
64mV
5.73A
RSENSE
≤
≈ 0.01Ω
Choosing 1% resistors: R = 25k and R = 78.7k yields
an output voltage of 3.32V.
A
B
1. Are the top N-channel MOSFETs MTOP1 and MTOP2
located within 1cm of each other with a common drain
ThepowerdissipationonthetopsideMOSFETcanbeeasily
estimated. Choosing a Fairchild FDS6982S dual MOSFET
connection at C ? Do not attempt to split the input
IN
decoupling for the two channels as it can cause a large
results in: R
= 0.035Ω/0.022Ω, C
= 215pF. At
DS(ON)
MILLER
resonant loop.
maximum input voltage with T(estimated) = 50°C:
2. Are the signal and power grounds kept separate? The
combined IC signal ground pin and the ground return
3.3V
22V
2
PMAIN
=
5A 1+ 0.005 50°C – 25°C
(
)
(
)
(
)
of C
must return to the combined C
(–) ter-
INTVCC
OUT
2 5A
0.035Ω + 22V
) (
2.5Ω 215pF •
) (
(
)
(
)
minals. The path formed by the top N-channel MOSFET,
2
Schottky diode and the C capacitor should have short
IN
1
1
leads and PC trace lengths. The output capacitor (–)
terminals should be connected as close as possible
to the (–) terminals of the input capacitor by placing
the capacitors next to each other and away from the
Schottky loop described above.
+
350kHz = 331mW
(
)
5V – 2.3V 2.3V
A short-circuit to ground will result in a folded back cur-
rent of:
95ns 22V
34mV
0.01Ω 2
1
3. DotheLTC3890-1V pins’resistivedividersconnectto
(
)
FB
ISC =
–
= 3.18A
the (+) terminals of C ? The resistive divider must be
OUT
4.7µH
connected between the (+) terminal of C
and signal
OUT
ground. The feedback resistor connections should not
be along the high current input feeds from the input
capacitor(s).
with a typical value of R
and δ = (0.005/°C)(25°C)
DS(ON)
= 0.125. The resulting power dissipated in the bottom
MOSFET is:
2
P
= 3.18A 1.125 0.022Ω
(
) (
) (
)
SYNC
= 250mW
which is less than under full-load conditions.
38901fb
26
LTC3890-1
APPLICATIONS INFORMATION
–
+
suggest noise pickup at the current or voltage sensing
inputs or inadequate loop compensation. Overcompensa-
tion of the loop can be used to tame a poor PC layout if
regulatorbandwidthoptimizationisnotrequired.Onlyafter
each controller is checked for its individual performance
should both controllers be turned on at the same time.
A particularly difficult region of operation is when one
controller channel is nearing its current comparator trip
pointwhentheotherchannelisturningonitstopMOSFET.
This occurs around 50% duty cycle on either channel due
to the phasing of the internal clocks and may cause minor
duty cycle jitter.
4. Are the SENSE and SENSE leads routed together with
minimumPCtracespacing?Thefiltercapacitorbetween
+
–
SENSE and SENSE should be as close as possible
to the IC. Ensure accurate current sensing with Kelvin
connections at the SENSE resistor.
5. Is the INTV decoupling capacitor connected close
CC
to the IC, between the INTV and the power ground
CC
pins? This capacitor carries the MOSFET drivers’ cur-
rent peaks. An additional 1µF ceramic capacitor placed
immediatelynexttotheINTV andPGNDpinscanhelp
CC
improve noise performance substantially.
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), andboostnodes(BOOST1, BOOST2)away
from sensitive small-signal nodes, especially from
the opposites channel’s voltage and current sensing
feedback pins. All of these nodes have very large and
fast moving signals and therefore should be kept on
the output side of the LTC3890-1 and occupy minimum
PC trace area.
Reduce V from its nominal level to verify operation of
IN
the regulator in dropout. Check the operation of the un-
dervoltage lockout circuit by further lowering V while
IN
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher out-
put currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
7.Useamodifiedstargroundtechnique:alowimpedance,
large copper area central grounding point on the same
side of the PC board as the input and output capacitors
with tie-ins for the bottom of the INTV decoupling
CC
capacitor, the bottom of the voltage feedback resistive
divider and the SGND pin of the IC.
PC Board Layout Debugging
for inductive coupling between C , Schottky and the top
IN
Start with one controller on at a time. It is helpful to use
a DC-50MHz current probe to monitor the current in the
inductorwhiletestingthecircuit.Monitortheoutputswitch-
ing node (SW pin) to synchronize the oscilloscope to the
internal oscillator and probe the actual output voltage as
well. Check for proper performance over the operating
voltage and current range expected in the application.
The frequency of operation should be maintained over the
input voltage range down to dropout and until the output
load drops below the low current operation threshold—
typically 25% of the maximum designed current level in
Burst Mode operation.
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
SGND pin of the IC.
An embarrassing problem, which can be missed in an
otherwise properly working switching regulator, results
when the current sensing leads are hooked up backwards.
The output voltage under this improper hookup will still
be maintained but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
Thedutycyclepercentageshouldbemaintainedfromcycle
to cycle in a well-designed, low noise PCB implementa-
tion. Variation in the duty cycle at a subharmonic rate can
38901fb
27
LTC3890-1
APPLICATIONS INFORMATION
ITH1
TRACK/SS1
PGOOD1
TG1
R
PU1
V
PULL-UP
V
PGOOD1
FB1
L1
R
R1*
C1*
SENSE
D1*
V
+
–
OUT1
SENSE1
SENSE1
SW1
C
B1
LTC3890-1
BOOST1
M1
M2
BG1
FREQ
R
IN
C
OUT1
V
f
IN
1µF
CERAMIC
IN
PLLIN/MODE
RUN1
+
C
VIN
PGND
GND
RUN2
+
EXTV
CC
CC
C
IN
+
V
SGND
C
IN
INTVCC
–
INTV
SENSE2
C
OUT2
1µF
C2*
CERAMIC
+
SENSE2
BG2
R2*
M4
L2
M3
D2*
BOOST2
V
FB2
C
B2
SW2
TG2
ITH2
R
SENSE
V
OUT2
TRACK/SS2
38901 F11
*R1, R2, C1, C2, D1, D2 ARE OPTIONAL.
Figure 11. Recommended Printed Circuit Layout Diagram
38901fb
28
LTC3890-1
APPLICATIONS INFORMATION
SW1
L1
R
SENSE1
V
OUT1
D1
C
R
L1
OUT1
V
IN
R
IN
C
IN
SW2
L2
R
SENSE2
V
OUT2
D2
C
R
L2
OUT2
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
38901 F12
Figure 12. Branch Current Waveforms
38901fb
29
LTC3890-1
TYPICAL APPLICATIONS
+
–
SENSE1
SENSE1
INTV
C1
1nF
CC
R
B1
100k
100k
R
PGOOD1
BG1
A1
31.6k
V
FB1
C
100pF
R
ITH1A
MBOT1
R
SENSE1
8mΩ
V
3.3V
5A
OUT1
C
1000pF
ITH1
SW1
ITH1
34.8k
L1
BOOST1
ITH1
4.7µH
C
OUT1
470µF
C
LTC3890-1
C
SS1
0.01µF
B1
0.1µF
TRACK/SS1
TG1
MTOP1
D1
V
IN
V
IN
9V TO 60V
C
IN
220µF
PLLIN/MODE
SGND
INTV
CC
C
INT
4.7µF
PGND
EXTV
V
CC
OUT2
RUN1
RUN2
FREQ
R
FREQ
41.2k
D2
TG2
MTOP2
C
SS2
0.01µF
C
0.1µF
B2
L2
8µH
R
TRACK/SS2
ITH2
BOOST2
SW2
SENSE2
R
ITH2
34.8k
C
470pF
10mΩ
V
8.5V
3A
ITH2
R
OUT2
C
OUT2
A2
10.5k
330µF
MBOT2
V
BG2
FB2
R
B2
100k
–
SENSE2
C2
1nF
+
SENSE2
38901 TA02a
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-472KL
L2: COILCRAFT SER1360-802KL
C
C
: SANYO 6TPE470M
: SANYO 10TPE330M
OUT1
OUT2
D1, D2: DFLS1100
Figure 13. High Efficiency Dual 8.5V/3.3V Step-Down Converter
Efficiency and Power Loss
vs Output Current
Efficiency vs Load Current
Efficiency vs Input Voltage
10000
1000
100
100
90
80
70
60
50
40
30
20
100
90
80
70
60
50
40
30
20
100
98
96
94
92
90
88
86
84
V
V
= 12V
IN
OUT
V
= 8.5V
BURST EFFICIENCY
OUT
= 3.3V
V
OUT
= 3.3V
V
= 8.5V
OUT2
CCM LOSS
BURST LOSS
PULSE-SKIPPING
LOSS
10
V
OUT1
= 3.3V
CCM EFFICIENCY
1
PULSE-SKIPPING
EFFICIENCY
10
0
10
0
82
80
V
IN
= 12V
I
= 2A
LOAD
5
0.1
10
0.0001 0.001
0.01
0.1
1
0.0001 0.001
0.01
0.1
1
10
0
10 15 20 25 30 35 40 45 50 55 60
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
INPUT VOLTAGE (V)
38901 TA02b
38901 TA02c
38901 TA02d
38901fb
30
LTC3890-1
TYPICAL APPLICATIONS
High Efficiency 8.5V Dual-Phase Step-Down Converter
+
–
SENSE1
SENSE1
INTV
C1
1nF
CC
R
B1
100k
100k
R
A1
PGOOD1
BG1
10.5k
V
FB1
MBOT1
C
100pF
ITH1A
L1
8µH
V
8.5V
6A
OUT1
SW1
R
ITH1
34.8k
R
SENSE1
BOOST1
ITH1
10mΩ
C
OUT1
330µF
C
ITH1
C
C
SS1
0.01µF
LTC3890-1
B1
470pF
0.1µF
TRACK/SS1
TG1
MTOP1
D1
V
IN
INTV
V
R
CC
IN
MODE
9V TO 60V
C
100k
IN
220µF
PLLIN/MODE
SGND
INTV
CC
C
INT
4.7µF
R
PGND
RUN
1000k
V
OUT
EXTV
CC
V
IN
RUN1
RUN2
FREQ
D2
R
FREQ
41.2k
TG2
MTOP2
C
0.1µF
B2
TRACK/SS2
C
ITH2
100pF
R
L2
8µH
BOOST2
SW2
SENSE2
10mΩ
ITH2
C
OUT2
MBOT2
BG2
330µF
V
FB2
–
+
SENSE2
C2
1nF
SENSE2
38901 TA03
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1, L2: COILCRAFT SER1360-802KL
C
, C
: SANYO 10TPE330M
OUT1 OUT2
D1, D2: DFLS1100
38901fb
31
LTC3890-1
TYPICAL APPLICATIONS
High Efficiency Dual 12V/5V Step-Down Converter
+
SENSE1
INTV
CC
C1
1nF
R
B1
100k
100k
–
PGOOD1
BG1
SENSE1
R
A1
6.98k
V
FB1
C
100pF
R
ITH1A
MBOT1
R
SENSE1
9mΩ
V
12V
3A
OUT1
C
470pF
ITH1
ITH1
SW1
34.8k
L1
8µH
C
OUT1
BOOST1
ITH1
180µF
C
LTC3890-1
TRACK/SS1
C
SS1
0.01µF
B1
0.47µF
TG1
MTOP1
D1
V
IN
V
IN
12.5V TO 60V
C
IN
220µF
PLLIN/MODE INTV
SGND
CC
C
INT
4.7µF
PGND
EXTV
CC
RUN1
RUN2
FREQ
R
FREQ
D2
41.2k
TG2
MTOP2
C
0.47µF
C
SS2
0.01µF
B2
L2
4.7µH
R
BOOST2
SW2
SENSE2
TRACK/SS2
ITH2
10mΩ
V
5V
5A
OUT2
R
ITH2
20k
C
470pF
ITH2
C
OUT2
MBOT2
BG2
470µF
R
A2
18.7k
V
FB2
–
+
SENSE2
R
B2
C2
1nF
100k
SENSE2
38901 TA04
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-802KL
L2: COILCRAFT SER1360-472KL
C
C
: 16SVP180MX
OUT1
OUT2
: SANYO 6TPE470M
D1, D2: DFLS1100
38901fb
32
LTC3890-1
TYPICAL APPLICATIONS
High Efficiency Dual 24V/5V Step-Down Converter
R
B1
487k
+
SENSE1
INTV
CC
C1
1nF
C
33pF
F1
100k
–
SENSE1
PGOOD1
BG1
R
A1
16.9k
V
FB1
C
100pF
R
ITH1A
L1
22µH
MBOT1
MTOP1
R
SENSE1
V
24V
1A
25mΩ
OUT1
C
680pF
ITH1
46k
ITH1
SW1
BOOST1
ITH1
C
OUT1
22µF
C
LTC3890-1
TRACK/SS1
C
SS1
0.01µF
B1
×2 CERAMIC
0.47µF
TG1
D1
V
IN
V
IN
28V TO 60V
C
IN
220µF
PLLIN/MODE INTV
SGND
CC
C
INT
4.7µF
PGND
EXTV
CC
RUN1
RUN2
FREQ
R
FREQ
D2
60k
TG2
MTOP2
C
0.47µF
C
SS2
0.01µF
R
B2
L2
4.7µH
R
BOOST2
SW2
SENSE2
TRACK/SS2
ITH2
10mΩ
V
C
470pF
ITH2
20k
OUT2
5V
5A
ITH2
C
OUT2
MBOT2
BG2
470µF
V
FB2
R
A2
R
B2
100k
18.7k
–
+
SENSE2
SENSE2
C2
1nF
38901 TA05
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: SUMIDA CDR7D43MN
L2: COILCRAFT SER1360-472KL
C
: KEMET T525D476MO16E035
: SANYO 6TPE470M
OUT1
OUT2
C
D1, D2: DFLS1100
38901fb
33
LTC3890-1
PACKAGE DESCRIPTION
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.386 – .393*
(9.804 – 9.982)
.045 ±.005
.033
(0.838)
REF
28 27 26 25 24 23 22 21 20 19 18 17 1615
.254 MIN
.150 – .165
.229 – .244
.150 – .157**
(5.817 – 6.198)
(3.810 – 3.988)
.0165 ±.0015
.0250 BSC
1
2
3
4
5
6
7
8
9 10 11 12 13 14
RECOMMENDED SOLDER PAD LAYOUT
.015 ± .004
(0.38 ± 0.10)
.0532 – .0688
(1.35 – 1.75)
× 45°
.004 – .0098
(0.102 – 0.249)
.0075 – .0098
(0.19 – 0.25)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
.008 – .012
.0250
(0.635)
BSC
GN28 (SSOP) 0204
(0.203 – 0.305)
TYP
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
38901fb
34
LTC3890-1
REVISION HISTORY
REV
DATE
01/11 Added MP-grade and H-grade. Changes reflected throughout the data sheet.
10/12 Added INTV to Absolute Maximum Ratings
DESCRIPTION
PAGE NUMBER
A
1-36
2
B
CC
38901fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
35
LTC3890-1
TYPICAL APPLICATION
High Efficiency Dual 12V/3.3V Step-Down Converter
+
SENSE1
INTV
CC
C1
1nF
R
B1
100k
100k
–
PGOOD1
BG1
SENSE1
R
A1
6.98k
V
FB1
C
100pF
R
ITH1A
MBOT1
R
SENSE1
9mΩ
V
12V
3A
OUT1
C
470pF
ITH1
ITH1
SW1
34.8k
L1
8µH
C
OUT1
BOOST1
ITH1
180µF
C
LTC3890-1
C
SS1
0.01µF
B1
0.47µF
TRACK/SS1
TG1
MTOP1
D1
V
IN
V
IN
12.5V TO 60V
C
IN
220µF
PLLIN/MODE
SGND
INTV
CC
C
INT
4.7µF
PGND
V
OUT1
EXTV
CC
RUN1
RUN2
FREQ
R
FREQ
D2
41.2k
TG2
MTOP2
C
0.47µF
C
SS2
0.01µF
B2
L2
4.7µH
R
BOOST2
SW2
SENSE2
TRACK/SS2
ITH2
10mΩ
V
3.3V
5A
OUT2
R
ITH2
34.8k
C
1000pF
ITH2
C
C
ITH2A
OUT2
MBOT2
BG2
100pF
470µF
V
FB2
R
A2
–
+
SENSE2
31.6k
R
B2
C2
1nF
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-802KL
100k
L2: COILCRAFT SER1360-472KL
SENSE2
C
C
: 16SVP180MX
: SANYO 6TPE470M
OUT1
OUT2
D1, D2: DFLS1100
38901 TA06
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT3891
60V, Low I , Synchronous Step-Down DC/DC Controller
PLL Fixed Frequency 50kHz to 900kHz, 4V ≤ V ≤ 60V,
IN
Q
0.8V ≤ V
≤ 24V, TSSOP-20E, 3mm × 4mm QFN-20
OUT
LTC3857/LTC3857-1/ Low I , Dual Output 2-Phase Synchronous Step-Down
PLL Fixed Frequency 50kHz to 900kHz, 4V ≤ V ≤ 38V,
IN
Q
LTC3858/LTC3858-1 DC/DC Controllers with 99% Duty Cycle
0.8V ≤ V
≤ 24V, I = 50µA/170µA
OUT Q
LTC3834/LTC3834-1/ Low I , Single Output Synchronous Step-Down
PLL Fixed Frequency 140kHz to 650kHz, 4V ≤ V ≤ 36V,
IN
Q
LTC3835/LTC3835-1 DC/DC Controllers with 99% Duty Cycle
0.8V ≤ V
≤ 10V, I = 30µA/80µA
OUT Q
LTC3810
100V Synchronous Step-Down DC/DC Controller
Constant On-Time Valley Current Mode, 4V ≤ V ≤ 100V,
IN
0.8V ≤ V
≤ 0.93V , SSOP-28
IN
OUT
LTC3859
Low I , Triple Output Buck/Buck/Boost Synchronous
Outputs (≥5V) Remain in Regulation Through Cold Crank,
2.5V ≤ V ≤ 38V, V Up to 24V, V Up to 60V
Q
DC/DC Controller
IN
OUT(BUCK)
OUT(BOOST)
38901fb
LT 1012 REV B • PRINTED IN USA
36 LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
●
●
LINEAR TECHNOLOGY CORPORATION 2010
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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