ADP3179JRU-REEL7 [ONSEMI]
IC SWITCHING CONTROLLER, PDSO20, TSSOP-20, Switching Regulator or Controller;
| 型号： | ADP3179JRU-REEL7 |
| 厂家： | 
ONSEMI |
| 描述： | IC SWITCHING CONTROLLER, PDSO20, TSSOP-20, Switching Regulator or Controller 开关 光电二极管 |
| 文件： | 总16页 (文件大小：226K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
4-Bit Programmable
Synchronous Buck Controllers
a
ADP3159/ADP3179
FEATURES
FUNCTIONAL BLOCK DIAGRAM
Optimally Compensated Active Voltage Positioning
with Gain and Offset Adjustment (ADOPT™) for
Superior Load Transient Response
Complies with VRM 8.4 Specifications with Lowest
System Cost
4-Bit Digitally Programmable 1.3 V to 2.05 V Output
N-Channel Synchronous Buck Driver
Two On-Board Linear Regulator Controllers
Total Accuracy ꢀ0.8% Over Temperature
High Efficiency Current-Mode Operation
Short Circuit Protection
VCC
CT
ADP3159/ADP3179
SET
UVLO
& BIAS
DRVH
DRVL
OSCILLATOR
RESET
PWM
DRIVE
CROWBAR
REFERENCE
REF
GND
DAC+20%
DAC–20%
V
LR1
PWRGD
LRFB1
Power Good Output
Overvoltage Protection Crowbar Protects Micro-
processors with No Additional External Components
LRDRV1
V
LR2
CS–
CS+
FB
APPLICATIONS
CMP
LRFB2
LRDRV2
COMP
Core Supply Voltage Generation for:
– +
Intel Pentium® III
g
m
Intel Celeron™
REF
VID DAC
VID3
VID2
VID1
VID0
GENERAL DESCRIPTION
of output capacitors and smallest footprint. Unlike voltage-
mode and standard current-mode architectures, active voltage
positioning adjusts the output voltage as a function of the load
current so it is always optimally positioned for a system tran-
sient. The devices also provide accurate and reliable short
circuit protection and adjustable current limiting. They also
include an integrated overvoltage crowbar function to protect
the microprocessor from destruction in case the core supply
exceeds the nominal programmed voltage by more than 20%.
The ADP3159 and ADP3179 are highly efficient output syn-
chronous buck switching regulator controllers optimized for
converting a 5 V main supply into the core supply voltage
required by high-performance processors. These devices use an
internal 4-bit DAC to read a voltage identification (VID) code
directly from the processor, which is used to set the output
voltage between 1.3 V and 2.05 V. They use a current mode,
constant off-time architecture to drive two N-channel
MOSFETs at a programmable switching frequency that can be
optimized for regulator size and efficiency.
The ADP3159 and ADP3179 contain two fixed-output volt-
age linear regulator controllers that are designed to drive
external N-channel MOSFETs. The outputs are internally
fixed at 2.5 V and 1.8 V in the ADP3159, while the ADP3179
provides adjustable output, which is set using an external
resistor divider. These linear regulators are used to generate
the auxiliary voltages (AGP, GTL, etc.) required in most moth-
erboard designs, and have been designed to provide a high
bandwidth load-transient response.
The ADP3159 and ADP3179 also use a unique supplemental
regulation technique called Analog Devices Optimal Position-
ing Technology (ADOPT) to enhance load transient
performance. Active voltage positioning results in a dc/dc con-
verter that meets the stringent output voltage specifications
for high-performance processors, with the minimum number
The ADP3159 and ADP3179 are specified over the commercial
temperature range of 0°C to 70°C and are available in a 20-lead
TSSOP package.
ADOPT is a trademark of Analog Devices, Inc.
Pentium is a registered trademark of Intel Corporation.
Celeron is a trademark of Intel Corporation.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, norforanyinfringementsofpatentsorotherrightsofthirdpartiesthat
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
Fax: 781/326-8703
www.analog.com
© Analog Devices, Inc., 2001
ADP3159/ADP3179–SPECIFICATIONS1
(VCC = 12 V, TA = 0ꢁC to 70ꢁC, unless otherwise noted.)
Parameter
Symbol
Conditions
Min
Typ
Max Unit
FEEDBACK INPUT
Output Accuracy
1.3 V Output
1.65 V Output
2.05 V Output
Line Regulation
Crowbar Trip Point
Crowbar Reset Point
Crowbar Response Time
VFB
Figure 1
Figure 1
Figure 1
1.289
1.637
2.034
1.3
1.311 V
1.663 V
2.066 V
%
1.65
2.05
0.06
120
50
∆VOUT
VCROWBAR
VCC = 10 V to 14 V
% of Nominal DAC Voltage
% of Nominal DAC Voltage
Overvoltage to DRVL Going High
115
40
125
60
%
%
ns
tCROWBAR
400
VID INPUTS
Input Low Voltage
Input High Voltage
Input Current
Pull-Up Resistance
Internal Pull-Up Voltage
VIL(VID)
VIH(VID)
IVID
0.6
250
5.7
V
V
µA
kΩ
V
2.0
VID(X) = 0 V
185
30
5.4
RVID
20
5.0
OSCILLATOR
Off Time
CT Charge Current
TA = 25°C, CT = 200 pF
TA = 25°C, VOUT in Regulation
TA = 25°C, VOUT = 0 V
3.5
130
25
4.0
150
35
4.5
170
45
µs
µA
µA
ICT
ERROR AMPLIFIER
Output Resistance
RO(ERR)
gm(ERR)
IO(ERR)
1
mΩ
Transconductance
2.05
600
2.2
625
3.0
750
500
2.35 mmho
Output Current
FB– Forced to VOUT – 3%
µA
V
mV
kHz
Maximum Output Voltage
Output Disable Threshold
–3 dB Bandwidth
VCOMP(MAX) FB– Forced to VOUT – 3%
VCOMP(OFF)
BWERR
900
COMP = Open
CURRENT SENSE
Threshold Voltage
VCS(TH)
FB– Forced to VOUT – 3%
FB– ≤ 0.45 V
0.8 V ≤ COMP ≤ 1 V
CS+ = CS– = VOUT
CS+ – (CS–) > 87 mV to DRVH
Going Low
69
35
78
45
1
0.5
50
87
54
5
mV
mV
mV
µA
Input Bias Current
Response Time
I
CS+, ICS–
5
tCS
ns
OUTPUT DRIVERS
Output Resistance
Output Transition Time
RO(DRV(X))
tR, tF
IL = 50 mA
CL = 3000 pF
6
80
Ω
ns
LINEAR REGULATORS
Feedback Current
ILRFB(X)
VLRFB(1)
0.3
2.5
1.0
1.8
1.0
1
µA
V
V
V
V
V
LR1 Feedback Voltage
ADP3159 Figure 2, VCC = 4.5 V to 12.6 V
2.44
2.56
1.03
1.85
1.03
ADP3179 Figure 2, VCC = 2-4.5 V to 12.6 V 0.97
ADP3159 Figure 2, VCC = 4.5 V to 12.6 V 1.75
ADP3179 Figure 2, VCC = 2-4.5 V to 12.6 V 0.97
LR2 Feedback Voltage
Driver Output Voltage
VLRFB(2)
VLRDRV(X)
VCC = 4.5 V, VLRFB(X) = 0 V
4.2
POWER GOOD COMPARATOR
Undervoltage Threshold
Undervoltage Hysteresis
Overvoltage Threshold
Overvoltage Reset Point
Output Voltage Low
VPWRGD(UV) % of Nominal DAC Voltage
% of Nominal DAC Voltage
VPWRGD(OV) % of Nominal DAC Voltage
% of Nominal DAC Voltage
75
80
85
%
5
%
115
40
120
50
250
250
125
60
500
%
%
mV
ns
VOL(PWRGD) IPWRGD(SINK) = 1 mA
Response Time
SUPPLY
DC Supply Current2
UVLO Threshold Voltage
UVLO Hysteresis
ICC
VUVLO
7
7
1
9
7.25
1.2
mA
V
V
6.75
0.8
NOTES
1All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC).
2Dynamic supply current is higher due to the gate charge being delivered to the external MOSFETs.
Specifications subject to change without notice.
–2–
REV. A
ADP3159/ADP3179
ABSOLUTE MAXIMUM RATINGS*
PIN FUNCTION DESCRIPTIONS
Pin Mnemonic Function
1, 19 NC No Connection.
2–5 VID3, VID2, Voltage Identification DAC Inputs. These
VID1, VID0 pins are pulled up to an internal reference,
providing a Logic One if left open. The
VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +15 V
DRVH, DRVL, LRDRV1, LRDRV2 . . . . –0.3 V to VCC + 0.3 V
All Other Inputs and Outputs . . . . . . . . . . . . –0.3 V to +10 V
Operating Ambient Temperature Range . . . . . . . 0°C to 70°C
Operating Junction Temperature . . . . . . . . . . . . . . . . . 125°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
θJA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 143°C/W
Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . 300°C
Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . 215°C
Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220°C
DAC output programs the FB regulation
voltage from 1.3 V to 2.05 V.
6
PWRGD
Open drain output that signals when the
output voltage is in the proper operating
range.
*This is a stress rating only; operation beyond these limits can cause the device to
be permanently damaged. Unless otherwise specified, all voltages are referenced
to GND.
7, 15 LRFB1,
LRFB2
Feedback connections for the linear
regulator controllers.
8, 14 LRDRV1,
LRDRV2
Gate drives for the respective linear
regulator N-channel MOSFETs.
9
FB
Feedback Input. Error amplifier input for
remote sensing of the output voltage.
Current Sense Negative Node. Negative
input for the current comparator.
10
CS–
PIN CONFIGURATION
RU-20
11
CS+
Current Sense Positive Node. Positive
input for the current comparator. The
output current is sensed as a voltage at this
pin with respect to CS–.
External capacitor connected from CT to
ground sets the Off-time of the device.
Error Amplifier Output and Compensation
Point. The voltage at this output programs
the output current control level between
CS+ and CS–.
1
2
20
GND
NC
VID0
19 NC
12
13
CT
3
18
17
16
15
14
13
12
11
DRVH
DRVL
VCC
VID1
4
VID2
COMP
ADP3159/
ADP3179
5
VID3
6
PWRGD
LRFB1
LRDRV1
FB
TOP VIEW
(Not to Scale)
LRFB2
LRDRV2
COMP
CT
7
8
16
17
VCC
DRVL
Supply Voltage for the device.
9
Low-Side MOSFET Drive. Gate drive for
the synchronous rectifier N-channel
MOSFET. The voltage at DRVL swings
from GND to VCC.
10
CS+
CS–
NC = NO CONNECT
18
20
DRVH
GND
High-side MOSFET Drive. Gate drive for
the buck switch N-channel MOSFET.
The voltage at DRVH swings from GND
to VCC.
Ground Reference. GND should have a
low impedance path to the source of hte
synchronous MOSFET.
ORDERING GUIDE
Temperature
Range
LDO
Voltage
Package
Description
Package
Option
Model
ADP3159JRU
ADP3179JRU
0°C to 70°C
0°C to 70°C
2.5 V, 1.8 V
Adjustable
Thin Shrink Small Outline
Thin Shrink Small Outline
RU-20
RU-20
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the ADP3159 and the ADP3179 feature proprietary ESD protection circuitry, permanent damage
may occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
REV. A
–3–
–Typical Performance Characteristics
ADP3159/ADP3179
TEK RUN
TRIG'D
60
50
40
30
20
10
0
V
CC
1
V
CORE
2
CH1
5.00V B
CH2 500mV B M 10.0ms A CH1
W
5.90V
0
100
200
300
400
500
600
700
800
W
OSCILLATOR FREQUENCY – kHz
0.00000 s
TPC 4. Power-On Start-Up Waveform
TPC 1. Supply Current vs. Operating Frequency Using
MOSFETs of Figure 3
TEK RUN
DRVH
TRIG'D
25
T
= 25ꢁC
A
V
= 1.65V
OUT
20
15
10
1
5
0
DRVL
CH1
–0.5
0
0.5
5.00V
B
CH2
5.00V B M 1.00ꢂs
A
CH1
5.90V
W
W
OUTPUT ACCURACY – % of Nominal
–2.6500ꢂs
TPC 5. Output Accuracy Distribution
TPC 2. Gate Switching Waveforms Using MOSFETs of
Figure 3
TEK RUN
DRVH
TRIG'D
DRVL
CH1
CH1
2.00V
B
CH2 2.00V B M 1.00ns
W
A
5.88V
W
150.000ꢂs
TPC 3. Driver Transition Waveforms Using MOSFETs of
Figure 3
–4–
REV. A
ADP3159/ADP3179
Analog Devices Optimal Positioning Technology (ADOPT)
adjusts the output voltage as a function of the load current so
that it is always optimally positioned for a load transient. Stan-
dard (passive) voltage positioning, sometimes recommended for
use with other architectures, has poor dynamic performance
which renders it ineffective under the stringent repetitive tran-
sient conditions specified in Intel VRM documents. Consequently,
such techniques do not allow the minimum possible number of
output capacitors to be used. ADOPT, as used in the ADP3159
and ADP3179, provides a bandwidth for transient response that
is limited only by parasitic output inductance. This yields opti-
mal load transient response with the minimum number of
output capacitors.
ADP3159/
ADP3179
1
2
GND
NC
20
19
VID0
NC
3
DRVH 18
VID1
4-BIT CODE
4
17
16
15
14
13
12
11
VID2
DRVL
VCC
5
12V
VID3
+
1ꢂF
100nF
6
PWRGD
LRFB1
LRDRV1
FB
LRFB2
LRDRV2
COMP
CT
7
8
100ꢃ
V
9
FB
100nF
10
CS–
CS+
AD820
Cycle-by-Cycle Operation
NC = NO CONNECT
1.2V
During normal operation (when the output voltage is regulated),
the voltage error amplifier and the current comparator are the
main control elements. During the on-time of the high-side
MOSFET, the current comparator monitors the voltage between
the CS+ and CS– pins. When the voltage level between the two
pins reaches the threshold level, the DRVH output is switched to
ground, which turns off the high-side MOSFET. The timing
capacitor CT is then charged at a rate determined by the off-time
controller. While the timing capacitor is charging, the DRVL
output goes high, turning on the low-side MOSFET. When the
voltage level on the timing capacitor has charged to the upper
threshold voltage level, a comparator resets a latch. The output
of the latch forces the low-side drive output to go low and the
high-side drive output to go high. As a result, the low-side switch
is turned off and the high-side switch is turned on. The sequence
is then repeated. As the load current increases, the output volt-
age starts to decrease. This causes an increase in the output of
the voltage-error amplifier, which, in turn, leads to an increase
in the current comparator threshold, thus tracking the load current. To
prevent cross conduction of the external MOSFETs, feedback is
incorporated to sense the state of the driver output pins. Before
the low-side drive output can go high, the high-side drive output
must be low. Likewise, the high-side drive output is unable to
go high while the low-side drive output is high.
Figure 1. Closed Loop Output Voltage Accuracy
Test Circuit
ADP3159/
ADP3179
1
GND 20
NC
2
3
19
VID0
VID1
VID2
NC
18
17
16
15
14
13
12
11
DRVH
DRVL
VCC
VCC
4
+
1ꢂF
100nF
5
VID3
6
V
PWRGD
LRFB1
LRDRV1
LRFB2
LRDRV2
COMP
CT
LR2
V
7
LR1
10nF
8
10nF
9
FB
10
CS–
CS+
NC = NO CONNECT
Figure 2. Linear Regulator Output Voltage Accuracy
Test Circuit
THEORY OF OPERATION
The ADP3159 and ADP3179 use a current-mode, constant
off-time control technique to switch a pair of external N-channel
MOSFETs in a synchronous buck topology. Constant off-time
operation offers several performance advantages, including that no
slope compensation is required for stable operation. A unique
feature of the constant off-time control technique is that since
the off-time is fixed, the converter’s switching frequency is a
function of the ratio of input voltage to output voltage. The fixed
off-time is programmed by the value of an external capacitor
connected to the CT pin. The on-time varies in such a way
that a regulated output voltage is maintained as described below
in the cycle-by-cycle operation. The on-time does not vary under
fixed input supply conditions, and it varies only slightly as a func-
tion of load. This means that the switching frequency remains
fairly constant in a standard computer application.
Power Good
The ADP3159 has an internal monitor that senses the output
voltage and drives the PWRGD pin of the device. This pin is an
open drain output whose high level (when connected to a pull-up
resistor) indicates that the output voltage has been within a 20%
regulation band of the targeted value for more than 500 ms. The
PWRGD pin will go low if the output is outside the regulation
band for more than 500 ms.
Output Crowbar
An added feature of using an N-channel MOSFET as the syn-
chronous switch is the ability to crowbar the output with the same
MOSFET. If the output voltage is 20% greater than the targeted
value, the controller IC will turn on the lower MOSFET,
which will current-limit the source power supply or blow its fuse,
pull down the output voltage, and thus save the microprocessor
from destruction. The crowbar function releases at approximately
50% of the nominal output voltage. For example, if the output
is programmed to 1.5 V, but is pulled up to 1.85 V or above, the
crowbar will turn on the lower MOSFET. If in this case the output
is pulled down to less than 0.75 V, the crowbar will release,
allowing the output voltage to recover to 1.5 V if the fault
condition has been removed.
Active Voltage Positioning
The output voltage is sensed at the CS– pin. A voltage error
amplifier, (gm), amplifies the difference between the output
voltage and a programmable reference voltage. The reference
voltage is programmed to between 1.3 V and 2.05 V by an inter-
nal 4-bit DAC that reads the code at the voltage identification
(VID) pins (Refer to Table I for output voltage vs. VID pin code
information). A unique supplemental regulation technique called
REV. A
–5–
ADP3159/ADP3179
D2
MBR052LT1
5V STANDBY
12V
D3
MBR052LT1
C6
1ꢂF
L2
1ꢂH
5V
+
+
+
C7
22ꢂF
C8
1000ꢂF
C9
1000ꢂF
ADP3159/
ADP3179
Q4
NC
U1 GND
1
2
20
19
18
17
SUB45N03-13L
1000ꢂFx5
R12
4mꢃ
VCC CORE
1.30VTO
2.05V
24mꢃ (EACH)
VID0
VID1
VID2
NC
DRVH
DRVL
5V
L1
1.7ꢂH
+
+
+
+
+
3
15A
FROM CPU
R1
10kꢃ
Q3
4
SUB75N03-07
C17 C18 C19 C20 C21
5
VID3
VCC 16
POWER
GOOD
PWRGD
LRFB1
LRDRV1
FB
LRFB2
15
14
13
12
11
6
C11
68pF
3.3V
LRDRV2
COMP
CT
7
C2
68pF
3.3V
R8
R11
C16
8
78.7kꢃ
10kꢃ
1ꢂF
9
C15
R2
10kꢃ
C4
2.7nF
Q2
SUB45N03-13L
1ꢂF
CS–
CS+
10
Q1
R7
10.5kꢃ
SUB45N03-13L
V
NC = NO CONNECT
LR2
1.8V,
2A
+
C5
100ꢂF
V
C3
150pF
LR1
R4
220ꢃ
2.5V, 2A
+
C1
100ꢂF
R3
220ꢃ
C10
1nF
Figure 3. 15 A Pentium III Application Circuit
On-board Linear Regulator Controllers
start-up the linear outputs will track the 3.3 V supply up until
they reach their respective regulation points, regardless of the
state of the 12 V supply. Once the 12 V supply has exceeded the
5 VSB supply by more than a diode drop, the controller IC
will track the 12 V supply. Once the 12 V supply has risen
above the UVLO value, the switching regulator will begin its
start-up sequence.
The ADP3159 and ADP3179 include two linear regulator controllers
to provide a low cost solution for generating additional supply rails.
In the ADP3159, these regulators are internally set to 2.5 V (LR1)
and 1.8 V (LR2) with 2.5% accuracy. The ADP3179 is designed
to allow the outputs to be set externally using a resistor divider.
The output voltage is sensed by the high input impedance LRFB(x)
pin and compared to an internal fixed reference.
Table I. Output Voltage vs. VID Code
The LRDRV(x) pin controls the gate of an external N-channel
MOSFET resulting in a negative feedback loop. The only addi-
tional components required are a capacitor and resistor for
stability. Higher output voltages can be generated by placing
a resistor divider between the linear regulator output and its
respective LRFB pin. The maximum output load current is
determined by the size and thermal impedance of the external
power MOSFET that is placed in series with the supply and
controlled by the ADP3159.
VID3
VID2
VID1
VID0
VOUT(NOM)
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1.30 V
1.35 V
1.40 V
1.45 V
1.50 V
1.55 V
1.60 V
1.65 V
1.70 V
1.75 V
1.80 V
1.85 V
1.90 V
1.95 V
2.00 V
2.05 V
The linear regulator controllers have been designed so that they
remain active even when the switching controller is in UVLO
mode to ensure that the output voltages of the linear regulators
will track the 3.3 V supply as required by Intel design specifica-
tions. By diode ORing the VCC input of the IC to the 5 VSB
and 12 V supplies as shown in Figure 3, the switching output
will be disabled in standby mode, but the linear regulators will
begin conducting once VCC rises above about 1 V. During
–6–
REV. A
ADP3159/ADP3179
APPLICATION INFORMATION
RSENSE is the resistance of the sense resistor
Specifications for a Design Example
The design parameters for a typical 750 MHz Pentium III appli-
cation (shown in Figure 3) are as follows:
(estimated value: 4 mΩ)
RL is the resistance of the inductor
(estimated value: 3 mΩ)
Input Voltage: (VIN) = 5 V
Inductance Selection
Auxiliary Input: (VCC) = 12 V
The choice of inductance determines the ripple current in the
inductor. Less inductance leads to more ripple current, which
increases the output ripple voltage and the conduction losses in
the MOSFETs, but allows using smaller-size inductors and, for
a specified peak-to-peak transient deviation, output capacitors
with less total capacitance. Conversely, a higher inductance means
lower ripple current and reduced conduction losses, but requires
larger-size inductors and more output capacitance for the same
peak-to-peak transient deviation. The following equation shows
the relationship between the inductance, oscillator frequency,
peak-to-peak ripple current in an inductor and input and
output voltages.
Output Voltage (VVID) = 1.7 V
Maximum Output Current (IO(MAX)) = 15 A
Minimum Output Current (IO(MIN)) = 1 A
Static tolerance of the supply voltage for the processor core
(∆VO) = +40 mV (–80 mV) = 120 mV
Transient tolerance (for less than 2 µs) of the supply voltage
for the processor core when the load changes between the
minimum and maximum values with a di/dt of 20 A/µs
(∆VO(TRANSIENT)) = +80 mV (–130 mV) = 210 mV
Input current di/dt when the load changes between the mini-
mum and maximum values < 0.1 A/µs.
VOUT × tOFF
IL(RIPPLE )
L =
(4)
The above requirements correspond to Intel’s published power
supply requirements based on VRM 8.4 guidelines.
For 4 A peak-to-peak ripple current, which corresponds to
approximately 25% of the 15 A full-load dc current in an induc-
tor, Equation 4 yields an inductance of:
CT Selection for Operating Frequency
The ADP3159 uses a constant off-time architecture with tOFF
determined by an external timing capacitor CT. Each time the
high-side N-channel MOSFET switch turns on, the voltage across
CT is reset to 0 V. During the off-time, CT is discharged by a
constant current of 150 µA. Once CT reaches 3.0 V, a new
on-time cycle is initiated. The value of the off-time is calculated
using the continuous-mode operating frequency. Assuming a
nominal operating frequency (fNOM) of 200 kHz at an output
voltage of 1.7 V, the corresponding off-time is:
1.7V × 3.3 µs
L =
= 1.4 µH
4 A
A 1.5 µH inductor can be used, which gives a calculated ripple
current of 3.8 A at no load. The inductor should not saturate at
the peak current of 17 A and should be able to handle the sum
of the power dissipation caused by the average current of 15 A
in the winding and the core loss.
Designing an Inductor
VOUT
VIN
1
fNOM
Once the inductance is known, the next step is either to design an
inductor or find a standard inductor that comes as close as
possible to meeting the overall design goals. The first decision
in designing the inductor is to choose the core material. There
are several possibilities for providing low core loss at high frequen-
tOFF = 1–
×
(1)
1.7V
5V
1
tOFF = 1−
×
= 3.3 µs
200 kHz
®
cies. Two examples are the powder cores (e.g., Kool-Mµ from
The timing capacitor can be calculated from the equation:
Magnetics, Inc.) and the gapped soft ferrite cores (e.g., 3F3 or 3F4
from Philips). Low frequency powdered iron cores should be
avoided due to their high core loss, especially when the inductor
value is relatively low and the ripple current is high.
tOFF × ICT 3.3 µs ×150 µA
VT(TH )
CT =
=
≈ 150 pF
(2)
3V
Two main core types can be used in this application. Open
magnetic loop types, such as beads, beads on leads, and rods
and slugs, provide lower cost but do not have a focused mag-
netic field in the core. The radiated EMI from the distributed
magnetic field may create problems with noise interference in
the circuitry surrounding the inductor. Closed-loop types, such
as pot cores, PQ, U, and E cores, or toroids, cost more, but
have much better EMI/RFI performance. A good compromise
between price and performance are cores with a toroidal shape.
The converter only operates at the nominal operating frequency
at the above-specified VOUT and at light load. At higher values
of VOUT, or under heavy load, the operating frequency decreases
due to the parasitic voltage drops across the power devices. The
actual minimum frequency at VOUT = 1.7 V is calculated to be
195 kHz (see Equation 3), where:
R
DS(ON)HSF is the resistance of the high-side MOSFET
(estimated value: 14 mΩ)
R
DS(ON)LSF is the resistance of the low-side MOSFET
(estimated value: 6 mΩ)
VIN – IO(MAX ) ×(RDS(ON )HSF + RSENSE + RL ) –VOUT
tOFF VIN – IO(MAX ) ×(RDS(ON )HSF + RSENSE + RL – RDS(ON )LSF )
1
fMIN
=
×
(3)
)
REV. A
–7–
ADP3159/ADP3179
There are many useful references for quickly designing a power
inductor. Table II gives some examples.
The output ripple voltage can be factored into the calculation by
summing the output ripple current with the maximum output
current to determine an effective maximum dynamic current
change. The remaining errors are summed separately according
to the formula:
Table II. Magnetics Design References
Magnetic Designer Software
Intusoft (http://www.intusoft.com)
VWIN = (V∆ –VVID × 2 kVID )×
(5)
Designing Magnetic Components for High-Frequency DC-DC
Converters
IO
IO + IO∆
kCSF 2
2
2
2
2
1–
kRCS
+
+ kRT + kEA = 95 mV
McLyman, Kg Magnetics
ISBN 1-883107-00-08
where kVID = 0.5% is the initial programmed voltage tolerance
from the graph of TPC 6, kRCS = 2% is the tolerance of the
current sense resistor, kCSF = 10% is the summed tolerance of
the current sense filter components, kRT = 2% is the tolerance of
the two termination resistors added at the COMP pin, and
Selecting a Standard Inductor
The companies listed in Table III can provide design consul-
tation and deliver power inductors optimized for high power
applications upon request.
Table III. Power Inductor Manufacturers
k
EA = 8% accounts for the IC current loop gain tolerance
including the gm tolerance.
Coilcraft
(847) 639-6400
http://www.coilcraft.com
Coiltronics
(561) 752-5000
http://www.coiltronics.com
The remaining window is then divided by the maximum output
current plus the ripple to determine the maximum allowed ESR
and output resistance:
VWIN
95 mV
RE(MAX ) = ROUT(MAX )
=
=
= 5 mΩ
(6)
IO + IO∆ 15 A + 3.8 A
The output filter capacitor bank must have an ESR of less than
5 mΩ. One can, for example, use five ZA series capacitors from
Rubycon which would give an ESR of 4.8 mΩ. Without ADOPT
voltage positioning, the ESR would need to be less than 3 mΩ,
yielding a 50% increase to eight Rubycon output capacitors.
Sumida Electric Company
(408) 982-9660
http://www.sumida.com
COUT Selection—Determining the ESR
The required equivalent series resistance (ESR) and capacitance
drive the selection of the type and quantity of the output capaci-
tors. The ESR must be small enough to contain the voltage
deviation caused by a maximum allowable CPU transient cur-
rent within the specified voltage limits, giving consideration also
to the output ripple and the regulation tolerance. The capaci-
tance must be large enough that the voltage across the capacitor,
which is the sum of the resistive and capacitive voltage deviations,
does not deviate beyond the initial resistive deviation while the
inductor current ramps up or down to the value corresponding
to the new load current. The maximum allowed ESR also repre-
COUT—Checking the Capacitance
As long as the capacitance of the output capacitor is above a
critical value and the regulating loop is compensated with ADOPT,
the actual value has no influence on the peak-to-peak deviation
of the output voltage to a full step change in the load current.
The critical capacitance can be calculated as follows:
IO
COUT(CRIT )
=
× L
RE ×VOUT
×1.5 µH = 2.6 mF
(7)
15 A
5 mΩ ×1.7
=
sents the maximum allowed output resistance, ROUT
.
The critical capacitance for the five ZA series Rubycon capaci-
tors is 2.6 mF while the equivalent capacitance is 5 mF. The
capacitance is safely above the critical value.
The cumulative errors in the output voltage regulation cuts into
the available regulation window, VWIN. When considering dynamic
load regulation this relates directly to the ESR. When consider-
ing dc load regulation, this relates directly to the programmed
output resistance of the power converter.
RSENSE
The value of RSENSE is based on the maximum required output
current. The current comparator of the ADP3159 has a mini-
mum current limit threshold of 69 mV. Note that the 69 mV
value cannot be used for the maximum specified nominal cur-
rent, as headroom is needed for ripple current and tolerances.
Some error sources, such as initial voltage accuracy and ripple
voltage, can be directly deducted from the available regulation
window, while other error sources scale proportionally to the
amount of voltage positioning used, which, for an optimal design,
should utilize the maximum that the regulation window will allow.
The error determination is a closed-loop calculation, but it can
be closely approximated. To maintain a conservative design while
avoiding an impractical design, various error sources should
be considered and summed statistically.
–8–
REV. A
ADP3159/ADP3179
The current comparator threshold sets the peak of the inductor
current yielding a maximum output current, IO, which equals
twice the peak inductor current value less half of the peak-to-
peak inductor ripple current. From this the maximum value of
RSENSE is calculated as:
The maximum duty ratio of the low-side (synchronous rectifier)
MOSFET is:
(13)
DLSF(MAX ) = 1– DHSF(MAX ) = 54%
The maximum rms current of the high-side MOSFET is:
2
2
IL(VALLEY ) +(IL(VALLEY ) × IL(PEAK ) )+ IL(PEAK )
VCS(CL)(MIN )
69 mV
15 A +1.9 A
IRMSHSF
IRMSHSF
=
=
DHSF(MAX ) ×
RSENSE
≤
=
= 4 mΩ
3
IL(RIPPLE )
(8)
(14)
IO
+
13.1 A2 +(13.1 A ×16.1 A)+16.1 A2
2
36% ×
= 8.8 A rms
3
In this case, 4 mΩ was chosen as the closest standard value.
The maximum rms current of the low-side MOSFET is:
Once RSENSE has been chosen, the output current at the point
where current limit is reached, IOUT(CL), can be calculated using
the maximum current sense threshold of 87 mV:
2
2
IL(VALLEY ) + IL(VALLEY ) × IL(PEAK ) + IL(PEAK )
IRMSLSF
IRMSLSF
=
=
DLSF(MAX ) ×
3
(15)
13.1 A2 +(13.1 A ×16.1 A)+16.1 A2
VCS(CL)(MAX ) IL(RIPPLE )
54% ×
= 10.8 A rms
IOUT(CL)
=
–
3
RSENSE
87 mV 3.8 A
2
The RDS(ON) for each MOSFET can be derived from the allowable
dissipation. If 10% of the maximum output power is allowed for
MOSFET dissipation, the total dissipation will be:
(9)
=
–
≈ 20 A
4 mΩ
2
At output voltages below 450 mV, the current sense threshold is
reduced to 54 mV, and the ripple current is negligible. There-
fore, at dead short the output current is reduced to:
(16)
PD(FETs) = 0.1×VOUT × IOUT(MAX ) = 2.26W
Allocating half of the total dissipation for the high-side MOSFET
and half for the low-side MOSFET and assuming that switching
losses are small relative to the dc conduction losses, the required
minimum MOSFET resistances will be:
54 mV
IOUT(SC)
=
= 13.5 A
(10)
4 mΩ
PHSF
1.13W
8.8A2
To safely carry the current under maximum load conditions, the
sense resistor must have a power rating of at least:
RDS(ON )HSF
≤
≤
=
=
= 15 mΩ
= 10 mΩ
(17)
2
IHSF
(11)
PR
= (IO )2 × RSENSE = (20 A)2 × 4 mΩ = 1.6W
SENSE
PLSF
1.13W
10.8 A2
RDS(ON )LSF
(18)
2
Power MOSFETs
ILSF
Two external N-channel power MOSFETs must be selected for
use with the ADP3159, one for the main switch and an identical
one for the synchronous switch. The main selection parameters
for the power MOSFETs are the threshold voltage (VGS(TH)) and
the ON-resistance (RDS(ON)).
Note that there is a trade-off between converter efficiency and
cost. Larger MOSFETs reduce the conduction losses and allow
higher efficiency, but increase the system cost. If efficiency is not a
major concern, a Vishay-Siliconix SUB45N03-13L (RDS(ON)
10 mΩ nominal, 16 mΩ worst-case) for the high-side and a
Vishay-Siliconix SUB75N03-07 (RDS(ON) = 6 mΩ nominal,
10 mΩ worst-case) for the low-side are good choices.
=
The minimum input voltage dictates whether standard threshold
or logic-level threshold MOSFETs must be used. For VIN > 8 V,
standard threshold MOSFETs (VGS(TH) < 4 V) may be used. If
V
IN is expected to drop below 8 V, logic-level threshold MOSFETs
The high-side MOSFET dissipation is:
(VGS(TH) < 2.5 V) are strongly recommended. Only logic-level
MOSFETs with VGS ratings higher than the absolute maximum
value of VCC should be used.
VIN × IL(PEAK ) × QG × fMIN
2
PDHSF = IRMSHSF × RDS(ON )
+
2 × IG
(19)
5V ×15 A × 70 nC ×195 kHz
2 ×1 A
PDHSF = 8.8 A2 ×16 mΩ +
= 1.75W
The maximum output current IO(MAX) determines the RDS(ON)
requirement for the two power MOSFETs. When the ADP3159
is operating in continuous mode, the simplifying assumption can
be made that one of the two MOSFETs is always conducting
the average load current. For VIN = 5 V and VOUT = 1.65 V, the
maximum duty ratio of the high-side FET is:
where the second term represents the turn-off loss of the
MOSFET. In the second term, QG is the gate charge to be removed
from the gate for turn-off and IG is the gate current. From the
data sheet, QG is 70 nC and the gate drive current provided by
the ADP3159 is about 1 A.
DHSF(MAX ) = 1–( fMIN × tOFF
)
The low-side MOSFET dissipation is:
(12)
2
DHSF(MAX ) = 1– 195 kHz × 3.3 µs = 36%
(
)
PDLSF = IRMSLSF × RDS(ON )
(20)
PDLSF = 10.8 A2 ×10 mΩ = 1.08W
Note that there are no switching losses in the low-side MOSFET.
REV. A
–9–
ADP3159/ADP3179
Surface mount MOSFETs are preferred in CPU core converter
applications due to their ability to be handled by automatic
assembly equipment. The TO-263 package offers the power
handling of a TO-220 in a surface-mount package. However,
this package still needs adequate copper area on the PCB to
help move the heat away from the package.
and this will produce an output voltage deviation equal to the
ESR of the output capacitor array times the load current change.
TRIG'D
TEK RUN: 200kS/s SAMPLE
The junction temperature for a given area of 2-ounce copper
can be approximated using:
T = θ × PD +T
(21)
(
)
J
JA
A
assuming:
θ
θ
θ
JA = 45°C/W for 0.5 in2
JA = 36°C/W for 1 in2
JA = 28°C/W for 2 in2
2
For 1 in2 of copper area attached to each transistor and an
ambient temperature of 50°C:
T
T
JHSF = (36°C/W × 1.48 W) + 50°C = 103°C
JLSF = (36°C/W × 1.08 W ) + 50°C = 89°C
CH1
100mV
CH2
M 250ꢂs
CH2
680mV
Figure 4. Transient Response of the Circuit of Figure 3
All of the above-calculated junction temperatures are safely
below the 175°C maximum specified junction temperature of
the selected MOSFETs.
100
90
80
70
60
50
CIN Selection and Input Current di/dt Reduction
In continuous inductor-current mode, the source current of the
high-side MOSFET is approximately a square wave with a duty
ratio equal to VOUT/VIN and an amplitude of one-half of the
maximum output current. To prevent large voltage transients, a
low ESR input capacitor sized for the maximum rms current
must be used. The maximum rms capacitor current is given by:
40
30
2
IC(RMS ) = IO DHSF − DHSF
=
20
10
0
(22)
15 A 0.36 – 0.362 = 7.2 A
0
2
4
6
8
10
12
14
16
18
20
OUTPUT CURRENT – A
For a ZA-type capacitor with 1000 µF capacitance and 6.3 V
voltage rating, the ESR is 24 mΩ and the maximum allowable
ripple current at 100 kHz is 2 A. At 105°C, at least four such
capacitors must be connected in parallel to handle the calculated
ripple current. At 50°C ambient, however, a higher ripple cur-
rent can be tolerated, so three capacitors in parallel are adequate.
Figure 5. Efficiency vs. Load Current of the Circuit
of Figure 3
To correctly implement active voltage positioning, the low fre-
quency output impedance (i.e., the output resistance) of the
converter should be made equal to the maximum ESR of the
output capacitor array. This can be achieved by having a single-
pole roll-off of the voltage gain of the gm error amplifier, where
the pole frequency coincides with the ESR zero of the output
capacitor. A gain with single-pole roll-off requires that the gm
amplifier output pin be terminated by the parallel combination
of a resistor and capacitor. The required resistor value can be
calculated from the equation:
The ripple voltage across the three paralleled capacitors is:
ESRC(IN )
DHSF
nC × CIN × fMAX
VC(IN )RIPPLE = IO
×
+
nC
(23)
24 mΩ
3
36%
VC(IN )RIPPLE = 15 A ×
+
= 129 mV
3 ×1000 µF ×195 kHz
To further reduce the effect of the ripple voltage on the system
supply voltage bus, and to reduce the input-current di/dt to
below the recommended maximum of 0.1 A/ms, an additional
small inductor (L > 1 µH @ 10 A) should be inserted between
the converter and the supply bus.
ROGM × RTOTAL 1MΩ × 9.1kΩ
RCOMP
where:
RTOTAL
=
=
= 9.2 kΩ
(24)
(25)
ROGM – RTOTAL 1MΩ – 9.1kΩ
nI × RSENSE 25 × 4 mΩ
=
=
= 9.1kΩ
Feedback Compensation for Active Voltage Positioning
Optimized compensation of the ADP3159 allows the best pos-
sible containment of the peak-to-peak output voltage deviation.
Any practical switching power converter is inherently limited by
the inductor in its output current slew rate to a value much less
than the slew rate of the load. Therefore, any sudden change of
load current will initially flow through the output capacitors,
gm × RE(MAX ) 2.2 mmho × 5 mΩ
In Equations 24 and 25, ROGM is the internal resistance of the
m amplifier, nI is the division ratio from the output voltage to
signal of the gm amplifier to the PWM comparator, and gm is the
g
transconductance of the gm amplifier itself.
–10–
REV. A
ADP3159/ADP3179
Although a single termination resistor equal to RCOMP would
yield the proper voltage positioning gain, the dc biasing of that
resistor would determine how the regulation band is centered
(i.e., offset). Note that sometimes the specified regulation band
is asymmetrical with respect to the nominal VID voltage. With
the ADP3159, the offset is already considered part of the design
procedure—no special provision is required. To accomplish the
dc biasing, it is simplest to use two resistors to terminate the gm
amplifier output, with the lower resistor (RB) tied to ground and
the upper resistor (RA) to the 12 V supply of the IC. The values
of these resistors can be calculated using:
heavy loads where the voltage is “positioned” near one of the
extremes of the regulation window rather than near the nominal
center value. It must be noted and understood that this low gain
characteristic (i.e., loose dc load regulation) is inherently required
to allow improved transient containment (i.e., to achieve tighter
ac load regulation). That is, the dc load regulation is intentionally
sacrificed (but kept within specification) in order to minimize
the number of capacitors required to contain the load transients
produced by the CPU.
3.3V
ADP3159/ADP3179
1ꢂF
VDIV
12V
RA
=
=
= 79.1kΩ
(26)
gm ×(VOUT(OS) + K) 2.2 mmho ×(22 mV + 4.7 ×10–2
)
V
where K is a constant determined by internal characteristics of
the ADP3159, peak-to-peak inductor current ripple (IRIPPLE),
and the current sampling resistor (RSENSE). K can be calculated
using Equations 28 and 29. VDIV is the resistor divider supply
voltage (e.g., the recommended 12 V supply) and VOUT(OS) is
the output voltage offset from the nominal VID-programmed
value under no load condition. This offset is given by Equation 30.
LR2
1kꢃ
LRDRV1
2.5V, 2.2A
R
S
68pF
250mꢃ
LRFB1
2.5V
10kꢃ
100ꢂF
Figure 6. Adding Overcurrent Protection to the
Linear Regulator
The closest 1% value for RA is 78.7 kΩ. This value is then used
to solve for RB:
Linear Regulators
The two linear regulators provide a low cost, convenient and
versatile solution for generating additional supply rails. The
maximum output load current is determined by the size and
thermal impedance of the external N-channel power MOSFET
that is placed in series with the supply. The output voltage is
sensed at the LRFB pin and compared to an internal reference
voltage in a negative feedback loop which keeps the output voltage
in regulation. If the load is reduced or increased, the MOSFET
drive will also be reduced or increased by the controller IC to
provide a well-regulated 2.5% accurate output voltage.
RA × RCOMP 78.7 kΩ × 9.2 kΩ
RA – RCOMP 78.7 kΩ – 9.2 kΩ
RB
=
=
= 10.4 kΩ
(27)
(28)
The nearest 1% value of 10.5 kΩ was chosen for RB.
IL(RIPPLE )
(RSENSE × nI )
gm × RTOTAL
VGNL
VCC
K =
×
+
–
2
gm × RTOTAL 2 × gm ROGM
3.8 A
2
= 4.7 ×10–2
4 mΩ × 25
2.2 mmho × 9.1kΩ
1.174
12V
K =
×
+
−
2.2 mmho × 9.1kΩ 2 × 2.2 mmho ×130 kΩ
The LRFB threshholds of the ADP3159 are internally set at
2.5 V(LRFB1) and 1.8 V(LRFB2), while the LRFB pins of the
ADP3179 are compared to an internal 1 V reference. This allows
the use of an external resistor divider network to program the
linear regulator output voltage. The correct resistor values for
setting the output voltage of the linear regulators in the
ADP3179 can be determined using:
IL(RIPPLE ) × RSENSE × nI
VIN −VVID
VGNL =VGNLO
VGNL = 1V +
+
−
× tD × RSENSE × nI
2
L
(29)
(30)
3.8 A × 4 mΩ × 25
5V –1.7V
1.5 µH
−
× 75 ns × 4 mΩ × 25 = 1.174V
2
RE(MAX ) × IL(RIPPLE )
VOUT(OS) = VOUT(MAX ) −VVID
−
−VVID × kVID
(
)
2
RU + RL
5 mΩ × 3.8 A
VOUT(LR) =VLRFB
×
VOUT(OS) = 40 mV −
−1.7V × 5 ×10–3 = 22 mV
(32)
RL
2
Assuming that RL =10 kΩ, VOUT(LR) = 1.2 V and rearranging
Finally, the compensating capacitance is determined from the
equality of the pole frequency of the error amplifier gain and the
zero frequency of the impedance of the output capacitor:
equation 32 to solve for RU yields:
10 kΩ × VOUT(LR) −VLRFB
(
)
(33)
RU =
RU =
COUT × ESR 5 mF × 4.8 mΩ
VLRFB
COC
=
=
= 2.6 nF
(31)
RTOTAL
9.1kΩ
10 kΩ × 1.2V −1V
(
)
= 2 kΩ
The closest standard value for COC is 2.7 nF
1V
Trade-Offs Between DC Load Regulation and AC Load
Regulation
Casual observation of the circuit operation—e.g., with a voltmeter
—would make it appear that the dc load regulation appears to
be rather poor compared to a conventional regulator (see Figure
4). This would be especially noticeable under very light or very
REV. A
–11–
ADP3159/ADP3179
Efficiency of the Linear Regulators
The efficiency and corresponding power dissipation of each
of the linear regulators are not determined by the controller
IC. Rather, these are a function of input and output voltage and
load current. Efficiency is approximated by the formula:
LAYOUT AND COMPONENT PLACEMENT GUIDELINES
The following guidelines are recommended for optimal perfor-
mance of a switching regulator in a PC system:
General Recommendations
1. For best results, a four-layer PCB is recommended. This
should allow the needed versatility for control circuitry
interconnections with optimal placement, a signal ground
plane, power planes for both power ground and the input
power (e.g., 5 V), and wide interconnection traces in the
rest of the power delivery current paths.
VOUT
VIN
η = 100% ×
(34)
The corresponding power dissipation in the MOSFET, together
with any resistance added in series from input to output, is
given by:
2. Whenever high currents must be routed between PCB
layers, vias should be used liberally to create several parallel
current paths so that the resistance and inductance introduced
by these current paths is minimized and the via current
rating is not exceeded.
(35)
PLDO = (VIN –VOUT )× IOUT
Minimum power dissipation and maximum efficiency are accom-
plished by choosing the lowest available input voltage that exceeds
the desired output voltage. However, if the chosen input source
is itself generated by a linear regulator, its power dissipation will
be increased in proportion to the additional current it must
now provide.
3. If critical signal lines (including the voltage and current
sense lines of the controller IC) must cross through
power circuitry, it is best if a ground plane can be inter-
posed between those signal lines and the traces of the
power circuitry. This serves as a shield to minimize noise
injection into the signals at the cost of making signal
ground a bit noisier.
Implementing Current Limit for the Linear Regulators
The circuit of Figure 6 gives an example of a current limit pro-
tection circuit that can be used in conjunction with the linear
regulators. The output voltage is internally set by the LRFB pin.
The value of the current sense resistor may be calculated as
follows:
4. The GND pin of the controller IC should connect first to
a ceramic bypass capacitor (on the VCC pin) and then into
the power ground plane. However, the ground plane should
not extend under other signal components, including the
ADP3159 itself.
540 mV 540 mV
RS ≅
=
= 250 mΩ
(36)
IO(MAX )
2.2 A
The power rating of the current sense resistor must be at least:
5. The output capacitors should also be connected as closely
as possible to the load (or connector) that receives the
power (e.g., a microprocessor core). If the load is distributed,
the capacitors should also be distributed, and generally in
proportion to where the load tends to be more dynamic. It
is also advised to keep the planar interconnection path short
(i.e., have input and output capacitors close together).
2
(37)
PD(R = RS × IO(MAX) = 1.2W
)
S
The maximum linear regulator MOSFET junction temperature
with a shorted output is:
TJ(MAX ) = TA +(θJC ×VIN × IO(MAX )
TJ(MAX ) = 50°C + (1.4°C/W ×(3.3V × 2.2 A) = 60°C
)
6. Absolutely avoid crossing any signal lines over the switching
power path loop, described below.
(38)
which is within the maximum allowed by the MOSFET’s data
sheet specification. The maximum MOSFET junction tempera-
ture at nominal output is:
Power Circuitry
7. The switching power path should be routed on the PCB to
encompass the smallest possible area in order to minimize
radiated switching noise energy (i.e., EMI). Failure to take
proper precaution often results in EMI problems for the
entire PC system as well as noise-related operational prob-
lems in the power converter control circuitry. The switching
power path is the loop formed by the current path through
the input capacitors, the two FETs, and the power Schottky
diode, if used, including all interconnecting PCB traces and
planes. The use of short and wide interconnection traces is
especially critical in this path for two reasons: it minimizes
the inductance in the switching loop, which can cause high-
energy ringing, and it accommodates the high current demand
with minimal voltage loss.
TJ(NOM) = TA +(θJC ×(VIN –VOUT )× IO(NOM)
TJ(NOM) = 50°C + (1.4°C/W ×(3.3V – 2.5 V )× 2 A) = 52°C
)
(39)
This example assumes an infinite heatsink. The practical limita-
tion will be based on the actual heatsink used.
–12–
REV. A
ADP3159/ADP3179
8. A power Schottky diode (1 ~ 2 A dc rating) placed from the
lower MOSFET’s source (anode) to drain (cathode) will
help to minimize switching power dissipation in the upper
MOSFET. In the absence of an effective Schottky diode,
this dissipation occurs through the following sequence of
switching events. The lower MOSFET turns off in advance
of the upper MOSFET turning on (necessary to prevent
cross-conduction). The circulating current in the power
converter, no longer finding a path for current through the
channel of the lower MOSFET, draws current through the
inherent body-drain diode of the MOSFET. The upper
MOSFET turns on, and the reverse recovery characteristic
of the lower MOSFET’s body-drain diode prevents the drain
voltage from being pulled high quickly. The upper MOSFET
then conducts very large current while it momentarily has a
high voltage forced across it, which translates into added
power dissipation in the upper MOSFET. The Schottky diode
minimizes this problem by carrying a majority of the circu-
lating current when the lower MOSFET is turned off, and
by virtue of its essentially nonexistent reverse recovery time.
10. The output power path, though not as critical as the switch-
ing power path, should also be routed to encompass a small
area. The output power path is formed by the current path
through the inductor, the current sensing resistor, the out-
put capacitors, and back to the input capacitors.
11. For best EMI containment, the ground plane should extend
fully under all the power components. These are: the input
capacitors, the power MOSFETs and Schottky diode, the
inductor, the current sense resistor, any snubbing elements
that might be added to dampen ringing, and the output
capacitors.
Signal Circuitry
12. The output voltage is sensed and regulated between the
GND pin (which connects to the signal ground plane) and
the CS– pin. The output current is sensed (as a voltage) and
regulated between the CS– pin and the CS+ pin. In order to
avoid differential mode noise pickup in those sensed signals,
their loop areas should be small. Thus the CS– trace should
be routed atop the signal ground plane, and the CS+ and
CS– traces should be routed as a closely coupled pair (CS+
should be over the signal ground plane as well).
9. Whenever a power-dissipating component (e.g., a power
MOSFET) is soldered to a PCB, the liberal use of vias,
both directly on the mounting pad and immediately sur-
rounding it, is recommended. Two important reasons for
this are: improved current rating through the vias (if it is
a current path), and improved thermal performance—
especially if the vias extend to the opposite side of the PCB
where a plane can more readily transfer the heat to the air.
13. The CS+ and CS– traces should be Kelvin-connected to
the current sense resistor so that the additional voltage drop
due to current flow on the PCB at the current sense resistor
connections does not affect the sensed voltage. It is desir-
able to have the ADP3159 close to the output capacitor
bank and not in the output power path, so that any voltage
drop between the output capacitors and the GND pin is
minimized, and voltage regulation is not compromised.
REV. A
–13–
ADP3159/ADP3179
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
20-Lead TSSOP
(RU-20)
0.260 (6.60)
0.252 (6.40)
20
11
0.177 (4.50)
0.169 (4.30)
0.256 (6.50)
0.246 (6.25)
1
10
PIN 1
0.006 (0.15)
0.002 (0.05)
0.0433 (1.10)
MAX
8ꢁ
0ꢁ
0.0256 (0.65)
BSC
0.0118 (0.30)
0.0075 (0.19)
0.028 (0.70)
0.020 (0.50)
SEATING
PLANE
0.0079 (0.20)
0.0035 (0.090)
–14–
REV. A
ADP3159/ADP3179
Revision History
Location
Page
Global change from ADP3159 to ADP3159/ADP3179
Data Sheet changed from REV. 0 to REV. A.
Edits to GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Edits to FUNCTIONAL BLOCK DIAGRAM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Addition to LINEAR REGULATORS section of the SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
Edit to ERROR AMPLIFIER section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
Addition to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Edits to the On-board Linear Regulator Controllers section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Edits to Figure 3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Edits to Equation 24 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Edits to Feedback Compensation for Active Voltage Positioning section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Edit to Equation 27 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Addition of new text to Linear Regulators section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
REV. A
–15–
–16–
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