MP28313CS-LF [MPS]
Switching Regulator, Current-mode, 3.4A, 340kHz Switching Freq-Max, PDSO8, ROHS COMPLIANT, MS-012AA, SOIC-8;
| 型号： | MP28313CS-LF |
| 厂家： | 
MONOLITHIC POWER SYSTEMS |
| 描述： | Switching Regulator, Current-mode, 3.4A, 340kHz Switching Freq-Max, PDSO8, ROHS COMPLIANT, MS-012AA, SOIC-8 开关 光电二极管 |
| 文件： | 总11页 (文件大小：331K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MP28313
2A, 16V, 340KHz Synchronous Rectified
Step-Down Converter
The Future of Analog IC Technology
DESCRIPTION
FEATURES
The MP28313 is a monolithic synchronous buck
regulator. The device integrates 130mΩ
MOSFETS that provide 2A continuous load
current over a wide operating input voltage of
5V to 16V. Current mode control provides fast
transient response and cycle-by-cycle current
limit.
•
•
•
•
•
•
•
•
•
•
2A Output Current
Wide 5V to 16V Operating Input Range
Integrated 130mΩ Power MOSFET Switches
Output Adjustable from 0.923V to 13V
Up to 93% Efficiency
Programmable Soft-Start
Stable with Low ESR Ceramic Output Capacitors
Fixed 340KHz Frequency
An adjustable soft-start prevents inrush current
at turn-on. Shutdown mode drops the supply
current to 1µA.
Cycle-by-Cycle Over Current Protection
Input Under Voltage Lockout
This device, available in an 8-pin SOIC
package, provides a very compact system
solution with minimal reliance on external
components.
APPLICATIONS
•
•
•
•
•
Distributed Power Systems
Networking Systems
FPGA, DSP, ASIC Power Supplies
Green Electronics/ Appliances
Notebook Computers
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
TYPICAL APPLICATION
C5
10nF
Efficiency vs
Load Current
INPUT
5V to 16V
100
95
90
85
80
75
70
65
60
55
50
V
= 3.3V
OUT
2
1
IN
BS
OUTPUT
3.3V
2A
V
= 2.5V
3
5
7
8
OUT
EN
SS
SW
MP28313
FB
GND
COMP
4
6
C3
3.3nF
0
0.5
1.0
1.5
2.0
2.5
LOAD CURRENT (A)
MP28313_EC01
MP28313 Rev. 1.5
9/27/2010
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1
MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
ORDERING INFORMATION
Part Number
MP28313CS*
MP28313DS
Package
Top Marking
MP28313CS
MP28313DS
Free Air Temperature (TA)
0°C to +85°C
SOIC8
–40°C to +85°C
* For Tape & Reel, add suffix –Z (e.g. MP28313CS–Z)
For RoHS compliant packaging, add suffix –LF (e.g. MP28313CS–LF–Z)
PACKAGE REFERENCE
TOP VIEW
BS
IN
1
2
3
4
8
7
6
5
SS
EN
SW
GND
COMP
FB
MP28313_PD01
ABSOLUTE MAXIMUM RATINGS (1)
Supply Voltage VIN .......................–0.3V to +22V
Switch Voltage VSW .................. –1V to VIN +0.3V
Boost Voltage VBS ..........VSW – 0.3V to VSW + 6V
All Other Pins.................................–0.3V to +6V
Thermal Resistance (4)
SOIC8 .....................................90 ...... 45...°C/W
θJA
θJC
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ(MAX), the junction-to-
ambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD(MAX)=(TJ(MAX)-
TA)/ θJA. Exceeding the maximum allowable power dissipation
will cause excessive die temperature, and the regulator will go
into thermal shutdown. Internal thermal shutdown circuitry
protects the device from permanent damage.
Continuous Power Dissipation
(TA = +25°C)(2)
…………………………………………………1.4W
Junction Temperature...............................150°C
Lead Temperature ....................................260°C
Storage Temperature ............. –65°C to +150°C
Recommended Operating Conditions (3)
Input Voltage VIN .................................5V to 16V
Output Voltage VOUT.....................0.923V to 13V
MP28313CS Operating Junct. Temp (TJ).............
……………………………………….0°C to +85°C
MP28313DS Operating Junct. Temp (TJ).............
……………………………………...-40°C to +85°C
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7, 4-layer PCB..
MP28313 Rev. 1.5
9/27/2010
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MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, TA = +25°C, unless otherwise noted.
Parameter
Symbol Condition
Min
Typ
1
Max
10
Units
µA
Shutdown Supply Current
Supply Current
VEN = 0V
VEN = 2.0V; VFB = 1.0V
1.3
1.8
mA
V
0.909
.906
.905
0.923
0.937
.941
.941
5V ≤ VIN ≤ 16V, TA =+25°C
TA = 0°C to +85°C
Feedback Voltage
VFB
V
TA = -40°C to +85°C
Feedback Overvoltage
Threshold
1.1
400
130
V
Error Amplifier Voltage Gain (5)
AEA
V/V
mΩ
High-Side Switch On Resistance
RDS(ON)1
(5)
Low-Side Switch On Resistance
RDS(ON)2
130
mΩ
(5)
High-Side Switch Leakage
Current
VEN = 0V, VSW = 0V
Minimum Duty Cycle
10
µA
Upper Switch Current Limit
Oscillation Frequency
2.6
3.4
A
Fosc1
Fosc2
300
340
KHz
Short Circuit Oscillation
Frequency
VFB = 0V
100
KHz
Maximum Duty Cycle
DMAX VFB = 1.0V
VEN Rising
87
90
220
1.5
%
ns
V
Minimum On Time (5)
340
2.0
EN Shutdown Threshold Voltage
1.1
EN Shutdown Threshold Voltage
Hysteresis
210
mV
EN Lockout Threshold Voltage
EN Lockout Hysterisis
2.2
2.5
2.7
V
210
mV
Input Under Voltage Lockout
Threshold
VIN Rising
3.80
4.10
210
4.40
V
Input Under Voltage Lockout
Threshold Hysteresis
mV
Soft-Start Current
Soft-Start Period
Thermal Shutdown (5)
VSS = 0V
6
µA
ms
°C
CSS = 0.1µF
15
145
160
Note:
5) Guaranteed by design, not tested.
MP28313 Rev. 1.5
9/27/2010
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MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
PIN FUNCTIONS
Pin # Name Description
High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET
switch. Connect a 0.01µF or greater capacitor from SW to BS to power the high side switch.
1
2
BS
IN
Power Input. IN supplies the power to the IC, as well as the step-down converter switches.
Drive IN with a 5V to 16V power source. Bypass IN to GND with a suitably large capacitor to
eliminate noise on the input to the IC. See Input Capacitor.
Power Switching Output. SW is the switching node that supplies power to the output. Connect
the output LC filter from SW to the output load. Note that a capacitor is required from SW to
BS to power the high-side switch.
3
4
5
SW
GND Ground.
Feedback Input. FB senses the output voltage to regulate that voltage. Drive FB with a
resistive voltage divider from the output voltage. The feedback threshold is 0.923V. See
FB
Setting the Output Voltage.
Compensation Node. COMP is used to compensate the regulation control loop. Connect a
series RC network from COMP to GND to compensate the regulation control loop. In some
cases, an additional capacitor from COMP to GND is required. See Compensation
Components.
6
COMP
Enable Input. EN is a digital input that turns the regulator on or off. Drive EN high to turn on
the regulator, drive it low to turn it off. Pull up with 100kΩ resistor for automatic startup.
7
8
EN
SS
Soft-Start Control Input. SS controls the soft start period. Connect a capacitor from SS to GND
to set the soft-start period. A 0.1µF capacitor sets the soft-start period to 15ms. To disable the
soft-start feature, leave SS unconnected.
MP28313 Rev. 1.5
9/27/2010
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MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS
VIN = 12V, VO = 3.3V, L = 10µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless otherwise noted.
Shutdown through Enable
Startup through Enable
Steady State Test
V
I
= 12V, V
= 3.3V
= 1A (Resistance Load)
V
I
= 12V, V
= 3.3V
= 1A (Resistance Load)
V
I
= 12V, V
OUT
= 3.3V
IN OUT
IN
OUT
IN
= 0A, I = 8.2mA
OUT
OUT
OUT
IN
V
IN
V
20mV/div.
V
EN
5V/div.
EN
5V/div.
V
OUT
V
V
OUT
OUT
20mV/div.
2V/div.
2V/div.
I
L
1A/div.
I
L
I
L
1A/div.
1A/div.
V
SW
V
SW
V
10V/div.
SW
10V/div.
10V/div.
2ms/div.
2ms/div.
MP28313-TPC03
MP28313-TPC01
MP28313-TPC02
Light Load Operation
No Load
Heavy Load Operation
2A Load
Medium Load Operation
1A Load
V
V
IN, AC
IN, AC
V
IN, AC
200mV/div.
20mV/div.
200mV/div.
V
V
O, AC
O, AC
V
O, AC
20mV/div.
20mV/div.
20mV/div.
I
L
I
I
L
L
1A/div.
1A/div.
1A/div.
V
V
V
SW
SW
SW
10V/div.
10V/div.
10V/div.
MP28313-TPC04
MP28313-TPC05
MP28313-TPC06
Short Circuit
Recovery
Load Transient
Short Circuit
Protection
V
OUT
V
V
OUT
OUT
2V/div.
2V/div.
200mV/div.
I
L
1A/div.
I
L
2A/div.
I
L
I
2A/div.
LOAD
1A/div.
MP28313-TPC07
MP28313-TPC08
MP28313-TPC09
MP28313 Rev. 1.5
9/27/2010
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MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
OPERATION
The converter uses internal N-Channel
MOSFET switches to step-down the input
voltage to the regulated output voltage. Since
the high side MOSFET requires a gate voltage
greater than the input voltage, a boost capacitor
connected between SW and BS is needed to
drive the high side gate. The boost capacitor is
charged from the internal 5V rail when SW is low.
FUNCTIONAL DESCRIPTION
The MP28313 is a synchronous rectified,
current-mode, step-down regulator. It regulates
input voltages from 5V to 16V down to an
output voltage as low as 0.923V, and supplies
up to 2A of load current.
The MP28313 uses current-mode control to
regulate the output voltage. The output voltage
is measured at FB through a resistive voltage
divider and amplified through the internal
transconductance error amplifier. The voltage at
the COMP pin is compared to the switch current
measured internally to control the output
voltage.
When the MP28313 FB pin exceeds 20% of the
nominal regulation voltage of 0.923V, the over
voltage comparator is tripped and the COMP
pin and the SS pin are discharged to GND,
forcing the high-side switch off.
+
CURRENT
2
IN
OVP
SENSE
AMPLIFIER
+
--
--
+
1.1V
0.3V
5V
RAMP
CLK
OSCILLATOR
100/340KHz
5
FB
1
3
BS
--
--
+
S
Q
Q
--
+
+
SW
R
CURRENT
COMPARATOR
8
6
SS
ERROR
AMPLIFIER
0.923V
COMP
4
GND
OVP
1.2V
+
--
2.5V
EN
IN
IN < 4.10V
EN OK
LOCKOUT
COMPARATOR
7
+
--
EN
INTERNAL
REGULATORS
5V
SHUTDOWN
COMPARATOR
1.5V
MP28313_F01_BD01
Figure 1—Functional Block Diagram
MP28313 Rev. 1.5
9/27/2010
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MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
APPLICATIONS INFORMATION
COMPONENT SELECTION
INPUT
5V to 16V
2
1
IN
BS
OUTPUT
3.3V
2A
3
5
7
8
EN
SS
SW
MP28313
FB
GND
COMP
4
6
Figure 2—Application Circuit
Setting the Output Voltage
determining the inductance to use is to allow
the peak-to-peak ripple current in the inductor
to be approximately 30% of the maximum
switch current limit. Also, make sure that the
peak inductor current is below the maximum
switch current limit. The inductance value can
be calculated by:
The output voltage is set using a resistive
voltage divider from the output voltage to FB pin.
The voltage divider divides the output voltage
down to the feedback voltage by the ratio:
R2
VFB = VOUT
R1+ R2
⎛
⎜
⎝
⎞
⎟
⎟
VOUT
VOUT
VIN
Where VFB is the feedback voltage and VOUT is
the output voltage.
⎜
L =
× 1−
fS × ∆IL
⎠
Thus the output voltage is:
Where VOUT is the output voltage, VIN is the
input voltage, fS is the switching frequency, and
∆IL is the peak-to-peak inductor ripple current.
R1+ R2
VOUT = 0.923 ×
R2
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
R2 can be as high as 100kΩ, but a typical value
is 10kΩ. Using the typical value for R2, R1 is
determined by:
⎛
⎜
⎝
⎞
VOUT
VOUT
VIN
R1 = 10.83 × (VOUT − 0.923) (kꢀ)
⎜
⎟
⎟
⎠
ILP = ILOAD
+
× 1−
2× fS ×L
For example, for a 3.3V output voltage, R2 is
10kꢀ, and R1 is 26.1kꢀ.
Where ILOAD is the load current.
The choice of which style inductor to use mainly
depends on the price vs. size requirements and
any EMI requirements.
Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor will result in less ripple current that will
result in lower output ripple voltage. However,
the larger value inductor will have a larger
physical size, higher series resistance, and/or
lower saturation current. A good rule for
MP28313 Rev. 1.5
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MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
⎛
⎜
⎝
⎞
⎟
⎟
⎠
ILOAD
VOUT
VIN
VOUT
Optional Schottky Diode
⎜
∆V
=
×
× 1−
IN
C1× fS
V
IN
During the transition between high-side switch
and low-side switch, the body diode of the low-
side power MOSFET conducts the inductor
current. The forward voltage of this body diode
is high. An optional Schottky diode may be
paralleled between the SW pin and GND pin to
improve overall efficiency. Table 1 lists example
Schottky diodes and their Manufacturers.
Where C1 is the input capacitance value.
Output Capacitor
The output capacitor is required to maintain the
DC output voltage. Ceramic, tantalum, or low
ESR electrolytic capacitors are recommended.
Low ESR capacitors are preferred to keep the
output voltage ripple low. The output voltage
ripple can be estimated by:
Table 1—Diode Selection Guide
Voltage/Current
⎛
⎜
⎝
⎞
⎟
⎟
⎠
⎛
⎜
⎝
⎞
⎟
⎟
⎠
VOUT
VOUT
VIN
1
Part Number
B130
Rating
30V, 1A
30V, 1A
30V, 1A
Vendor
⎜
⎜
∆VOUT
=
× 1−
× RESR
+
fS × L
8 × fS × C2
Diodes, Inc.
Diodes, Inc.
SK13
Where C2 is the output capacitance value and
RESR is the equivalent series resistance (ESR)
value of the output capacitor.
MBRS130
International
Rectifier
In the case of ceramic capacitors, the
impedance at the switching frequency is
dominated by the capacitance. The output
voltage ripple is mainly caused by the
capacitance. For simplification, the output
voltage ripple can be estimated by:
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required
to supply the AC current to the step-down
converter while maintaining the DC input
voltage. Use low ESR capacitors for the best
performance. Ceramic capacitors are preferred,
but tantalum or low-ESR electrolytic capacitors
may also suffice. Choose X5R or X7R
dielectrics when using ceramic capacitors.
⎛
⎜
⎝
⎞
VOUT
VOUT
VIN
⎜
⎟
⎟
⎠
∆VOUT
=
× 1−
2
8 × fS × L × C2
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the
output ripple can be approximated to:
Since the input capacitor (C1) absorbs the input
switching current it requires an adequate ripple
current rating. The RMS current in the input
capacitor can be estimated by:
VOUT
VOUT
VIN
⎛
⎞
∆VOUT
=
× ⎜1−
⎟ ×RESR
⎜
⎟
⎛
⎞
⎟
VOUT
VOUT
fS ×L
⎝
⎠
⎜
IC1 = ILOAD
×
× 1−
⎜
⎝
⎟
⎠
V
V
IN
IN
The characteristics of the output capacitor also
affect the stability of the regulation system. The
MP28313 can be optimized for a wide range of
capacitance and ESR values.
The worst-case condition occurs at VIN = 2VOUT
,
where IC1 = ILOAD/2. For simplification, choose
the input capacitor whose RMS current rating
greater than half of the maximum load current.
Compensation Components
MP28313 employs current mode control for
easy compensation and fast transient response.
The system stability and transient response are
controlled through the COMP pin. COMP pin is
the output of the internal transconductance
error amplifier. A series capacitor-resistor
combination sets a pole-zero combination to
control the characteristics of the control system.
The input capacitor can be electrolytic, tantalum
or ceramic. When using electrolytic or tantalum
capacitors, a small, high quality ceramic
capacitor, i.e. 0.1µF, should be placed as close
to the IC as possible. When using ceramic
capacitors, make sure that they have enough
capacitance to provide sufficient charge to
prevent excessive voltage ripple at input. The
input voltage ripple for low ESR capacitors can
be estimated by:
MP28313 Rev. 1.5
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MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
The DC gain of the voltage feedback loop is
given by:
in slower line and load transient responses,
while higher crossover frequencies could cause
system instability. A good rule of thumb is to set
the crossover frequency below one-tenth of the
switching frequency.
VFB
AVDC = RLOAD × GCS × AEA
×
VOUT
Where AVEA is the error amplifier voltage gain;
GCS is the current sense transconductance and
To optimize the compensation components, the
following procedure can be used.
RLOAD is the load resistor value.
1. Choose the compensation resistor (R3,
Figure2) to set the desired crossover frequency.
The system has two poles of importance. One
is due to the compensation capacitor (C3,
Figure2) and the output resistor of the error
amplifier, and the other is due to the output
capacitor and the load resistor. These poles are
located at:
Determine the R3 value by the following
equation:
2π × C2 × fC VOUT 2π × C2 × 0.1× fS VOUT
R3 =
×
<
×
GEA × GCS
VFB
GEA × GCS
VFB
Where fC is the desired crossover frequency
which is typically below one tenth of the
switching frequency.
GEA
fP1
=
2π× C3× AVEA
1
fP2
=
2. Choose the compensation capacitor (C3,
Figure2) to achieve the desired phase margin.
For applications with typical inductor values,
setting the compensation zero, fZ1, below one-
forth of the crossover frequency provides
sufficient phase margin.
2π × C2× RLOAD
Where
GEA
is
the
error
amplifier
transconductance.
The system has one zero of importance, due to the
compensation capacitor (C3) and the compensation
resistor (R3). This zero is located at:
Determine the C3 value by the following equation:
4
1
C3 >
fZ1
=
2π × R3 × fC
2π × C3×R3
Where R3 is the compensation resistor.
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero,
due to the ESR and capacitance of the output
capacitor, is located at:
3. Determine if the second compensation
capacitor (C6, Figure2) is required. It is required
if the ESR zero of the output capacitor is
located at less than half of the switching
frequency, or the following relationship is valid:
1
fESR
=
2π × C2× RESR
fS
2
1
<
2π × C2× RESR
In this case, a third pole set by the
compensation capacitor (C6, Figure2) and the
compensation resistor (R3) is used to
compensate the effect of the ESR zero on the
loop gain. This pole is located at:
If this is the case, then add the second
compensation capacitor (C6, Figure2) to set the
pole fP3 at the location of the ESR zero.
Determine the C6 value by the equation:
1
C2 × RESR
fP3
=
C6 =
2π× C6×R3
R3
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
where the feedback loop has the unity gain is
important. Lower crossover frequencies result
MP28313 Rev. 1.5
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MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
External Bootstrap Diode
An external bootstrap diode may enhance the
efficiency of the regulator, the applicable
conditions of external BST diode are:
z
z
VOUT=5V or 3.3V; and
Duty cycle is high: D=
VOUT
VIN
>65%
In these cases, an external BST diode is
recommended from the output of the voltage
regulator to BST pin, as shown in Figure3
External BST Diode
IN4148
BST
CBST
MP28313
5V or 3.3V
SW
+
COUT
L
Figure 3—Add Optional External Bootstrap
Diode to Enhance Efficiency
The recommended external BST diode is
IN4148, and the BST cap is 0.1~1µF.
MP28313 Rev. 1.5
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MP28313 – 2A, 16V, 340KHz SYNCHRONOUS RECTIFIED, STEP-DOWN CONVERTER
PACKAGE INFORMATION
SOIC8
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
MP28313 Rev. 1.5
9/27/2010
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2010 MPS. All Rights Reserved.
11
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