MAX17597ATE+T [MAXIM]
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| 型号： | MAX17597ATE+T |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | 暂无描述 稳压器 控制器 |
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19-6178; Rev 0; 1/12
E V A L U A T I O N K I T A V A I L A B L E
General Description
Benefits and Features
The MAX17595/MAX17596/MAX17597 is a family of peak-
current-mode controllers which contain all the circuitry
required for the design of wide input-voltage flyback
and boost regulators. The MAX17595 offers optimized
input rising and falling thresholds for universal input
AC-DC converters and telecom DC-DC (36V–72V input
range) power supplies. The MAX17596 offers input rising
and falling thresholds suitable for low-voltage DC-DC
applications (4.5V–36V input range). The MAX17597
offers all circuitry needed to implement a boost converter
controller. All three controllers contain a built-in gate
driver for external n-channel MOSFETs.
S Peak Current Mode Offline (Universal Input AC)
and Telecom (36V–72V) Flyback Controller
(MAX17595)
S Peak-Current-Mode DC-DC Flyback Controller
(4.5V–36V Input Range) (MAX17596)
S Peak-Current-Mode Nonsynchronous Boost PWM
Controller (4.5V–36V Input Range) (MAX17597)
S Current Mode Control Provides Excellent
Transient Response
S Low 20µA Startup Supply Current
S 100kHz to 1MHz Programmable Switching
The MAX17595/MAX17596/MAX17597 house an inter-
nal error amplifier with 1% accurate reference, useful
in implementations without the need for an external
reference. The switching frequency is programmable
from 100kHz to 1MHz with an accuracy of 8% using an
external resistor, allowing optimization of magnetic and
filter components, resulting in compact and cost-effective
power conversion solutions. For EMI sensitive applica-
tions, the MAX17595/MAX17596/MAX17597 family incor-
porates a programmable-frequency dithering scheme,
enabling low-EMI spread-spectrum operation.
Frequency
S Programmable Frequency Dithering for Low-EMI
Spread-Spectrum Operation
S Switching Frequency Synchronization
S Adjustable Current Limit with External Current-
Sense Resistor
S Fast Cycle-By-Cycle Peak Current Limiting
S Hiccup-Mode Short-Circuit Protection
S Overtemperature Protection
An EN/UVLO input allows the user to start the
power supply precisely at the desired input voltage,
while also functioning as an on/off pin. The OVI pin
enables implementation of an input overvoltage protection
scheme, ensuring that the converter shuts down when
the DC input voltage exceeds a set maximum value. The
SS pin allows programmable soft-start time for the power
converter, and helps limit inrush current during startup.
The MAX17595/MAX17596/MAX17597 family also allows
the designer to choose between voltage soft-start and
current soft-start modes, useful in optoisolated designs.
A programmable slope compensation scheme is pro-
vided to enhance the stability of the peak-current-mode
control scheme.
S Programmable Soft-Start and Slope Compensation
S Programmable Voltage or Current Soft-Start
Schemes
S Input Overvoltage Protection
S Space-Saving, 3mm x 3mm TQFN Package
Applications
Universal Input Offline AC-DC Power Supplies
Wide-Range DC-Input Flyback/Boost Battery
Chargers
Battery-Powered Applications
Industrial, Telecom, and Automotive Applications
Hiccup-mode overcurrent protection and thermal
shutdown are provided to minimize dissipation in
overcurrent and overtemperature fault conditions. The IC
is available in a space-saving 16-pin, 3mm x 3mm TQFN
package with 0.5mm lead spacing.
Ordering Information/Selector Guide appears at end of
data sheet.
For related parts and recommended products to use with this part,
refer to www.maxim-ic.com/MAX17595.related.
����������������������������������������������������������������� Maxim Integrated Products
1
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
ABSOLUTE MAXIMUM RATINGS
V
V
V
to SGND..........................................................-0.3V to +40V
Maximum Input/Output Current (Continuous)
V , NDRV ........................................................................100mA
IN
IN
DRV
DRV
to SGND..................................-0.3V to +16V (MAX17595)
to SGND..........-0.3V to +6V (MAX17596 and MAX17597)
NDRV (pulsed, for less than 100ns) .................................... Q1A
Continuous Power Dissipation TQFN (single-layer board)
(derate 20.8mW/NC above +70NC)............................1666mW
Operating Temperature Range........................ -40NC to +125NC
Storage Temperature Range............................ -65NC to +150NC
Junction Temperature .....................................................+150NC
Lead Temperature (soldering, 10s) ................................+300NC
Soldering Temperature (reflow) ......................................+260NC
NDRV to SGND .................................... -0.3V to +(V
EN/UVLO to SGND.................................. -0.3V to +(V + 0.3)V
OVI, RT, DITHER, COMP, SS, FB,
SLOPE to SGND .................................................... -0.3V to +6V
CS to SGND ............................................................-0.8V to +6V
PGND to SGND....................................................-0.3V to +0.3V
+ 0.3)V
DRV
IN
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional opera-
tion of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
PACKAGE THERMAL CHARACTERISTICS (Note 1)
Junction-to-Ambient Thermal Resistance (q )..............48°C/W
Junction-to-Case Thermal Resistance (q ).....................7°C/W
JA
JC
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-
layer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial.
ELECTRICAL CHARACTERISTICS
(V = 12V (for the MAX17595, bring V up to 21V for startup), V
= V
= V
= V
= V
= V
= 0V,
SGND
IN
IN
CS
SLOPE
DITHER
FB
OVI
V
= +2V; NDRV, SS, COMP are unconnected, R = 25kI, C
= 1FF, C
= 1FF, T = T = -40NC to +125NC, unless
EN/UVLO
RT
VIN
VDRV
A
J
otherwise noted. Typical values are at T = T = +25NC.) (Note 2)
A
J
PARAMETER
INPUT SUPPLY (V
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
)
IN
MAX17595
MAX17596/MAX17597
8
29
36
V
Voltage Range
V
V
V
IN
IN
4.5
18.5
3.5
6.5
3.3
MAX17595
20
4
21.5
4.4
V
V
Bootstrap UVLO Wakeup
Bootstrap UVLO Shutdown
V
IN-UVR
V
rising #
IN
IN
MAX17596/MAX17597
MAX17595
7
7.7
IN
V
V
V
V
V
falling $
< UVLO
= 0V
IN-UVF
IN
Level
MAX17596/MAX17597
3.9
4.25
V
Supply Start-Up Current
I
IN
VIN-
STARTUP
20
32
32
FA
IN
(Under UVLO)
V
V
Supply Shutdown Current
Supply Current
I
20
2
FA
IN
IN-SH
EN
I
Switching, f
= 400kHz
mA
IN
IN-SW
SW
V
CLAMP (INC) (MAX17595 ONLY)
IN
MAX17595, I
(Note 3)
= 2mA sinking, V = 0V
EN
VIN
V
Clamp Voltage
V
30
33
36
V
IN
INC
ENABLE (EN)
V
1.16
1.1
1.21
1.15
1.26
1.2
V
V
V
rising #
falling $
ENR
EN
EN
EN
EN Undervoltage Threshold
EN Input Leakage Current
V
V
ENF
I
= 1.5V, T = +25NC
-100
+100
nA
EN
A
����������������������������������������������������������������� Maxim Integrated Products
2
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
ELECTRICAL CHARACTERISTICS (continued)
(V = 12V (for the MAX17595, bring V up to 21V for startup), V
= V
= V
= V
= V
= V
= 0V,
SGND
IN
IN
CS
SLOPE
DITHER
FB
OVI
V
= +2V; NDRV, SS, COMP are unconnected, R = 25kI, C
= 1FF, C
= 1FF, T = T = -40NC to +125NC, unless
EN/UVLO
RT
VIN
VDRV
A
J
otherwise noted. Typical values are at T = T = +25NC.) (Note 2)
A
J
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
INTERNAL LDO (VDRV)
8V < V < 15V and 0mA < I
IN
(MAX17595)
< 50mA
< 50mA
VDRV
7.1
7.4
7.7
5.1
V
Output Voltage Range
V
V
DRV
DRV
6V < V < 12V and 0mA < I
IN
(MAX17596/MAX17597)
VDRV
4.7
70
4.9
V
V
Current Limit
Dropout
I
100
mA
V
DRV
VDRV-MAX
V
= 4.5V, I
= 20mA (MAX17596/
IN
VDRV
V
4.2
DRV
VDRV-DO
MAX17597)
OVERVOLTAGE PROTECTION (OVI)
V
1.16
1.1
1.21
1.15
2
1.26
1.2
V
V
rising #
OVIR
OVI
OVI Overvoltage Threshold
V
V
falling $
OVIF
OVI
OVI Masking Delay
t
Fs
OVI-MD
OVI Input Leakage Current
OSCILLATOR (RT)
I
V
= 1V, T = +25NC
-100
+100
nA
OVI
OVI
A
NDRV Switching Frequency
Range
f
100
-8
1000
+8
kHz
%
SW
NDRV Switching Frequency
Accuracy
(MAX17595/MAX17596)
(MAX17597)
46
90
48
50
95
Maximum Duty Cycle
D
%
MAX
92.5
SYNCHRONIZATION (DITHER)
Synchronization Logic-High
Input
V
3
V
HI-SYNC
Synchronization Pulse Width
50
ns
Hz
Synchronization Frequency
Range
f
(MAX17595/MAX17596)
1.1 x f
1.8 x f
SW
SYNCIN
SW
DITHERING RAMP GENERATOR (DITHER)
Charging Current
V
V
= 0V
45
43
50
50
2
55
57
FA
FA
V
DITHER
Discharging Current
= 2.2V
DITHER
Ramp-High Trip Point
Ramp-Low Trip Point
0.4
V
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3
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
ELECTRICAL CHARACTERISTICS (continued)
(V = 12V (for the MAX17595, bring V up to 21V for startup), V
= V
= V
= V
= V
= V
= 0V,
SGND
IN
IN
CS
SLOPE
DITHER
FB
OVI
V
= +2V; NDRV, SS, COMP are unconnected, R = 25kI, C
= 1FF, C
= 1FF, T = T = -40NC to +125NC, unless
EN/UVLO
RT
VIN
VDRV
A
J
otherwise noted. Typical values are at T = T = +25NC.) (Note 2)
A
J
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
SOFT-START/SOFT-STOP (SS)
Soft-Start Charging Current
I
9
10
5
11
5.6
FA
FA
V
SSCH
Soft-Stop Discharging Current
SS Bias Voltage
I
For soft-stop enabled parts
Soft-stop completion
4.4
1.19
SSDISCH
V
1.21
0.15
1.23
SS
SS Discharge Threshold
NDRV DRIVER (NDRV)
Pulldown Impedance
Pullup Impedance
Peak Sink Current
Peak Source Current
Fall Time
V
V
SSDISCH
R
I
I
(sinking) = 100mA
(sourcing) = 5mA
1.37
4.26
1.5
0.9
10
3
I
I
NDRV-N
NDRV
NDRV
R
8.5
NDRV-P
C
C
C
C
= 10nF
= 10nF
= 1nF
A
NDRV
NDRV
NDRV
NDRV
A
t
ns
ns
NDRV-F
Rise Time
t
= 1nF
20
NDRV-R
CURRENT-LIMIT COMPARATOR (CS)
Cycle-by-Cycle Peak -Current-
Limit Threshold
V
290
340
-122
305
360
-102
70
320
380
-82
mV
mV
mV
ns
CS-PEAK
Cycle-by-Cycle Runaway
Current-Limit Threshold
V
CS-RUN
Cycle-by-Cycle Reverse-
Current Limit Threshold
V
CS-REV
Current-Sense Leading-Edge
Blanking Time
t
From NDRV rising # edge
CS-BLANK
From CS rising (10mV overdrive) to
NDRV falling (excluding leading
edge blanking)
Propagation Delay from
Comparator Input to NDRV
t
40
ns
PDCS
Number of Consecutive Peak-
Current-Limit Events to Hiccup
N
N
8
1
event
event
HICCUP-P
Number of Runaway-Current-
Limit Events to Hiccup
HICCUP-R
Overcurrent Hiccup Timeout
Minimum On-Time
32768
130
cycle
ns
t
90
170
ON-MIN
SLOPE COMPENSATION (SLOPE)
Slope Bias Current
I
9
10
11
FA
kI
SLOPE
Slope Resistor Range
25
200
Slope Voltage Range to
Enable Current Soft-Start and
Minimum Slope Compensation
0
200
mV
����������������������������������������������������������������� Maxim Integrated Products
4
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
ELECTRICAL CHARACTERISTICS (continued)
(V = 12V (for the MAX17595, bring V up to 21V for startup), V
= V
= V
= V
= V
= V
= 0V,
SGND
IN
IN
CS
SLOPE
DITHER
FB
OVI
V
= +2V; NDRV, SS, COMP are unconnected, R = 25kI, C
= 1FF, C
= 1FF, T = T = -40NC to +125NC, unless
EN/UVLO
RT
VIN
VDRV
A
MIN
4
J
otherwise noted. Typical values are at T = T = +25NC.) (Note 2)
A
J
PARAMETER
SYMBOL
CONDITIONS
TYP
MAX
UNITS
Slope Voltage Range to
Enable Voltage Soft-Start and
Minimum Slope Compensation
V
Slope Voltage Range to
Enable Voltage Soft-Start
and Programmable Slope
Compensation
0.2
4
V
Slope Compensation Ramp
140
165
50
190
mV/Fs
mV/Fs
R
= 100kW
SLOPE
Default Slope Compensation
Ramp
V
< 0.2V or 4V < V
SLOPE
SLOPE
PWM COMPARATOR
Comparator Offset Voltage
Current-Sense Gain
V
V
- V
CS
1.65
1.75
1.81
1.97
13
2
V
PWM-OS
COMP
A
DCOMP/DCS (T = +25NC)
2.15
20
V/V
FA
CS-PWM
A
CS Peak Slope Ramp Current
I
Ramp current peak (T = +25NC)
A
CSSLOPE
Comparator Propagation
Delay
Change in V = 10mV (including internal
CS
lead-edge blanking)
t
110
ns
PWM
ERROR AMPLIFIER
FB Reference Voltage
FB Input Bias Current
Voltage Gain
V
V
V
, when I
= 0 and V = 1.8V
COMP
1.19
-100
1.21
1.23
V
nA
REF
FB
COMP
I
= 1.5V, T = +25NC
+100
FB
FB
A
A
EAMP
Gm
80
1.8
10
dB
mS
MHz
FA
Transconductance
1.5
2.1
Transconductance Bandwidth
Source Current
BW
Open-loop (gain = 1), -3dB frequency
V
= 1.8V, V = 1V
80
80
120
120
210
210
COMP
COMP
FB
Sink Current
V
= 1.8V, V = 1.75V
FA
FB
THERMAL SHUTDOWN
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis
Temperature rising
+160
20
NC
NC
Note 2: All devices 100% production tested at T = +25°C. Limits over temperature are guaranteed by design.
A
Note 3: The MAX17595 is intended for use in universal input power supplies. The internal clamp circuit at V is used to prevent
IN
the bootstrap capacitor from changing to a voltage beyond the absolute maximum rating of the device when EN is low
(shutdown mode). Externally limit the maximum current to V (hence to clamp) to 2mA (max) when EN is low.
IN
����������������������������������������������������������������� Maxim Integrated Products
5
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Typical Operating Characteristics
VCC A J
(V = 15V, V
= +2V, COMP = open, C
= 1FF, C
= 1FF, T = T = -40NC to +125NC, unless otherwise noted.)
IN
EN/UVLO
VIN
BOOTSTRAP UVLO WAKE-UP LEVEL
V
IN
WAKE-UP LEVEL vs. TEMPERATURE
V FALLING THRESHOLD
IN
vs. TEMPERATURE (MAX17595)
(MAX17596/MAX17597)
vs. TEMPERATURE (MAX17595)
MAX17595/6/7 toc01
MAX17595/6/7 toc02
MAX17595/6/7 toc03
20.04
20.03
20.02
20.01
20.00
19.99
19.98
4.15
4.10
4.05
4.00
3.95
3.90
7.025
7.020
7.015
7.010
7.005
7.000
6.995
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
EN/UVLO RISING THRESHOLD
vs. TEMPERATURE
V
FALLING THRESHOLD vs. TEMPERATURE
IN
(MAX17596/MAX17597)
MAX17595/6/7 toc05
MAX17595/6/7 toc04
1.209
1.208
1.207
1.206
1.205
1.204
1.203
1.202
4.00
3.95
3.90
3.85
3.80
3.75
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
EN/UVLO FALLING THRESHOLD
vs. TEMPERATURE
OVI RISING THRESHOLD
vs. TEMPERATURE
MAX17595/6/7 toc06
MAX17595/6/7 toc07
1.149
1.148
1.147
1.146
1.145
1.211
1.210
1.209
1.208
1.207
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
����������������������������������������������������������������� Maxim Integrated Products
6
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Typical Operating Characteristics (continued)
(V = 15V, V
IN
= +2V, COMP = open, C
= 1FF, C
= 1FF, T = T = -40NC to +125NC, unless otherwise noted.)
EN/UVLO
VIN
VCC A J
OVI FALLING THRESHOLD
vs. TEMPERATURE
V
IN
SUPPLY CURRENT UNDER UVLO
SWITCHING CURRENT
vs. TEMPERATURE
MAX17595/6/7 toc10
vs. TEMPERATURE
MAX17595/6/7 toc08
MAX17595/6/7 toc09
1.1505
1.1500
1.1495
1.1490
1.1485
1.1480
25.5
24.5
23.5
22.5
21.5
20.5
19.5
2.5
2.4
2.3
2.2
2.1
2.0
1.9
1.8
1.7
1.6
1.5
1.1475
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
NDRV SWITCHING FREQUENCY
vs. RESISTOR
NDRV SWITCHING FREQUENCY
vs. TEMPERATURE
MAX17595/6/7 toc11
MAX17595/6/7 toc12
1000
900
800
700
600
500
400
300
200
100
0
950
850
750
650
550
450
350
250
150
50
R
= 10kI
RT
R
= 100kI
RT
5
15 25 35 45 55 65 75 85 95
-40 -20
0
20 40 60 80 100 120
TEMPERATURE (°C)
FREQUENCY SELECTION RESISTOR (kI)
SWITCHING WAVEFORMS (MAX17595)
FREQUENCY DITHERING vs. R
DITHER
MAX17595/6/7 toc13
MAX17595/6/7 toc14
14
12
10
8
V
DRAIN
100V/div
6
I
PRI
4
1A/div
2
200 300 400 500 600 700 800 900 1000
(kI)
4µs/div
R
DITHER
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7
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Typical Operating Characteristics (continued)
(V = 15V, V
= +2V, COMP = open, C
= 1FF, C
= 1FF, T = T = -40NC to +125NC, unless otherwise noted.)
IN
EN/UVLO
VIN
VCC A J
ENABLE STARTUP
ENABLE SHUTDOWN
HICCUP OPERATION
MAX17595/6/7 toc17
MAX17595/6/7 toc15
MAX17595/6/7 toc16
EN/UVLO
5V/div
EN/UVLO
5V/div
V
OUT
10V/div
V
OUT
10V/div
V
OUT
V
DRAIN
10V/div
100V/div
COMP
1V/div
COMP
1V/div
I
PRI
2A/div
2ms/div
400µs/div
1ms/div
LOAD TRANSIENT RESPONSE
(15V OUTPUT)
SWITCHING CURRENT
vs. SWITCHING FREQUENCY
MAX17595/6/7 toc18
MAX17595/6/7 toc19
2.5
2.3
2.1
1.9
1.7
1.5
V
(AC)
OUT
0.5V/div
I
LOAD
0.5A/div
100 200 300 400 500 600 700 800 900 1000
SWITCHING FREQUENCY (Hz)
20ms/div
BODE PLOT (15V OUTPUT)
EFFICIENCY GRAPH (15V OUTPUT)
MAX17595/6/7 toc20
MAX17595/6/7 toc21
100
90
80
70
60
50
40
30
20
10
0
V
= 120V
DC
PHASE
36°/div
BANDWIDTH = 8.8kHz
PHASE MARGIN = 64°
GAIN
10dB/div
6 8 1
2
4
6 8 1
2
4 6 8
0
0.2 0.4 0.6 0.8 1.0 1.2 1.4
LOAD CURRENT (A)
����������������������������������������������������������������� Maxim Integrated Products
8
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Pin Configuration
TOP VIEW
12
11
10
9
SGND
SS
13
14
V
8
7
6
5
DRV
V
IN
MAX17595
MAX17596
MAX17597
FB
EN/UVLO 15
16
EP
4
COMP
OVI
+
1
2
3
TQFN
Pin Description
PIN
NAME
FUNCTION
1, 12
N.C.
No Connection
Slope Compensation Input. A resistor, R
, connected from SLOPE to SGND programs the
SLOPE
amount of slope compensation with reference-voltage soft-start mode. Connecting this pin to
SGND enables duty-cycle soft-start with minimum slope compensation of 50mV/Fs. Setting V
> 4V enables reference voltage soft-start with minimum slope compensation of 50mV/Fs.
2
3
4
SLOPE
RT
SLOPE
Switching Frequency Programming Resistor Connection. Connect resistor R from RT to SGND to
RT
set the PWM switching frequency.
Frequency Dithering Programming or Synchronization Connection. For spread-spectrum frequency
operation, connect a capacitor from DITHER to SGND, and a resistor from DITHER to RT. To
synchronize the internal oscillator to the externally applied frequency, connect DITHER/SYNC to
the synchronization pulse.
DITHER/SYNC
Transconductance Amplifier Output. Connect the frequency compensation network between
COMP and SGND.
5
6
7
COMP
FB
Transconductance Amplifier Inverting Input
Soft-Start Capacitor Pin for Flyback Regulator. Connect a capacitor C from SS to SGND to set
SS
the soft-start time interval.
SS
8
9
SGND
CS
Signal Ground. Connect SGND to the signal ground plane.
Current-Sense Input. Peak-current-limit trip voltage is 300mV.
Power Ground. Connect PGND to the power ground plane.
External Switching nMOS Gate-Driver Output
10
11
PGND
NDRV
����������������������������������������������������������������� Maxim Integrated Products
9
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Pin Description (continued)
PIN
NAME
FUNCTION
Linear Regulator Output and Driver Input. Connect input bypass capacitor from V
close as possible to the IC.
to SGND as
DRV
13
V
DRV
Internal V
a 1FF minimum ceramic capacitor.
Regulator Input. Connect V to the input voltage source. Bypass V to PGND with
IN IN
DRV
14
15
V
IN
Enable/Undervoltage Lockout. To externally program the UVLO threshold of the input supply,
connect a resistive divider between input supply, EN, and SGND.
EN/UVLO
Overvoltage Comparator Input. Connect a resistive divider between the input supply, OVI, and
SGND to set the input overvoltage threshold.
16
—
OVI
EP
Exposed Pad
Input Voltage Range (V
)
IN
Detailed Description
The MAX17595 has different rising and falling under-
The MAX17595 offers a bootstrap UVLO wakeup level
of 20V with a wide hysteresis of 15V minimum, and is
optimized for implementing isolated and non-isolated
universal (85V to 265V AC) offline single-switch flyback
converter or telecom (36V to 72V) power supplies. The
MAX17596/MAX17597 offer a UVLO wakeup level of
4.4V and are well-suited for low-voltage DC-DC flyback/
boost power supplies. An internal 1% reference (1.21V)
can be used to regulate the output down to 1.21V in
nonisolated flyback and boost applications. Additional
semi-regulated outputs, if needed, can be generated
by using additional secondary windings on the flyback
converter transformer.
voltage lockout (UVLO) thresholds on the V pin than
IN
the thresholds of the MAX17596/MAX17597. The thresh-
olds for the MAX17595 are optimized for implementing
power supply startup schemes, typically used for offline
AC-DC power supplies. The MAX17595 is well-suited for
operation from rectified DC bus in AC-DC power-supply
applications, which are typical of front-end industrial
power-supply applications. As such, the MAX17595 has no
limitationonmaximuminputvoltage,aslongastheexternal
components are rated suitably and the maximum
operating voltages of the MAX17595 are respected.
The MAX17595 can be successfully used in universal
input (85V to 265V AC) rectified bus applications, in recti-
fied 3-phase DC bus applications, and in telecom (36V to
72V DC) applications.
The MAX17595/MAX17596/MAX17597 family utilizes
peak-current-mode control and external compensation
for optimizing closed-loop performance. The devices
include cycle-by-cycle peak current limit, and eight
consecutive occurrences of current-limit-event trig-
ger hiccup mode, which protects external com-
ponents by halting switching for a period of 32,768
cycles. The devices also include voltage soft-start
for nonisolated designs, and current soft-start for
isolated designs to allow monotonic and smooth rise of the
outpu voltage during startup. The voltage and current
soft-start modes can be selected using the SLOPE pin.
See Figure 1 for more information.
The MAX17596/MAX17597 are intended to implement
flyback (isolated and nonisolated) and boost convert-
ers. The V pin of the MAX17596/MAX17597 has a
IN
maximum operating voltage of 36V. The MAX17596/
MAX17597 implement rising and falling thresholds on
the V pin that assume power-supply startup schemes
IN
typical of low-voltage DC-DC applications, down to
an input voltage of 4.5V DC. Therefore, flyback/boost
converters with a 4.5V to 36V supply voltage range can
be implemented with the MAX17596/MAX17597.
���������������������������������������������������������������� Maxim Integrated Products 10
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
THERMAL SENSOR
50µA
7.5V (MAX17595)
OR
5V (MAX17596/
MAX17597)
MAX17595
MAX17596
MAX17597
DITHER
V
DRV
5V
LDO
LDO
CONTROL
AND
2V/0.4V
AV
DRIVER LOGIC
POK
HICCUP
V
DRV
V
IN
NDRV
DRIVER
PGND
8 PEAK EVENTS
OR 1 RUNAWAY
SSDONE
UVLO
1.21V
CHIPEN
OSC
OSC
PEAKLIM
COMP
EN/
UVLO
PGND
305mV
360mV
DITHER
(SYNC)
RUNAWAY
COMP
OVI
1.21V
BLANKING
70ns
CS
PWM COMP
RT
FIXED
OR VAR
10µA
10µA
CHIPPEN
SLOPE
SLOPE
DECODE
SS
SS
SSDONE
SS MODE OSC
1.23V
R
5µA
COMP
1X
CHIPPEN/
HICCUP
(FACTORY OPTION)
VOLTAGE
SOFT-START
1.21V
SS MODE
SS
R
SGND
FB
SS
CURRENT
SOFT-START
SS MODE
Figure 1. MAX17595/MAX17596/MAX17597 Block Diagram
���������������������������������������������������������������� Maxim Integrated Products 11
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
duration is twice that of the programmed soft-start period.
This is particularly useful in implementing controlled
shutdown of output voltage in isolated power converters.
Internal Linear Regulator (V
)
DRV
The internal functions and driver circuits are designed
to operate from 7.5V (MAX17595) or 5V (MAX17596/
MAX17597) power supply voltages. The MAX17595/
MAX17596/MAX17597 family has an internal linear regu-
Switching Frequency Selection (RT)
The ICs’ switching frequency is programmable between
lator that is powered from the V pin. The output of the
IN
100kHz and 1MHz with a resistor R connected between
RT
linear regulator is connected to the V
pin, and should
DRV
RT and SGND. Use the following formula to determine the
be decoupled with a 1FF capacitor to ground for stable
operation. The V regulator output supplies all the oper-
appropriate value of R needed to generate the desired
RT
DRV
output-switching frequency (f ):
SW
ating current of the MAX17595/MAX17596/MAX17597.
10
The maximum operating voltage on the V pin is 29V for
IN
10
R
=
the MAX17595, and 36V for the MAX17596/MAX17597.
RT
f
SW
n-Channel MOSFET Gate Driver (NDRV)
The MAX17595/MAX17596/MAX17597 family offers a
built-in gate driver for driving an external n-channel
MOSFET. The NDRV pin can source/sink currents in
excess of 650mA/1000mA.
where f
is the desired switching frequency.
SW
Frequency Dithering for
Spread-Spectrum Applications (Low EMI)
The switching frequency of the converter can be
dithered in a range of Q10% by connecting a capaci-
tor from DITHER/SYNC to SGND, and a resistor from
DITHER to RT, as shown in the Typical Operating Circuits.
Spread-spectrum modulation technique spreads the
energy of switching frequency and its harmonics over a
wider band while reducing their peaks, helping to meet
stringent EMI goals.
Maximum Duty Cycle
The MAX17595/MAX17596 operate at a maximum duty
cycle of 49%. The MAX17597 offers a maximum duty
cycle of 94% to implement flyback and boost converters
involving large input-to-output voltage ratios in DC-DC
applications. Slope compensation is necessary for stable
operation of peak-current-mode controlled converters
such as the MAX17595/MAX17596/MAX17597 at duty
cycles greater than 50%, in addition to the loop compen-
sation required for small signal stability. The MAX17595/
MAX17596/MAX17597 implement a SLOPE pin for this
purpose. See the Slope Compensation section for more
details.
Applications Information
Startup Voltage and Input Overvoltage
Protection Setting (EN/UVLO, OVI)
The devices’ EN/UVLO pin serves as an enable/disable
input, as well as an accurate programmable input UVLO
pin. The devices do not commence startup operation
unless the EN/UVLO pin voltage exceeds 1.21V (typ).
The devices turn off if the EN/UVLO pin voltage falls
below 1.15V (typ). A resistor-divider from the input DC
bus to ground can be used to divide down and apply a
Soft-Start (SS)
The MAX17595/MAX17596/MAX17597 devices imple-
ment soft-start operation for the flyback/boost regulator.
A capacitor connected to the SS pin programs the soft-
start period. The soft-start feature reduces input inrush
current during startup. The devices allow the end user
to select between voltage soft-start, usually preferred in
nonisolated applications, and current soft-start, which is
useful in isolated applications to get a monotonic and
smooth rise in output voltage. See the Input Voltage
fraction of the input DC voltage (V ) to the EN/UVLO
DC
pin. The values of the resistor-divider can be selected
so that the EN/UVLO pin voltage exceeds the 1.23V (typ)
turn-on threshold at the desired input DC bus voltage. The
same resistor-divider can be modified with an additional
Range (V ) section.
IN
resistor (R ) to implement input overvoltage protec-
OVI
tion in addition to the EN/UVLO functionality as shown
in Figure 2. When voltage at the OVI pin exceeds
1.21V (typ), the devices stop switching and resume
switching operations only if voltage at the OVI pin falls
below 1.15V (typ). For given values of startup DC input
Soft-Stop
A soft-stop feature can be requested from the factory.
This feature ramps down the duty cycle of operation of
the converter to zero in a controlled fashion, and enables
controlled ramp down of output voltage. The soft-stop
���������������������������������������������������������������� Maxim Integrated Products 12
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
voltage (V
) and input overvoltage-protection
START
voltage (V ), the resistor values for the divider can
OVI
be calculated as follows, assuming a 24.9kI resistor
R
R
R
DC1
DC2
DC3
for R
:
OVI
V
OVI
R
SUM
R
= R
×
OVI
−1 kI
EN
V
START
where R
is in kI, while V
and V
are in volts.
OVI
R
START
OVI
EN/UVLO
OVI
V
START
1.21
= R
+ R
EN
×
−1 kI
SUM
OVI
MAX17595
MAX17596
MAX17597
R
R
EN
where R
R
is in kI, while V is in volts.
START
,
EN
OVI
In universal AC input applications, R
to be implemented as equal resistors in series (R
might need
OVI
SUM
,
DC1
R
DC2
, and R ) so that voltage across each resistor is
DC
limited to its maximum operation voltage.
Figure 2. Programming EN/UVLO and OVI
R
SUM
3
R
= R
= R =
DC3
kI
DC1
DC2
(V
), and a bias voltage (V
IN
) that is bootstrapped
BIAS
OUT
to the V pin through the diode (D2). If V
exceeds
BIAS
For low-voltage DC-DC applications based on the
MAX17596/MAX17597, a single resistor can be used in
the sum of 7V, and the drop across D2 before the volt-
age on C falls below 7V, then the V voltage is
START
IN
the place of R
mately 40V.
, as the voltage across it is approxi-
SUM
sustained by V
operating with energy from V
, allowing the MAX17595 to continue
BIAS
. The large hysteresis
BIAS
(13V typ) of the MAX17595 allows for a small startup
capacitor (C ). The low startup current (20FA typ)
Startup Operation
START
The MAX17595 is optimized for implementing an offline
allows the use of a large startup resistor (R
),
START
single-switch flyback converter and has a 20V V UVLO
IN
thus reducing power dissipation at higher DC bus volt-
ages. Figure 3 shows the typical RC startup scheme
wake-up level with hysteresis of 15V (min). In offline
applications, a simple cost-effective RC startup circuit is
used. When the input DC voltage is applied, the startup
for the MAX17595, when the output voltage V
is
OUT
used as the bias voltage to sustain switching operation.
might need to be implemented as equal, multiple
resistor (R
) charges the startup capacitor (C
),
START
START
R
START
causing the voltage at the V pin to increase towards
IN
resistors in series (R , R , and R ) to share the
IN1
IN2
IN3
the wake-up V UVLO threshold (20V typ). During this
time, the MAX17595 draws a low startup current of 20FA
IN
applied high DC voltage in offline applications so
that the voltage across each resistor is limited to its
(typ) through R
. When the voltage at V reaches
START
IN
maximum continuous operating voltage rating. R
START
the wake-up V UVLO threshold, the MAX17595 com-
IN
and C
can be calculated as:
START
mences switching and control operations. In this con-
dition, the MAX17595 draws 2mA (typ) current from
Q
× f
6
t
SS
10
GATE SW
C
START
, when operated at 1MHz switching frequency,
C
= I
+
×
FF
START
IN
for its internal operation. In addition, the average value
of gate drive current is also drawn from C , which
10
START
where I is the supply current drawn at the V pin in
IN
GATE
used in nC, f
IN
is a function of the gate charge of the external MOSFET
used. Since this total current cannot be supported by
mA, Q
is the gate charge of the external MOSFET
SW
SS
is the switching frequency of the convert-
the current through R
, the voltage on C
starts
START
START
er in Hz, and t is the soft-start time programmed for the
to drop. When suitably configured, as shown in Figure
9, the external MOSFET is switched by the NDRV pin
and the flyback converter generates an output voltage
flyback converter in ms. See the Programming Soft-Start
of Flyback/Boost Converter (SS) section.
���������������������������������������������������������������� Maxim Integrated Products 13
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
The startup capacitor (C
) can be calculated as:
START
V
−10 × 50
(
)
START
R
=
kI
START
1+ C
Q
× f
t
SS
START
GATE SW
C
= I
+
×
FF
START
IN
6
10
10
where C
is the startup capacitor in FF.
START
where I is the supply current drawn at the V pin in
For designs that cannot accept power dissipation in the
startup resistors at high DC input voltages in offline appli-
cations, the startup circuit can be set up with a current
source instead of a startup resistor as shown in Figure 4.
IN
IN
mA, Q
used in nC, f
is the gate charge of the external MOSFET
GATE
is the switching frequency of the con-
SW
verter in kHz, and t is the soft-start time programmed
SS
for the flyback converter in ms.
V
DC
V
OUT
R
R
R
IN1
IN2
IN3
R
START
V
C
DC
F
V
OUT
MAX17595
NDRV
CS
V
IN
LDO
DRV
C
START
V
DRV
C
VDRV
Figure 3. MAX17595 RC-Based Startup Circuit
V
DC
R
R
R
IN1
IN2
IN3
R
SUM
V
V
OUT
DC
D1
C
OUT
V
OUT
MAX17595
R
ISRC
NDRV
CS
V
IN
LDO
DRV
C
START
V
DRV
R
S
C
VDRV
Figure 4. MAX17595 Current-Source-Based Startup Circuit
���������������������������������������������������������������� Maxim Integrated Products 14
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Resistors R
and R
can be calculated as:
incurred to supply the operating current of the MAX17596/
MAX17597 can be tolerated, the V pin is directly
SUM
ISRC
IN
V
START
10
connected to the DC input, as shown in Figure 5. In the
case of higher DC input voltages (e.g., 16V to 32V DC),
a startup circuit, such as that shown in Figure 6, can be
used to minimize power dissipation in the startup circuit.
In this startup scheme, the transistor (Q1) supplies the
switching current until a bias winding NB comes up. The
R
=
MW
MW
SUM
V
BEQ1
70
R
=
ISRC
The V
UVLO wakeup threshold of the MAX17596/
IN
resistor (R ) can be calculated as:
MAX17597 is set to 4.1V (typ) with a 200mV hysteresis,
optimized for low-voltage DC-DC applications down to
4.5V. For applications where the input DC voltage is low
enough (e.g., 4.5V to 5.5V DC) that the power loss
Z
R
= 9 ×(V
− 6.3) kW
Z
INMIN
V
DC
V
OUT
D1
V
IN
V
DRV
V
LDO
IN
C
OUT
C
DRV
Np
Ns
NDRV
CS
R
S
MAX17596
MAX17597
Figure 5. MAX17596/MAX17597 Typical Startup Circuit with V Connected Directly to DC Input
IN
V
DC
D1
R
Z
V
IN
V
DRV
NB
Q
LDO
1
C
Z
D1
6.3V
C
DRV
OUT
Np
Ns
V
IN
C
IN
NDRV
CS
R
S
MAX17596
MAX17597
Figure 6. MAX17596/MAX17597 Typical Startup Circuit with Bias Winding to Turn Off Q1 and Reduce Power Dissipation
���������������������������������������������������������������� Maxim Integrated Products 15
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Programming Soft-Start of
V
OUT
Flyback/Boost Converter (SS)
The soft-start period in the voltage soft-start scheme of
the devices can be programmed by selecting the value
of the capacitor connected from the SS pin to SGND. The
R
U
MAX17595
MAX17596
MAX17597
FB
capacitor C can be calculated as:
SS
C
= 8.2645× t nF
SS
R
B
SS
where t
is expressed in ms. The soft-start period in
SS
the current soft-start scheme depends on the load at the
output and the soft-start capacitor.
Figure 7. Programming Output Voltage
Programming Output Voltage
The devices incorporate an error amplifier with a 1% pre-
cision voltage reference that enables negative feedback
control of the output voltage. The output voltage of the
switching converter can be programmed by selecting the
when needed. Set the corner frequency between 10MHz
and 20MHz. After the leading-edge blanking time,
the device monitors V . The duty cycle is terminated
CS
immediately when V exceeds 300mV.
CS
values for the resistor-divider connected from V
, and
OUT
The devices offer a runaway current limit scheme that
protects the devices under high-input-voltage short-
circuit conditions when there is insufficient output volt-
age available to restore inductor current built up during
the on period of the flyback/boost converter. Either eight
consecutive occurrences of the peak-current-limit event
or one occurrence of the runaway current limit trigger a
hiccup mode that protects the converter by immediately
the flyback/boost output to ground, with the midpoint of
the divider connected to the FB pin (Figure 7). With R
B
selected in the 20kI to 50kI range, R can be calcu-
lated as:
U
V
OUT
R
= R
×
−1 kI, whereR is in kI.
U
B
B
1.21
suspending switching for a period of time (t
).
RSTART
Peak-Current-Limit Setting (CS)
The devices include a robust overcurrent protection
scheme that protects the device under overload and
This allows the overload current to decay due to power
loss in the converter resistances, load, and the output
diode of the flyback/boost converter before soft-start
is attempted again. The runaway current limit is set
short-circuit conditions. A current-sense resistor (R
CS
in the Typical Operating Circuits), connected between
the source of the MOSFET and PGND, sets the peak
current limit. The current-limit comparator has a voltage
at a V
of 360mV (typ). The peak-current-limit-
CS-PEAK
triggered hiccup operation is disabled until the end of
the soft-start period, while the runaway current-limit-
triggered hiccup operation is always enabled.
trip level (V
) of 300mV. Use the following equa-
CS-PEAK
tion to calculate the value of R
:
CS
Programming Slope
Compensation (SLOPE)
300mV
R
=
I
CS
I
MOSFET
The MAX17595/MAX17596 operate at a maximum duty
cycle of 49%. In theory, they do not require slope
compensation to prevent subharmonic instability that
occurs naturally in continuous-conduction mode (CCM)
peak-current-mode-controlled converters operating at
duty cycles greater than 50%. In practice, the MAX17595/
MAX17596 require a minimum amount of slope compen-
sation to provide stable operation. The devices allow the
user to program this default value of slope compensation
simply by leaving the SLOPE pin unconnected. It is rec-
ommended that discontinuous-mode designs also use
this minimum amount of slope compensation to provide
better noise immunity and jitter-free operation.
where I
is the peak current flowing through the
MOSFET
MOSFET. When the voltage produced by this current
(through the current-sense resistor) exceeds the current-
limit comparator threshold, the MOSFET driver (NDRV)
terminates the current on-cycle within 30ns (typ).
The devices implement 65ns of leading-edge blanking
to ignore leading-edge current spikes. These spikes are
caused by reflected secondary currents, capacitance
discharging current at the MOSFET’s drain, and gate
charging current. Use a small RC network for additional
filtering of the leading edge spike on the sense waveform
���������������������������������������������������������������� Maxim Integrated Products 16
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
The MAX17597 flyback/boost converter can be designed
Error Amplifier, Loop Compensation,
and Power Stage Design of
to operate in either discontinuous-conduction mode
(DCM) or to enter into the continuous-conduction mode
at a specific load condition for a given DC input
voltage. In continuous-conduction mode, the flyback/
boost converter needs slope compensation to avoid
subharmonic instability that occurs naturally over all
specified load and line conditions in peak-current-mode
controlled converters operating at duty cycles greater
than 50%. A minimum amount of slope signal is added to
the sensed current signal even for converters operating
below 50% duty to provide stable, jitter-free operation.
The SLOPE pin allows the user to program the necessary
slope compensation by setting the value of the resistor
Flyback/Boost Converter
The flyback/boost converter requires proper loop
compensation to be applied to the error-amplifier output
to achieve stable operation. The goal of the compensator
design is to achieve desired closed-loop bandwidth,
and sufficient phase margin at the crossover frequency
of the open-loop gain-transfer function of the converter.
The error amplifier provided in the devices is a transcon-
ductance amplifier. The compensation network used
to apply the necessary loop compensation is shown in
Figure 8.
The flyback/boost converter can be used to implement
the following converters and operating modes:
(R ) connected from the SLOPE pin to ground.
SLOPE
S
− 8
•ꢀ Nonisolated flyback converter in discontinuous-con-
E
R
=
kI
SLOPE
duction mode (DCM flyback)
1.55
where the slope (S ) is expressed in mV/Fs.
•ꢀ Nonisolated flyback converter in continuous-conduc-
E
tion mode (CCM flyback)
Frequency Dithering for
•ꢀ Boost converter in discontinuous-conduction mode
Spread-Spectrum Applications (Low EMI)
The switching frequency of the converter can be dithered
in a range of Q10% by connecting a capacitor from
DITHER/SYNC to SGND, and a resistor from DITHER
to RT as shown in the Typical Operating Circuits. This
results in lower EMI.
(DCM boost)
•ꢀ Boost converter in continuous-conduction mode
(CCM boost)
Calculations for loop-compensation values (R , C , and
C ) for these converter types and design procedures for
power-stage components are detailed in the following
sections.
Z
Z
P
A current source at DITHER/SYNC charges the capacitor
C
to 2V at 50FA. Upon reaching this trip point, it
DITHER
discharges C
to 0.4V at 50FA. The charging and
DITHER
discharging of the capacitor generates a triangular wave-
form on DITHER/SYNC with peak levels at 0.4V and 2V
and a frequency that is equal to:
COMP
50FA
f
=
TRI
MAX17595
R
Z
C
× 3.2V
DITHER
C
P
MAX17596
MAX17597
C
Z
typically, f
should be set close to 1kHz. The resistor
TRI
R
connected from DITHER/SYNC to RT deter-
DITHER
mines the amount of dither as follows:
R
RT
DITHER
%DITHER =
R
Figure 8. Error-Amplifier Compensation Network
where %DITHER is the amount of dither expressed as a
percentage of the switching frequency. Setting R
DITHER
to 10 x R generates Q10% dither.
RT
���������������������������������������������������������������� Maxim Integrated Products 17
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Maximum secondary peak current:
DCM Flyback
I
PRIPEAK
K
I
=
SECPEAK
Primary Inductance Selection
In a DCM flyback converter, the energy stored in the
primary inductance of the flyback transformer is
delivered entirely to the output. The maximum primary
inductance value for which the converter remains in DCM
at all operating conditions can be calculated as:
Maximum primary peak current:
I
xL
x f
PRI SW
+ V )
D
SECPEAK
I
= I
SECRMS PRIPEAK
3 x(V
OUT
2
V
×D
× 0.4
× f
(
≤
)
For the purpose of current-limit setting, I
lated as follows:
can be calcu-
INMIN
MAX
LIM
L
PRIMAX
V
+ V ×I
OUT
D
OUT SW
I
= I
×1.2
LIM PRIPEAK
where D
is chosen as 0.35 for the MAX17595/
MAX
MAX17596 and 0.7 for the MAX17597; V is the
voltage drop of the output rectifier diode on the secondary
D
Primary Snubber Selection
Ideally, the external MOSFET experiences a drain-source
voltage stress equal to the sum of the input voltage and
reflected voltage across the primary winding during the
off period of the MOSFET. In practice, parasitic inductors
and capacitors in the circuit, such as leakage inductance
of the flyback transformer, cause voltage overshoot and
ringing, in addition to the ideally expected voltage stress.
Snubber circuits are used to limit the voltage overshoots
to safe levels within the voltage rating of the external
MOSFET. The snubber capacitor can be calculated
using the following equation:
winding, and f
is the switching frequency of the power
SW
converter. Choose the primary inductance value to be
less than L
.
PRIMAX
Duty Cycle Calculation
The accurate value of the duty cycle (D ) for the
selected primary inductance (L ) can be calculated
NEW
PRI
using the following equation:
2.5×L
× V
(
+ V ×I × f
OUT SW
)
PRI
OUT
D
D
=
NEW
V
INMIN
2
2
2 ×L ×I
×K
Turns Ratio Calculation (Ns/Np)
Transformer turns ratio (K = Ns/Np) can be calculated as:
LK PRIPEAK
C
=
SNUB
2
V
OUT
V
+ V ×(1− D
)
(
)
OUT
V
D
MAX
MAX
where L
is the leakage inductance that can be
LK
K =
obtained from the transformer specifications (usually
1.5%–2% of the primary inductance).
×D
INMIN
The power to be dissipated in the snubber resistor is
calculated using the following formula:
Peak/RMS Current Calculation
The transformer manufacturer needs RMS current
values in the primary and secondary to design the wire
diameter for the different windings. Peak current calcula-
tions are useful in setting the current limit. Use the fol-
lowing equations to calculate the primary and secondary
peak and RMS currents.
2
P
= 0.833×L ×I
× f
SW
SNUB
LK PRIPEAK
The snubber resistor is calculated based on the equation
below:
2
6.25× V
Maximum primary peak current:
OUT
R
=
SNUB
2
P
×K
V
×D
× f
SNUB
INMIN
NEW
I
=
PRIPEAK
L
The voltage rating of the snubber diode is:
PRI SW
V
OUT
K
Maximum primary RMS current:
= I
V
= V
+ 2.5×
DSNUB
INMAX
D
NEW
3
I
×
PRIRMS PRIPEAK
���������������������������������������������������������������� Maxim Integrated Products 18
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
A) Capacitor selection based on switching ripple
Output Capacitor Selection
X7R ceramic output capacitors are preferred in industrial
applications due to their stability over temperature. The
output capacitor is usually sized to support a step load
of 50% of the maximum output current in the application,
so that the output-voltage deviation is contained to 3% of
the output-voltage change. The output capacitance can
be calculated as follows:
(MAX17596/MAX17597)
For DC-DC applications, X7R ceramic capacitors
are recommended due to their stability over the
operating temperature range. The ESR and ESL of a
ceramic capacitor are relatively low, so the ripple
voltage is dominated by the capacitive component.
For the flyback converter, the input capacitor sup-
plies the current when the main switch is on. Use the
following equation to calculate the input capacitor
for a specified peak-to-peak input switching ripple
I
× t
RESPONSE
STEP
C
t
=
OUT
∆V
OUT
(V
IN_RIP
):
0.33
1
≅
+
2
RESPONSE
f
f
SW
NEW
D
×I
1− 0.5×D
(
)
C
NEW PRIPEAK
C
=
IN
2× f
× V
SW
IN_RIP
where I
time of the controller, DV
age deviation, and f is the target closed-loop crossover
frequency. f is chosen to be one-tenth of the switching
is the load step, t
is the response
RESPONSE
STEP
is the allowable output volt-
OUT
B) Capacitor selection based on rectified line voltage
C
ripple (MAX17595)
C
SW
For the flyback converter, the input capacitor
supplies the input current when the diode rectifier is
frequency, f . For the flyback converter, the output
capacitor supplies the load current when the main
switch is on; therefore, the output voltage ripple is a
function of load current and duty cycle. Use the following
equation to calculate the output capacitor ripple:
off. The voltage discharge (V
average current, should be within the limits specified:
), due to the input
IN_RIP
0.5×I
×D
PRIPEAK
NEW
× V
IN_RIP
C
=
IN
2
f
RIPPLE
D
× I
− K ×I
NEW
PRIPEAK
OUT
∆V
=
COUT
2×I
× f
× C
where f
, the input AC ripple frequency equal
RIPPLE
PRIPEAK SW
OUT
to the supply frequency for half-wave rectification,
is two times the AC supply frequency for full-wave
rectification.
where I
minimum input voltage.
is load current and D
is the duty cycle at
OUT
NEW
Input Capacitor Selection
C) Capacitor selection based on holdup time require-
The MAX17595 is optimized to implement offline AC-DC
converters. In such applications, the input capacitor
must be selected based on either the ripple due to
the rectified line voltage, or based on holdup-time
requirements. Holdup time can be defined as the time
period over which the power supply should regulate
its output voltage from the instant the AC power fails.
The MAX17596/MAX17597 are useful in implementing
low-voltage DC-DC applications where the switching-
frequency ripple must be used to calculate the input
capacitor. In both cases, the capacitor must be sized to
meet RMS current requirements for reliable operation.
ments (MAX17595)
For a given output power (P
be delivered during holdup time (t
voltage at which the AC supply fails (V
the minimum DC bus voltage at which the converter
) that needs to
HOLDUP
), DC bus
HOLDUP
), and
INFAIL
can regulate the output voltages (V
), the input
INMIN
capacitor (C ) is estimated as:
IN
3×P
× t
HOLDUP
2
HOLDUP
2
C
=
IN
(V
− V
)
INFAIL
INMIN
the input capacitor RMS current can be calculated as:
2
0.6× V
× D
(
)
INMIN
MAX
×L
PRI
I
=
INCRMS
f
SW
���������������������������������������������������������������� Maxim Integrated Products 19
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
where:
External MOSFET Selection
A MOSFET selection criterion includes maximum drain
voltage, peak/RMS current in the primary, and the
maximum allowable power dissipation of the package
without exceeding the junction temperature limits. The
voltage seen by the MOSFET drain is the sum of the input
voltage, the reflected secondary voltage on the trans-
former primary, and the leakage inductance spike. The
I
OUT
f
=
P
π × V
× C
OUT
OUT
1
C
=
Z
π ×R × f
Z
1
P
C
=
P
π ×R × f
SW
MOSFET’s absolute maximum V rating must be higher
Z
DS
than the worst-case drain voltage:
f
is the switching frequency of the devices.
SW
V
+ V
OUT
DIODE
V
= V
+
× 2.5
CCM Flyback
DSMAX
INMAX
K
Transformer Turns Ratio Calculation
(K = Ns/Np)
The transformer turns ratio can be calculated using the
following formula:
The drain current rating of the external MOSFET is
selected to be greater than the worst-case peak-current-
limit setting.
Secondary Diode Selection
Secondary-diode selection criteria includes the maxi-
mum reverse voltage, average current in the secondary-
reverse recovery time, junction capacitance, and the
maximum allowable power dissipation of the package.
The voltage stress on the diode is the sum of the output
voltage and the reflected primary voltage. The maximum
operating reverse-voltage rating must be higher than the
worst-case reverse voltage:
V
+ V ×(1− D
)
MAX
(
)
OUT
V
D
K =
×D
INMIN
MAX
where D
is the duty cycle assumed at minimum
input (0.35 for the MAX17595/MAX17596 and 0.7 for the
MAX17597).
MAX
Primary Inductance Calculation
Calculate the primary inductance based on the ripple:
V
+ V ×(1− D
) ×K
(
=
)
V
= 1.25×(K × V
+ V
)
OUT
D
NOM
SW
SECDIODE
INMAX
OUT
L
PRI
2×I
×β × f
OUT
The current rating of the secondary diode should be
selected so that the power loss in the diode (given as
the product of forward-voltage drop and the average
diode current) should be low enough to ensure that the
junction temperature is within limits. This necessitates
where D
, the nominal duty cycle at nominal operating
NOM
DC input voltage V
, is given as:
INNOM
V
+ V ×K
D
(
)
OUT
D
=
NOM
+ V ×K
OUT D
V
+ V
that the diode current rating be in the order of 2 x I
(
)
INNOM
OUT
to 3 x I
. Select fast-recovery diodes with a recovery
OUT
The output current, down to which the flyback converter
should operate in CCM, is determined by selection of
the fraction A in the above primary inductance formula.
For example, A should be selected as 0.15 so that the
converter operates in CCM down to 15% of the maximum
output load current. Since the ripple in the primary current
waveform is a function of duty cycle and is maximum at
maximum DC input voltage, the maximum (worst-case)
load current down to which the converter operates in
CCM occurs at maximum operating DC input voltage.
time less than 50ns, or Schottky diodes with low junction
capacitance.
Error Amplifier Compensation Design
The loop compensation values are calculated as:
2
0.1× f
SW
1+
× V
× f
×I
OUT OUT
f
P
R
= 450×
Z
2×L
PRI SW
V
is the forward drop of the selected output diode at
D
maximum output current.
���������������������������������������������������������������� Maxim Integrated Products 20
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
of 50% of the maximum output current in the application
so that the output-voltage deviation is contained to 3% of
the output-voltage change. The output capacitance can
be calculated as:
Peak and RMS Current Calculation
RMS current values in the primary and secondary are
needed by the transformer manufacturer to design the
wire diameter for the different windings. Peak current
calculations are useful in setting the current limit. Use the
following equations to calculate the primary and secondary
peak and RMS currents.
I
× t
RESPONSE
STEP
C
=
OUT
∆V
OUT
0.33
1
Maximum primary peak current:
t
≅ (
+
)
RESPONSE
f
f
SW
I
×K
V
×D
C
OUT
INMIN MAX
I
=
+
PRIPEAK
1− D
2×L
× f
where I
is the load step, t
is the response
RESPONSE
MAX
PRI SW
STEP
time of the controller, DV
is the allowable output
OUT
Maximum primary RMS current:
2
voltage deviation, and f is the target closed-loop cross-
over frequency. f is chosen to be less than one-fifth of
the worst-case (lowest) RHP zero frequency f
right half-plane zero frequency is calculated as follows:
C
2
C
I
+ ∆I
− I
(
× ∆I
)
PRIPEAK
PRI
PRIPEAK
PRI
I
=
. The
PRIRMS
RHP
3
×
D
MAX
2
(1− D
) × V
OUT
MAX
f
=
where DI
waveform and is given by:
is the ripple current in the primary current
PRI
ZRHP
2
2× π ×D
×L
×I
×K
MAX
PRI OUT
V
×D
MAX
For the CCM flyback converter, the output capacitor
supplies the load current when the main switch is on;
therefore, the output voltage ripple is a function of load
current and duty cycle. Use the following equation to
estimate the output voltage ripple:
INMIN
L
∆I
=
PRI
× f
PRI SW
Maximum secondary peak current:
I
PRIPEAK
K
I
=
SECPEAK
I
×D
OUT
MAX
OUT
∆V
=
COUT
Maximum secondary RMS current:
f
× C
SW
2
2
I
+ ∆I
+ I
× ∆I
SECPEAK SEC
(
)
SECPEAK
SEC
I
=
Input Capacitor Selection
The design procedure for input capacitor selection is
SECRMS
3
× 1− D
identical to that outlined in the DCM Flyback section.
MAX
where DI
waveform and is given by:
is the ripple current in the secondary current
External MOSFET Selection
The design procedure for external MOSFET selection is
identical to that outlined in the DCM Flyback section.
SEC
V
×D
MAX
INMIN
∆I
=
SEC
Secondary-Diode Selection
The design procedure for secondary-diode selection is
identical to that outlined in the DCM Flyback section.
L
× f
×K
PRI SW
For the purpose of current-limit setting, the peak current
can be calculated as follows:
Error Amplifier Compensation Design
In the CCM flyback converter, the primary inductance
and the equivalent load resistance introduces a right
half-plane zero at the following frequency:
I
= I
×1.2
LIM PRIPEAK
Primary RCD Snubber Selection
The design procedure for primary RCD snubber selection
is identical to that outlined in the DCM Flyback section.
2
(1− D
) × V
MAX
OUT
f
=
ZRHP
Output Capacitor Selection
X7R ceramic output capacitors are preferred in industrial
applications due to their stability over temperature. The
output capacitor is usually sized to support a step load
2
2× π ×D
×L
×I
×K
MAX
PRI OUT
���������������������������������������������������������������� Maxim Integrated Products 21
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
The loop compensation values are calculated as:
where is I given by:
PK
2
2×(V
− V
)×I
225×I
(1− D
f
RHP
OUT
L
IN_MIN OUT
× f
OUT
)
I
=
R
=
× 1+
PK
Z
5 × f
INMIN SW
MAX
P
where f , the pole due to output capacitor and load is
P
L
is the minimum value of the input inductor taking
INMIN
given by:
into account tolerance and saturation effects.
(1+ D
) ×I
OUT
MAX
f
=
Output Capacitor Selection
P
2× π × C
× V
OUT
OUT
The output capacitance can be calculated as follows:
The above selection of R sets the loop-gain crossover
I
× t
RESPONSE
Z
STEP
C
t
=
OUT
frequency (f , where the loop gain equals 1) equal to
C
∆V
OUT
1/5th the right half-plane zero frequency.
0.33
1
f
ZRHP
≅ (
+
)
RESPONSE
f
≤
C
f
f
SW
C
5
where I
time of the controller, DV
voltage deviation, and f is the target closed-loop
crossover frequency. f is chosen to be one-tenth of
C
the switching frequency f . For the boost converter,
is the load step, t
is the response
RESPONSE
With the control loop zero placed at the load pole
frequency:
STEP
is the allowable output
OUT
C
1
C
=
Z
2π ×R × f
Z
P
SW
the output capacitor supplies the load current when the
main switch is on; therefore, the output voltage ripple is a
function of duty cycle and load current. Use the following
equation to calculate the output capacitor ripple:
With the high-frequency pole placed at half the switching
frequency:
1
C
=
P
π ×R × f
Z
SW
I
×L ×I
IN PK
OUT
∆V
=
COUT
V
× C
OUT
DCM Boost
INMIN
In a DCM boost converter, the inductor current returns to
zero in every switching cycle. Energy stored during the
on-time of the main switch Q1 is delivered entirely to the
load in each switching cycle.
Input Capacitor Selection
The input ceramic capacitor value required can be
calculated based on the ripple allowed on the input DC
bus. The input capacitor should be sized based on the
RMS value of the AC current handled by it. The calcula-
tions are:
Inductance Selection
The design procedure starts with calculating the boost
converter’s input inductor, such that it operates in
DCM at all operating line and load conditions. The
critical inductance required to maintain DCM operation is
calculated as:
3.75×I
OUT
×(1− D
C
=
IN
V
× f
)
MAX
INMIN SWMIN
The capacitor RMS can be calculated as:
I
2
PK
× 0.4
V
− V
× V
IN_MIN
I
=
(
)
OUT
IN_MIN
CIN_RMS
2× 3
L
≤
IN
2
I
× V
× f
SW
OUT
OUT
Error Amplifier Compensation Design
The loop compensation values for the error amplifier can
now be calculated as:
where V
is the minimum input voltage.
INMIN
Peak/RMS Currents Calculation
For the purposes of setting the current limit, the peak cur-
rent in the inductor can be calculated as:
G
×G ×10
M
DC
2× π × f
C
=
= G
(
×10 nF
DC
)
Z
SW
I
= I ×1.2
where G , the DC gain of the power stage, is given as:
LIM PK
DC
���������������������������������������������������������������� Maxim Integrated Products 22
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
The RMS current in the MOSFET is useful in estimating
2
8×(V
− V
)× f
× V
×L
IN
the conduction loss, and is given as:
3
OUT
INMIN
SW
OUT
G
=
DC
2
(2V
− V
) ×I
− V
OUT
INMIN
OUT
I
×L
× f
PK
INS SW
I
=
MOSFETRMS
3× V
V
× C
×(V
)
INMIN
INMIN
OUT
OUT
OUT
R
=
Z
I
× C ×(2V
− V
)
where I
operating input voltage, V
is the peak current calculated at the lowest
OUT
Z
OUT
INMIN
PK
.
INMIN
where V
is the minimum operating input voltage,
INMIN
CCM Boost
and I
is the maximum load current.
OUT
C
×ESR
OUT
R
In a CCM boost converter, the inductor current does
not return to zero during a switching cycle. Since
the MAX17597 implements a nonsynchronous boost
converter, the inductor current will enter DCM operation
at load currents below a critical value equal to half of the
peak-peak ripple in the inductor current.
C
=
P
Z
Slope Compensation
In theory, the DCM boost converter does not require
slope compensation for stable operation. In practice, the
converter needs a minimum amount of slope for good
noise immunity at very light loads. The minimum slope is
set for the MAX17596/MAX17597 by leaving the SLOPE
pin unconnected.
Inductor Selection
The design procedure starts with calculating the boost
converter’s input inductor at nominal input voltage for
a ripple in the inductor current equal to 30% of the
maximum input current.
Output Diode Selection
The voltage rating of the output diode for the boost
converter ideally equals the output voltage of the
boost converter. In practice, parasitic inductances and
capacitances in the circuit interact to produce voltage
overshoot during the turn-off transition of the diode that
occurs when the main switch Q1 turns on. The diode
rating should therefore be selected with the necessary
margin to accommodate this extra voltage stress. A volt-
V
×D ×(1− D)
IN
L
=
IN
0.3×I
× f
OUT SW
where D is the duty cycle calculated as:
+ V − V
V
OUT
D
IN
D =
V
+ V − (R ×I
)
OUT
D
DS OUT
age rating of 1.3 x V
margin in most cases.
provides the necessary design
OUT
V
is the voltage drop across the output diode of the
D
boost converter at maximum output current, and R
the resistance of the MOSFET in the on state.
is
DS
The current rating of the output diode should be selected
so that the power loss in the diode (given as the prod-
uct of forward-voltage drop and the average diode
current) should be low enough to ensure that the junction
temperature is within limits. This necessitates the diode
Peak/RMS Current Calculation
For the purposes of setting the current limit, the peak
current in the inductor and MOSFET can be calculated
as follows:
current rating to be in the order of 2 x I
to 3 x I
.
OUT
OUT
Select fast-recovery diodes with a recovery time less than
50ns or Schottky diodes with low junction capacitance.
V
×D
×(1− D
)
I
OUT
(1− D)
OUT
MAX
MAX
I
=
+
PK
L
× f
INMIN SW
MOSFET RMS Current Calculation
The voltage stress on the MOSFET ideally equals the
sum of the output voltage and the forward drop of the
output diode. In practice, voltage overshoot and ringing
occur due to action of circuit parasitic elements during
the turn-off transition. The MOSFET voltage rating should
be selected with the necessary margin to accommodate
×1.2 for D
< 0.5
MAX
0.25× V
I
OUT
(1− D)
OUT
And, I
=
+
PK
L
× f
INMIN SW
×1.2 forD
≥ 0.5
MAX
this extra voltage stress. A voltage rating of 1.3 x V
provides the necessary design margin in most cases.
OUT
���������������������������������������������������������������� Maxim Integrated Products 23
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
D
, the maximum duty cycle, is obtained by substituting
Error Amplifier Compensation Design
The loop compensation values for the error amplifier can
now be calculated as:
MAX
the minimum input operating voltage V
equation above for duty cycle. L
in the
INMIN
is the minimum
INMIN
value of the input inductor taking into account tolerance
and saturation effects.
2
250× V
× C
×(1− D
)
MIN
OUT
I
OUT
×L
R
=
Z
OUTMIN
IN
Output Capacitor Selection
The output capacitance can be calculated as follows:
where D
is the duty cycle at the highest operating
MIN
I
× t
RESPONSE
input voltage, and I
is the minimum load current.
STEP
OUTMIN
C
t
=
OUT
∆V
OUT
V
× C
OUT OUT
2×I
C
=
Z
0.33
1
×R
Z
OUT
1
≅ (
+
)
RESPONSE
f
f
SW
C
C
=
P
π × f
×R
Z
SW
where I
time of the controller, DV
voltage deviation, and f is the target closed-loop
crossover frequency. f is chosen to be one-tenth of
is the load step, t
is the response
RESPONSE
STEP
is the allowable output
OUT
C
Slope Compensation Ramp
The slope required to stabilize the converter at duty
cycles greater than 50% can be calculated as follows:
C
the switching frequency f . For the boost converter,
SW
the output capacitor supplies the load current when the
main switch is on; therefore, the output voltage ripple is a
function of duty cycle and load current. Use the following
equation to calculate the output capacitor ripple:
0.5×(0.82× V
− V
)
OUT
IN
INMIN
S
=
V/Fs,
E
L
where L is in µH.
IN
I
×D
MAX
OUT
C
Output Diodes Selection
∆V
=
COUT
× f
The design procedure for output-diode selection is
OUT SW
identical to that outlined in the DCM Boost section.
Input Capacitor Selection
MOSFET RMS Current Calculation
The voltage stress on the MOSFET ideally equals the
sum of the output voltage and the forward drop of the
output diode. In practice, voltage overshoot and ringing
occur due to action of circuit parasitic elements during
the turn-off transition. The MOSFET voltage rating should
be selected with the necessary margin to accommodate
The input ceramic capacitor value required can be
calculated based on the ripple allowed on the input DC
bus. The input capacitor should be sized based on the RMS
value of the AC current handled by it. The calculations are:
3.75×I
OUT
C
=
IN
V
× f
×(1− D
)
MAX
INMIN SW
this extra voltage stress. A voltage rating of 1.3 x V
OUT
The input capacitor RMS current can be calculated as:
∆I
provides the necessary design margin in most cases.
The RMS current in the MOSFET is useful in estimating
the conduction loss, and is given as:
LIN
I
=
CIN_RMS
2× 3
I
× D
MAX
OUT
where:
I
=
MOSFETRMS
(1− D
)
MAX
V
×D
×(1− D
)
MAX
OUT
MAX
∆I
=
LIN
where D
is the duty cycle at the lowest operating
MAX
L
× f
INMIN SW
input voltage, and I
is the maximum load current.
OUT
forD
< 0.5,
MAX
0.25× V
OUT
∆I
=
LIN
L
× f
INMIN SW
for D
≥ 0.5
MAX
���������������������������������������������������������������� Maxim Integrated Products 24
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Typical Operating Circuits
V
IN
V
OUT
R15
R16
V
OUT
D2
D6
D4
402kI
402kI
5
T1
8
7
C9
2.2µF
50V
C19
OPEN
R1
0I
C12
1µF
R14
402kI
6
1
V
IN
PGND
C13
22µF
C14
22µF
C15
22µF
C16
22µF
R1
10
1
AC1
AC2
C5
100µF
450V
4
3
R17
100kI
R18
100kI
C10
3300pF
D1
11
L1
6.6mH
GND0
D3
R8
1.5MI
2
12
2
C1
0.1µF/
275V AC
V
IN
R7
1.5MI
PGND
GND0
C6
0.47µF
PGND
V
IN
C7
47nF
SS
R10
0I
SLOPE
V
OUT
R19
0I
R23
OPEN
NDRV
PGND
SGND
DITHER/
SYNC
N1
SGND
R9
10kI
R26
DITHER/
SYNC
8.06kI
6
RT
5
4
1
V
DRV
R11
OPEN
R28
562kI
R25
OPEN
SGND
R20
100I
DITHER/
SYNC
PGND
MAX17595
DITHER/
SYNC
V
CS
IN
2
C2
SHORT
V
FB
C18
15000pF
R27
20kI
C11
330pF
R21
0.1I
SGND
COMP
R22
1.2kI
R2
2.67MI
R12
12.1kI
C17
47pF
V
DRV
V
FB
SGND
R3
C3
SHORT
C4
OPEN
C8
R13
10kI
2.67MI
2.2µF
FB
V
2
3
DRV
1
U3
V
DRV
R24
OPEN
R4
R29
49.9kI
N.C.
N.C.
2.67MI
PGND
EN/UVLO
OVI
EN/UVLO
R5
75kI
OVI
SGND
R6
24.9kI
SGND
SGND
SGND
Figure 9. MAX17595 Typical Application Circuit (Universal Offline Isolated Power Supply)
���������������������������������������������������������������� Maxim Integrated Products 25
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Typical Operating Circuits (continued)
V
V
OUT
IN
D2
T1
V
V
OUT
IN
C1
47µF
50V
C2
4.7µF
50V
C9
22µF
16V
C10
22µF
16V
C11
22µF
16V
R1
7.5kI
C4
33nF, 50V
18V TO 36V
INPUT
5V, 1.5A
OUTPUT
D1
C3
0.22µF
50V
GND
PGND
V
IN
SS
EP
C5
47nF
NDRV
CS
SLOPE
N1
R2
SHORT
R8
100I
V
OUT
R3
C6
R9
10kI
300pF
0.5I
FB
COMP
PGND
V
FB
R12
OPEN
MAX17596
R4
15kI
V
CC
R14
1kI
R15
V
DRV
30.3kI
C12
OPEN
C13
R16
20kI
V
DRV
4.7nF
C7
2.2µF
16V
V
FB
C14
33pF
SGND
RT
V
IN
R13
511I
U2
R10
17.4kI
R5
348kI
2
R17
EN/UVLO
OVI
U3
1
3
EN/UVLO
10kI
R11
OPEN
R6
20kI
C8
SHORT
OVI
DITHER
R7
10kI
PGND
Figure 10. MAX17596 Typical Application Circuit (Power Supply for DC-DC Applications)
���������������������������������������������������������������� Maxim Integrated Products 26
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Typical Operating Circuits (continued)
V
IN
V
IN
V
IN
C1
47µF
C2
0.1µF
10.8V TO
13.2V DC
EP
PGND
SS
C3
47µF
V
DRV
C4
2.2µF
V
DRV
R1
120kI
V
IN
SLOPE
FB
L1
220µH
R2
9.92kI
MAX17597
D1
SS26-TP
V
R3
184kI
OUT
24V, 0.3A
C7
4.7µF/35V
V
OUT
NDRV
CS
N1
R4
5kI
COMP
PGND
R8
100I
C5
47nF
C6
120pF
C8
300pF
R9
0.5I
V
IN
R10
17.4kI
RT
R5
481kI
EN/UVLO
OVI
R11
OPEN
R6
25kI
C9
SHORT
DITHER
R7
49.9kI
SGND
SGND
PGND
Figure 11. MAX17597 Typical Application Circuit (Nonsynchronous Boost Converter)
���������������������������������������������������������������� Maxim Integrated Products 27
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
except for a connection at the least noisy section of the
power ground plane, typically the return of the input filter
capacitor. The negative terminal of the filter capacitor, the
ground return of the power switch and current sensing
resistor, must be close together. PCB layout also affects
the thermal performance of the design. A number of
thermal vias that connect to a large ground plane should
be provided under the exposed pad of the part for
efficient heat dissipation. For a sample layout that
ensures first-pass success, refer to the MAX17595 evalu-
ation kit layout available at www.maxim-ic.com. For
universal AC input designs, follow all applicable safety
regulations. Offline power supplies can require UL, VDE,
and other similar agency approvals.
Layout, Grounding and Bypassing
All connections carrying pulsed currents must be very
short and as wide as possible. The inductance of these
connections must be kept to an absolute minimum due to
the high di/dt of the currents in high-frequency-switching
power converters. This implies that the loop areas for
forward and return pulsed currents in various parts of the
circuit should be minimized. Additionally, small current
loop areas reduce radiated EMI. Similarly, the heatsink
of the MOSFET presents a dV/dt source; therefore,
the surface area of the MOSFET heatsink should be
minimized as much as possible.
Ground planes must be kept as intact as possible. The
ground plane for the power section of the converter
should be kept separate from the analog ground plane,
Ordering Information/Selector Guide
TEMP
RANGE
PIN
PACKAGE
UVLO, V
IN
PART
FUNCTIONALITY
D
MAX
CLAMP
20V, Yes
4V, No
MAX17595ATE+ -40NC to +125NC
MAX17596ATE+ -40NC to +125NC
MAX17597ATE+ -40NC to +125NC
16 TQFN-EP*
16 TQFN-EP*
16 TQFN-EP*
Offline Flyback Controller
Low-Voltage DC-DC Flyback Controller
Boost Controller
46%
46%
93%
4V, No
+Denotes a lead(Pb)-free/RoHS-compliant package.
*Exposed pad.
Package Information
For the latest package outline information and land patterns (footprints), go to www.maxim-ic.com/packages. Note that a “+”, “#”, or
“-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains
to the package regardless of RoHS status.
PACKAGE TYPE
PACKAGE CODE
OUTLINE NO.
21-0136
LAND PATTERN NO.
90-0032
16 TQFN
T1633+4
���������������������������������������������������������������� Maxim Integrated Products 28
MAX17595/MAX17596/MAX17597
Peak-Current-Mode Controllers for
Flyback and Boost Regulators
Revision History
REVISION REVISION
PAGES
DESCRIPTION
CHANGED
NUMBER
DATE
0
1/12
Initial release
—
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied.
Maxim reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical
Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
29
©
2012 Maxim Integrated Products
Maxim is a registered trademark of Maxim Integrated Products, Inc.
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