VCA2613Y [BB]
Dual, VARIABLE GAIN AMPLIFIER with Low-Noise Preamp; 双可变增益放大器，具有低噪声前置放大器
| 型号： | VCA2613Y |
| 厂家： | 
BURR-BROWN CORPORATION |
| 描述： | Dual, VARIABLE GAIN AMPLIFIER with Low-Noise Preamp |
| 文件： | 总16页 (文件大小：341K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
VCA2613
V
C
A
2
6
1
3
SBOS179D – DECEMBER 2000 – REVISED OCTOBER 2004
Dual, VARIABLE GAIN AMPLIFIER
with Low-Noise Preamp
FEATURES
DESCRIPTION
The VCA2613 is a dual, Low-Noise Preamplifier (LNP), plus
low-noise Variable Gain Amplifier (VGA). The combination of
Active Termination (AT) and Maximum Gain Select (MGS)
allow for the best noise performance. The VCA2613 also
features low crosstalk and outstanding distortion perfor-
mance.
● LOW NOISE PREAMP:
Low Input Noise: 1.0nV/√Hz
Active Termination Noise Reduction
Switchable Termination Value
80MHz Bandwidth
5dB to 25dB Gain
The LNP has differential input and output capability and is
strappable for gains of 5dB, 17dB, 22dB or 25dB. Low input
impedance is achieved by AT, resulting in as much as a 4.6dB
improvement in noise figure over conventional shunt termina-
tion. The termination value can also be switched to accommo-
date different sources. The output of the LNP is available for
external signal processing.
Differential In and Out
● LOW NOISE VARIABLE GAIN AMPLIFIER:
Low Noise VCA: 3.3nV/√Hz, Differential
Programming Optimizes Noise Figure
24dB to 45dB Gain
40MHz Bandwidth
Differential In and Out
The variable gain is controlled by an analog voltage whose
gain varies from 0dB to the gain set by the MGS. The ability
to program the variable gain also allows the user to optimize
dynamic range. The VCA input can be switched from the
LNP to external circuits for different applications. The output
can be used in either a single-ended or differential mode to
drive high-performance Analog-to-Digital (A/D) converters,
and is cleanly limited for optimum overdrive recovery.
● LOW CROSSTALK: 52dB at Max Gain, 5MHz
● HIGH-SPEED VARIABLE GAIN ADJUST
● SWITCHABLE EXTERNAL PROCESSING
APPLICATIONS
The combination of low noise, gain, and gain range program-
mability makes the VCA2613 a versatile building block in a
number of applications where noise performance is critical.
The VCA2613 is available in a TQFP-48 package.
● ULTRASOUND SYSTEMS
● WIRELESS RECEIVERS
● TEST EQUIPMENT
Maximum Gain Select
VCACNTL
FBCNTL
LNPOUTN
VCAINN
MGS0 MGS1 MGS2
RF2
RF1
FBSW
FB
VCA2613
(1 of 2 Channels)
Analog
Control
Maximum Gain
Select
Input
LNPIN
P
VCAOUTN
LNPGS1
LNPGS2
Programmable
Gain Amplifier
24 to 45dB
Voltage
Controlled
Attenuator
Low Noise
Preamp
5dB to 25dB
LNP
Gain Set
LNPGS3
VCAOUT
P
LNPIN
N
LNPOUT
P
VCAIN
P
SEL
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
Copyright © 2000-2004, Texas Instruments Incorporated
www.ti.com
ABSOLUTE MAXIMUM RATINGS(1)
ELECTROSTATIC
DISCHARGE SENSITIVITY
This integrated circuit can be damaged by ESD. Texas Instru-
ments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
Power Supply (+VS) ............................................................................. +6V
Analog Input ............................................................. –0.3V to (+VS + 0.3V)
Logic Input ............................................................... –0.3V to (+VS + 0.3V)
Case Temperature ......................................................................... +100°C
Junction Temperature .................................................................... +150°C
Storage Temperature ...................................................... –40°C to +150°C
NOTE: (1) Stresses above those listed under “Absolute Maximum Ratings” may
cause permanent damage to the device. Exposure to absolute maximum
conditions for extended periods may affect device reliability.
ESD damage can range from subtle performance degrada-
tion to complete device failure. Precision integrated circuits
may be more susceptible to damage because very small
parametric changes could cause the device not to meet its
published specifications.
PACKAGE/ORDERING INFORMATION(1)
PACKAGE
DESIGNATOR
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
PRODUCT
PACKAGE-LEAD
VCA2613Y
TQFP-48
PFB
"
VCA2613
VCA2613Y/250
VCA2613Y/2K
Tape and Reel, 250
Tape and Reel, 2000
"
"
"
NOTE: (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet.
ELECTRICAL CHARACTERISTICS
At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted.
The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended unless otherwise noted.
VCA2613Y
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
PREAMPLIFIER
Input Resistance
Input Capacitance
Input Bias Current
CMRR
600
15
1
50
1
112
3.5
1.0
0.35
6.2
kΩ
pF
nA
dB
VPP
mVPP
f = 1MHz, VCACNTL = 0.2V
Preamp Gain = +5dB
Preamp Gain = +25dB
Preamp Gain = +5dB
Preamp Gain = +25dB
Independent of Gain
Maximum Input Voltage
Input Voltage Noise(1)
nV/
nV/
pA/
√Hz
√Hz
√Hz
Input Current Noise
Noise Figure, RS = 75Ω, RIN = 75Ω(1)
RF = 550Ω, Preamp Gain = 22dB,
PGA Gain = 39dB
dB
Bandwidth
Gain = 22dB
80
MHz
PROGRAMMABLE VARIABLE GAIN AMPLIFIER
Peak Input Voltage
–3dB Bandwidth
Slew Rate
Output Signal Range
Differential
2
40
300
2
1
VPP
MHz
V/µs
VPP
Ω
RL ≥ 500Ω Each Side to Ground
Output Impedance
f = 5MHz
Output Short-Circuit Current
±40
–71
–63
–80
–80
6
–68
±2
2.5
mA
dBc
dBc
dBc
dBc
VPP
dB
Third Harmonic Distortion
Second Harmonic Distortion
IMD, Two-Tone
f = 5MHz, VOUT = 1VPP, VCACNTL = 3.0V
f = 5MHz, VOUT = 1VPP, VCACNTL = 3.0V
–45
–45
V
V
OUT = 2VPP, f = 1MHz
OUT = 2VPP, f = 10MHz
1dB Compression Point
Crosstalk
Group Delay Variation
DC Output Level, VIN = 0
f = 5MHz, Output Referred, Differential
VOUT = 1VPP, f = 1MHz, Max Gain Both Channels
1MHz < f < 10MHz, Full Gain Range
ns
V
ACCURACY
Gain Slope
Gain Error
10.9
dB/V
dB
±1(2)
Output Offset Voltage
Total Gain
±50
21
50
mV
dB
dB
CNTL = 0.2V
CNTL = 3.0V
18
47
24
53
GAIN CONTROL INTERFACE
Input Voltage (VCACNTL) Range
Input Resistance
0.2 to 3.0
V
MΩ
µs
1
0.2
Response Time
45dB Gain Change, MGS = 111
Operating, Both Channels
POWER SUPPLY
Operating Temperature Range
Specified Operating Range
Power Dissipation
–40
4.75
+85
5.25
495
°C
V
mW
5.0
410
NOTE: (1) For preamp driving VGA. (2) Referenced to best fit dB-linear curve.
VCA2613
2
SBOS179D
www.ti.com
PIN CONFIGURATION
48 47 46 45 44 43 42 41 40 39 38 37
VDD
A
1
2
3
4
5
6
7
8
9
36 VDD
35 NC
34 NC
B
NC
NC
VCAINNA
VCAINPA
LNPOUTNA
LNPOUTPA
SWFBA
FBA
33 VCAINNB
32 VCAINPB
31 LNPOUTNB
30 LNPOUTPB
29 SWFBB
28 FBB
VCA2613
COMP1A 10
COMP2A 11
LNPINNA 12
27 COMP1B
26 COMP2B
25 LNPINNB
13 14 15 16 17 18 19 20 21 22 23 24
PIN DESCRIPTIONS
PIN
DESIGNATOR
DESCRIPTION
PIN
DESIGNATOR
DESCRIPTION
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
VDD
NC
NC
A
Channel A +Supply
Do Not Connect
Do Not Connect
25
26
27
28
29
30
31
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
LNPINNB
COMP2B
COMP1B
FBB
SWFBB
LNPOUTPB
LNPOUTNB
VCAINPB
VCAINNB
NC
Channel B LNP Inverting Input
Channel B Frequency Compensation 2
Channel B Frequency Compensation 1
Channel B Feedback Output
Channel B Switched Feedback Output
Channel B LNP Positive Output
Channel B LNP Negative Output
Channel B VCA Positive Input
Channel B VCA Negative Input
Do Not Connect
Do Not Connect
Channel B +Analog Supply
Channel B Analog Ground
Channel B VCA Negative Output
Channel B VCA Positive Output
Maximum Gain Select 3 (LSB)
Maximum Gain Select 2
Maximum Gain Select 1 (MSB)
VCA Control Voltage
VCAINNA
VCAINPA
LNPOUTNA
LNPOUTPA
SWFBA
FBA
COMP1A
COMP2A
LNPINNA
LNPGS3A
LNPGS2A
LNPGS1A
LNPINPA
Channel A VCA Negative Input
Channel A VCA Positive Input
Channel A LNP Negative Output
Channel A LNP Positive Output
Channel A Switched Feedback Output
Channel A Feedback Output
Channel A Frequency Compensation 1
Channel A Frequency Compensation 2
Channel A LNP Inverting Input
Channel A LNP Gain Strap 3
Channel A LNP Gain Strap 2
Channel A LNP Gain Strap 1
Channel A LNP Noninverting Input
+Supply for Internal Reference
0.01µF Bypass to Ground
NC
V
DDB
GNDB
VCAOUTNB
VCAOUTPB
MGS3
MGS2
MGS1
VCACNTL
VCAINSEL
FBSWCNTL
VCAOUTPA
VCAOUTNA
GNDA
VDDR
VBIAS
VCM
GNDR
LNPINPB
LNPGS1B
LNPGS2B
LNPGS3B
0.01µF Bypass to Ground
Ground for Internal Reference
Channel B LNP Noninverting Input
Channel B LNP Gain Strap 1
Channel B LNP Gain Strap 2
Channel B LNP Gain Strap 3
VCA Input Select, HI = External
Feedback Switch Control: HI = ON
Channel A VCA Positive Output
Channel A VCA Negative Output
Channel A Analog Ground
VCA2613
SBOS179D
3
www.ti.com
TYPICAL PERFORMANCE CURVES
At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted.
The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended unless otherwise noted. This results in a 6dB reduction in signal
amplitude compared to differential operation.
OUTPUT REFERRED NOISE vs VCACNTL
GAIN vs VCACNTL
2000
1800
1600
1400
1200
1000
800
65
60
55
50
45
40
35
30
25
20
15
MGS = 111
MGS = 110
RS = 50Ω
MGS = 101
MGS = 100
MGS = 111
MGS = 011
MGS = 010
600
400
MGS = 001
MGS = 000
200
MGS = 011
0
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0
VCACNTL (V)
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0
VCACNTL (V)
INPUT REFERRED NOISE vs VCACNTL
INPUT REFERRED NOISE vs RS
20
18
16
14
12
10
8
10.0
RS = 50Ω
1.0
MGS = 111
6
4
MGS = 011
2
0
0.1
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0
VCACNTL (V)
1
10
100
1000
RS (Ω)
NOISE FIGURE vs RS
NOISE FIGURE vs VCACNTL
9
8
7
6
5
4
3
2
1
0
20
18
16
14
12
10
8
6
4
2
0
10
100
1000
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0
VCACNTL (V)
RS (Ω)
VCA2613
4
SBOS179D
www.ti.com
op amp. The VCM node shown in the drawing is the VCM
output (pin 19). Typical R and C values are shown, yielding
a high-pass time constant similar to that of the LNP. If a
different common-mode referencing method is used, it is
important that the common-mode level be within 10mV of the
VCM output for proper operation.
THEORY OF OPERATION
The VCA2613 is a dual-channel system consisting of three
primary blocks: a Low Noise Preamplifier (LNP), a Voltage
Controlled Attenuator (VCA), and a Programmable Gain
Amplifier (PGA). For greater system flexibility, an onboard
multiplexer is provided for the VCA inputs, selecting either
the LNP outputs or external signal inputs. Figure 1 shows a
simplified block diagram of the dual-channel system.
1kΩ
External
InA
To VCAIN
47nF
Input
Signal
1kΩ
Channel A
Output
Channel A
Input
LNP
VCA
PGA
VCM (+2.5V)
Maximum
Gain
Select
Analog
Control
VCA
Control
MGS
FIGURE 2. Recommended Circuit for Coupling an External
Signal into the VCA Inputs.
Channel B
Input
Channel B
Output
VCA—OVERVIEW
LNP
VCA
PGA
The magnitude of the differential VCA input signal (from the
LNP or an external source) is reduced by a programmable
attenuation factor, set by the analog VCA Control Voltage
(VCACNTL) at pin 43. The maximum attenuation factor is
further programmable by using the three MGS bits (pins 40-
42). Figure 3 illustrates this dual-adjustable characteristic.
Internally, the signal is attenuated by having the analog
VCACNTL vary the channel resistance of a set of shunt-
connected FET transistors. The MGS bits effectively adjust
the overall size of the shunt FET by switching parallel
components in or out under logic control. At any given
maximum gain setting, the analog variable gain characteris-
tic is linear in dB as a function of the control voltage, and is
created as a piecewise approximation of an ideal dB-linear
transfer function. The VCA gain control circuitry is common
to both channels of the VCA2613.
External
InB
FIGURE 1. Simplified Block Diagram of the VCA2613.
LNP—OVERVIEW
The LNP input may be connected to provide active-feedback
signal termination, achieving lower system noise perfor-
mance than conventional passive shunt termination. Even
lower noise performance is obtained if signal termination is
not required. The unterminated LNP input impedance is
600kΩ. The LNP can process fully differential or single-
ended signals in each channel. Differential signal processing
results in significantly reduced 2nd-harmonic distortion and
improved rejection of common-mode and power-supply noise.
The first gain stage of the LNP is AC-coupled into its output
buffer with a 44µs time constant (3.6kHz high-pass charac-
teristic). The buffered LNP outputs are designed to drive the
succeeding VCA directly or, if desired, external loads as low
as 135Ω with minimal impact on signal distortion. The LNP
employs very low impedance local feedback to achieve
stable gain with the lowest possible noise and distortion.
Four pin-programmable gain settings are available: 5dB,
17dB, 22dB, and 25dB. Additional intermediate gains can be
programmed by adding trim resistors between the Gain Strap
programming pins.
0
Minimum Attenuation
–24
Maximum Attenuation
The common-mode DC level at the LNP output is nominally
2.5V, matching the input common-mode requirement of the
VCA for simple direct coupling. When external signals are
fed to the VCA, they should also be set up with a 2.5VDC
common-mode level. Figure 2 shows a circuit that demon-
strates the recommended coupling method using an external
–45
0
3.0V
Control Voltage
FIGURE 3. Swept Attenuator Characteristic.
VCA2613
SBOS179D
5
www.ti.com
PGA OVERVIEW AND OVERALL DEVICE
CHARACTERISTICS
The VCA2613 includes a built-in reference, common to both
channels, to supply a regulated voltage for critical areas of
the circuit. This reduces the susceptibility to power supply
variation, ripple, and noise. In addition, separate power
supply and ground connections are provided for each chan-
nel and for the reference circuitry, further reducing interchannel
cross-talk.
The differential output of the VCA attenuator is then amplified
by the PGA circuit block. This post-amplifier is programmed
by the same MGS bits that control the VCA attenuator,
yielding an overall swept-gain amplifier characteristic in which
the VCA • PGA gain varies from 0dB (unity) to a program-
mable peak gain of 24, 27, 30, 33, 36, 39, 42, or 45dB.
Further details regarding the design, operation and use of
each circuit block are provided in the following sections.
The GAIN vs VCACNTL curve in the typical characteristics
shows the composite gain control characteristic of the entire
VCA2613. Setting VCACNTL to 3.0V causes the digital MGS
gain control to step in 3dB increments. Setting VCACNTL to 0V
causes all the MGS-controlled gain curves to converge at
one point. The gain at the convergence point is the LNP gain
less 6dB, because the measurement setup looks at only one
side of the differential PGA output, resulting in 6dB lower
signal amplitude.
LOW NOISE PREAMPLIFIER (LNP)—DETAIL
The LNP is designed to achieve a low noise figure, especially
when employing active termination. Figure 4 is a simplified
schematic of the LNP, illustrating the differential input and
output capability. The input stage employs low resistance
local feedback to achieve stable low noise, low distortion
performance with very high input impedance. Normally, low
noise circuits exhibit high power consumption due to the
large bias currents required in both input and output stages.
The LNP uses a patented technique that combines the input
and output stages such that they share the same bias
current. Transistors Q4 and Q5 amplify the signal at the gate-
source input of Q4, the +IN side of the LNP. The signal is
further amplified by the Q1 and Q2 stage, and then by the final
Q3 and RL gain stage, which uses the same bias current as
the input devices Q4 and Q5. Devices Q6 through Q10 play
the same role for signals on the –IN side.
ADDITIONAL FEATURES—OVERVIEW
Overload protection stages are placed between the attenua-
tor and the PGA, providing a symmetrically clipped output
whenever the input becomes large enough to overload the
PGA. A comparator senses the overload signal amplitude
and substitutes a fixed DC level to prevent undesirable
overload recovery effects. As with the previous stages, the
VCA is AC-coupled into the PGA. In this case, the coupling
time constant varies from 5µs at the highest gain (45dB) to
59µs at the lowest gain (25dB).
The differential gain of the LNP is given in Equation (1):
(1)
RL
Gain = 2 •
RS
COMP2A
VDD
COMP1A
RL
93Ω
RL
93Ω
To Bias
Circuitry
Q9
Q2
LNPOUTN LNPOUT
P
Buffer
Buffer
CCOMP
4.7pF
(External
Capacitor)
Q3
Q8
RS1
105Ω
RS2
34Ω
RW
RW
LNPGS1
Q4
Q7
LNPINP
LNPINN
LNPGS2
LNPGS3
RS3
17Ω
Q10
Q1
To Bias
Circuitry
Q5
Q6
FIGURE 4. Schematic of the Low Noise Preamplifier (LNP).
6
VCA2613
SBOS179D
www.ti.com
where RL is the load resistor in the drains of Q3 and Q8, and
RS is the resistor connected between the sources of the input
transistors Q4 and Q7. The connections for various RS com-
To preserve the low noise performance of the LNP, the user
should take care to minimize resistance in the input lead. A
parasitic resistance of only 10Ω will contribute 0.4nV/√Hz
.
binations are brought out to device pins LNPGS1, LNPGS2
,
and LNPGS3 (pins 13-15 for channel A, 22-24 for channel B).
These Gain Strap pins allow the user to establish one of four
fixed LNP gain options as shown in Table I.
NOISE (nV/√Hz)
LNP GAIN (dB)
Input-Referred
Output-Referred
25
22
17
5
1.54
1.59
1.82
4.07
2260
1650
1060
597
LNP PIN STRAPPING
LNP GAIN (dB)
LNPGS1, LNPGS2, LNPGS3 Connected Together
LNPGS1 Connected to LNPGS3
LNPGS1 Connected to LNPGS2
All Pins Open
25
22
17
5
TABLE II. Noise Performance for MGS = 111 and VCACNTL = 3.0V.
The LNP is capable of generating a 2VPP differential signal.
The maximum signal at the LNP input is therefore 2VPP
divided by the LNP gain. An input signal greater than this
would exceed the linear range of the LNP, an especially
important consideration at low LNP gain settings.
TABLE I. Pin Strappings of the LNP for Various Gains.
It is also possible to create other gain settings by connecting
an external resistor between LNPGS1 on one side, and
LNPGS2 and/or LNPGS3 on the other. In that case, the
internal resistor values shown in Figure 4 should be com-
bined with the external resistor to calculate the effective
value of RS for use in Equation (1). The resulting expression
for external resistor value is given in Equation (2).
ACTIVE FEEDBACK WITH THE LNP
One of the key features of the LNP architecture is the ability
to employ active-feedback termination to achieve superior
noise performance. Active-feedback termination achieves a
lower noise figure than conventional shunt termination, es-
sentially because no signal current is wasted in the termina-
tion resistor itself. Another way to understand this is as
follows: Consider first that the input source, at the far end of
the signal cable, has a cable-matching source resistance of
RS. Using conventional shunt termination at the LNP input, a
second terminating resistor of value RS is connected to
ground. Therefore, the signal loss is 6dB due to the voltage
divider action of the series and shunt RS resistors. The
effective source resistance has been reduced by the same
factor of 2, but the noise contribution has been reduced by
only the √2, only a 3dB reduction. Therefore, the net theoreti-
cal SNR degradation is 3dB, assuming a noise-free amplifier
input. (In practice, the amplifier noise contribution will de-
grade both the unterminated and the terminated noise fig-
ures, somewhat reducing the distinction between them.)
2RS1RL + 2RFIXRL – Gain • RS1RFIX
REXT
=
(2)
Gain• RS1 – 2RL
where REXT is the externally selected resistor value needed
to achieve the desired gain setting, RS1 is the fixed parallel
resistor in Figure 4, and RFIX is the effective fixed value of the
remaining internal resistors: RS2, RS3, or (RS2 || RS3) depend-
ing on the pin connections.
Note that the best process and temperature stability will be
achieved by using the pre-programmed fixed gain options of
Table I, since the gain is then set entirely by internal resistor
ratios, which are typically accurate to ±0.5%, and track quite
well over process and temperature. When combining exter-
nal resistors with the internal values to create an effective RS
value, note that the internal resistors have a typical tempera-
ture coefficient of +700ppm/°C and an absolute value toler-
ance of approximately ±5%, yielding somewhat less predict-
able and stable gain settings. With or without external resis-
tors, the board layout should use short Gain Strap connec-
tions to minimize parasitic resistance and inductance effects.
See Figure 5 for an amplifier using active feedback. This
diagram appears very similar to a traditional inverting ampli-
fier. However, the analysis is somewhat different because
the gain A in this case is not a very large open-loop op amp
gain; rather it is the relatively low and controlled gain of the
LNP itself. Thus, the impedance at the inverting amplifier
terminal will be reduced by a finite amount, as given in the
familiar relationship of Equation (3):
The overall noise performance of the VCA2613 will vary as
a function of gain. Table II shows the typical input- and
output-referred noise densities of the entire VCA2613 for
maximum VCA and PGA gain; i.e., VCACNTL set to 3.0V and
all MGS bits set to 1. Note that the input-referred noise
values include the contribution of a 50Ω fixed source imped-
ance, and are therefore somewhat larger than the intrinsic
input noise. As the LNP gain is reduced, the noise contribu-
tion from the VCA/PGA portion becomes more significant,
resulting in higher input-referred noise. However, the output-
referred noise, which is indicative of the overall SNR at that
gain setting, is reduced.
RF
RIN
=
(3)
1+ A
(
)
where RF is the feedback resistor (supplied externally be-
tween the LNPINP and FB terminals for each channel), A is
the user-selected gain of the LNP, and RIN is the resulting
amplifier input impedance with active feedback. In this case,
unlike the conventional termination above, both the signal
voltage and the RS noise are attenuated by the same factor
VCA2613
SBOS179D
7
www.ti.com
VCA NOISE = 3.8nV√Hz, LNP GAIN = 20dB
LNP Noise
14
12
10
8
RF
nV/√Hz
6.0E-10
8.0E-10
1.0E-09
1.2E-09
1.4E-09
1.6E-09
1.8E-09
2.0E-09
RS
LNPIN
A
6
RIN
RIN
Active Feedback
4
RF
=
= RS
2
1 + A
0
RS
0
100 200 300 400 500 600 700 800 900 1000
Source Impedance (Ω)
A
RS
FIGURE 7. Noise Figure for Conventional Termination.
A switch, controlled by the FBSWCNTL signal on pin 45,
enables the user to reduce the feedback resistance by
adding an additional parallel component, connected between
the LNPINP and SWFB terminals. The two different values of
feedback resistance will result in two different values of
active-feedback input resistance. Thus, the active-feedback
impedance can be optimized at two different LNP gain
settings. The switch is connected at the buffered output of
the LNP and has an ON resistance of approximately 1Ω.
Conventional Cable Termination
FIGURE 5. Configurations for Active Feedback and Conven-
tional Cable Termination.
of two (6dB) before being re-amplified by the A gain setting.
This avoids the extra 3dB degradation due to the square-root
effect described above, the key advantage of the active
termination technique.
When employing active feedback, the user should be careful
to avoid low-frequency instability or overload problems. Fig-
ure 8 illustrates the various low-frequency time constants.
Referring again to the input resistance calculation of Equa-
tion (3), and considering that the gain term A falls off below
3.6kHz, it is evident that the effective LNP input impedance
will rise below 3.6kHz, with a DC limit of approximately RF. To
avoid interaction with the feedback pole/zero at low frequen-
cies, and to avoid the higher signal levels resulting from the
rising impedance characteristic, it is recommended that the
external RFCC time constant be set to about 5µs.
As mentioned above, the previous explanation ignored the
input noise contribution of the LNP itself. Also, the noise
contribution of the feedback resistor must be included for a
completely correct analysis. The curves given in Figures 6
and 7 allow the VCA2613 user to compare the achievable
noise figure for active and conventional termination methods.
The left-most set of data points in each graph give the results
for typical 50Ω cable termination, showing the worst noise
figure but also the greatest advantage of the active feedback
method.
RF
VCA NOISE = 3.8nV√Hz, LNP GAIN = 20dB
9
VCM
LNP Noise
nV/√Hz
8
CF
7
6
5
4
3
2
1
0
6.0E-10
8.0E-10
1.0E-09
1.2E-09
1.4E-09
1.6E-09
1.8E-09
2.0E-09
0.001µF
1MΩ
44pF
44pF
CC
Buffer
Buffer
LNPOUT
N
RS
LNPOUTP
Gain
Stage
1MΩ
0
100 200 300 400 500 600 700 800 900 1000
VCM
(VCA) LNP
Source Impedance (Ω)
FIGURE 6. Noise Figure for Active Termination.
FIGURE 8. Low Frequency LNP Time Constants.
VCA2613
8
SBOS179D
www.ti.com
Achieving the best active feedback architecture is difficult
with conventional op amp circuit structures. The overall gain
A must be negative in order to close the feedback loop, the
input impedance must be high to maintain low current noise
and good gain accuracy, but the gain ratio must be set with
very low value resistors to maintain good voltage noise.
Using a two-amplifier configuration (noninverting for high
impedance plus inverting for negative feedback reasons)
results in excessive phase lag and stability problems when
the loop is closed. The VCA2613 uses a patented architec-
ture that achieves these requirements, with the additional
benefits of low power dissipation and differential signal han-
dling at both input and output.
associated with the input connection. Equation 4 relates the
bandwidth to the various impedances that are connected to
the LNP.
A + 1 R + R
(
)
I
F
BW =
(4)
2πC(RI)(RF)
AVOIDING UNSTABLE PERFORMANCE
The VCA2612 and the VCA2613 are very similar in perfor-
mance in all respects, except in the area of noise perfor-
mance. See Figure 4 for a schematic of the LNP. The
improvement in noise performance is because the input
wiring resistor (RW) of the VCA2613, see Figure 4, has been
considerably reduced compared to the VCA2612. This brings
the input noise of the VCA2613 down to 1.0nV/√Hz com-
pared to VCA2612’s 1.25nV/√Hz. The input impedance at
the gate of either Q4 or Q7 can be approximated by the
network shown in Figure 11. The resistive component shown
in Figure 11 is negative, which gives rise to unstable behav-
ior when the signal source resistance has both inductive and
capacitive elements. It should be noted that this negative
resistance is not a physical resistor, but an equivalent resis-
tance that is a function of the devices shown in Figure 4.
Normally, when an inductor and capacitor are placed in
series or parallel, there is a positive resistance in the loop
that prevents unstable behavior.
For greatest flexibility and lowest noise, the user may wish to
shape the frequency response of the LNP. The COMP1 and
COMP2 pins for each channel (pins 10 and 11 for channel A,
pins 26 and 27 for channel B) correspond to the drains of Q3
and Q8, see Figure 4. A capacitor placed between these pins
will create a single-pole low-pass response, in which the
effective R of the RC time constant is approximately 186Ω.
COMPENSATION WHEN USING ACTIVE
FEEDBACK
The typical open-loop gain versus frequency characteristic
for the LNP is shown in Figure 9. The –3dB bandwidth is
approximately 180MHz and the phase response is such that
when feedback is applied the LNP will exhibit a peaked
response or might even oscillate. One method of compensat-
ing for this undesirable behavior is to place a compensation
capacitor at the input to the LNP, as shown in Figure 10. This
method is effective when the desired –3dB bandwidth is
much less than the open-loop bandwidth of the LNP. This
compensation technique also allows the total compensation
capacitor to include any stray or cable capacitance that is
24pF
–93Ω
57pF
–3dB Bandwidth
FIGURE 11. VCA2613 Input Impedance.
25dB
For the VCA2613, the situation can be remedied by placing
an external resistor with a value of approximately 15Ω or
higher in series with the input lead. The net series resistance
will be positive, and there will be no observed instability.
Although this technique will prevent oscillations, it is not
recommended, as it will also increase the input noise. A
4.7pF external capacitor must be placed between pins
COMP2A (pin 11) and LNPINPA (pin 16), and between pins
COMP2B (pin 26) and LNPINPB (pin 21). This has the result
of making the input impedance always capacitive due to the
feedback effect of the compensation capacitor and the gain
of the LNP. Using capacitive feedback, the LNP becomes
unconditionally stable, as there is no longer a negative
component to the input impedance. The compensation
capacitor mentioned above will be reflected to the input by
the formula:
180MHz
FIGURE 9. Open-Loop Gain Characteristic of LNP.
RF
RI
Input
C
A
Output
CIN = (A + 1)CCOMP
(5)
FIGURE 10. LNP with Compensation Capacitor.
VCA2613
SBOS179D
9
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The capacitance that is determined in Equation 5 should be
added to the capacitance shown in Equation 4 to determine
the overall bandwidth of the LNP. The LNPINNA (pin 12) and
the LNPINNB (pin 25) should be bypassed to ground by the
shortest means possible to avoid any inductance in the lead.
In addition to the analog VCACNTL gain setting input, the
attenuator architecture provides digitally programmable ad-
justment in eight steps, via the three Maximum Gain Setting
(MGS) bits. These adjust the maximum achievable gain
(corresponding to minimum attenuation in the VCA, with
VCACNTL = 3.0V) in 3dB increments. This function is accom-
plished by providing multiple FET sub-elements for each of
the Q1 to Q10 FET shunt elements (see Figure 12). In the
simplified diagram of Figure 13, each shunt FET is shown as
two sub-elements, QNA and QNB. Selector switches, driven by
the MGS bits, activate either or both of the sub-element FETs
to adjust the maximum RON and thus achieve the stepped
attenuation options.
LNP OUTPUT BUFFER
The differential LNP output is buffered by wideband class AB
voltage followers which are designed to drive low impedance
loads. This is necessary to maintain LNP gain accuracy,
since the VCA input exhibits gain-dependent input imped-
ance. The buffers are also useful when the LNP output is
brought out to drive external filters or other signal processing
circuitry. Good distortion performance is maintained with
buffer loads as low as 135Ω. As mentioned previously, the
buffer inputs are AC coupled to the LNP outputs with a
3.6kHz high-pass characteristic, and the DC common mode
level is maintained at the correct VCM for compatibility with
the VCA input.
The VCA can be used to process either differential or single-
ended signals. Fully differential operation will reduce 2nd-
harmonic distortion by about 10dB for full-scale signals.
Input impedance of the VCA will vary with gain setting, due
to the changing resistances of the programmable voltage
divider structure. At large attenuation factors (i.e., low gain
settings), the impedance will approach the series resistor
value of approximately 135Ω.
VOLTAGE-CONTROLLED ATTENUATOR (VCA)—DETAIL
As with the LNP stage, the VCA output is AC coupled into the
PGA. This means that the attenuation-dependent DC com-
mon-mode voltage will not propagate into the PGA, and so
the PGA’s DC output level will remain constant.
The VCA is designed to have a dB-linear attenuation charac-
teristic, i.e. the gain loss in dB is constant for each equal
increment of the VCACNTL control voltage. See
Figure 1 for a block diagram of the VCA. The attenuator is
essentially a variable voltage divider consisting of one series
input resistor, RS, and ten identical shunt FETs, placed in
parallel and controlled by sequentially activated clipping
amplifiers. Each clipping amplifier can be thought of as a
specialized voltage comparator with a soft transfer character-
istic and well-controlled output limit voltages. The reference
voltages V1 through V10 are equally spaced over the 0V to
3.0V control voltage range. As the control voltage rises
through the input range of each clipping amplifier, the ampli-
fier output will rise from 0V (FET completely ON) to VCM –VT
(FET nearly OFF), where VCM is the common source voltage
and VT is the threshold voltage of the FET. As each FET
approaches its OFF state and the control voltage continues
to rise, the next clipping amplifier/FET combination takes
over for the next portion of the piecewise-linear attenuation
characteristic. Thus, low control voltages have most of the
FETs turned ON, while high control voltages have most
turned OFF. Each FET acts to decrease the shunt resistance
of the voltage divider formed by RS and the parallel FET
network.
Finally, note that the VCACNTL input consists of FET gate
inputs. This provides very high impedance and ensures that
multiple VCA2613 devices may be connected in parallel with
no significant loading effects. The nominal voltage range for
the VCACNTL input spans from 0V to 3V. Over driving this
input (≤ 5V) does not affect the performance.
OVERLOAD RECOVERY CIRCUITRY—DETAIL
With a maximum overall gain of 70dB, the VCA2613 is prone
to signal overloading. Such a condition may occur in either
the LNP or the PGA depending on the various gain and
attenuation settings available. The LNP is designed to pro-
duce low-distortion outputs as large as 1VPP single-ended
(2VPP differential). Therefore the maximum input signal for
linear operation is 2VPP divided by the LNP differential gain
setting. Clamping circuits in the LNP ensure that larger input
amplitudes will exhibit symmetrical clipping and short recov-
ery times. The VCA itself, being basically a voltage divider,
is intrinsically free of overload conditions. However, the PGA
post-amplifier is vulnerable to sudden overload, particularly
at high gain settings. Rapid overload recovery is essential in
many signal processing applications such as ultrasound
imaging. A special comparator circuit is provided at the PGA
input which detects overrange signals (detection level de-
pendent on PGA gain setting). When the signal exceeds the
The attenuator is comprised of two sections, with five parallel
clipping amplifier/FET combinations in each. Special refer-
ence circuitry is provided so that the (VCM –VT) limit voltage
will track temperature and IC process variations, minimizing
the effects on the attenuator control characteristic.
VCA2613
10
SBOS179D
www.ti.com
Attenuator
Input
A1 - A10 Attenuator Stages
QS
Attenuator
Output
RS
Q1
A2
Q2
A3
Q3
A4
Q4
A5
Q5
Q6
A7
Q7
A8
Q8
A9
Q9
A10
Q10
VCM
A1
A6
C1
V1
C2
V2
C3
V3
C4
V4
C5
V5
C6
V6
C7
V7
C8
V8
C9
V9
C10
V10
Control
Input
C1 - C10 Clipping Amplifiers
0dB
–4.5dB
Attenuation Characteristic of Individual FETs
VCM-VT
0
V1
V2
V3
V4
V5
V6
V7
V8
V9
V10
Characteristic of Attenuator Control Stage Output
OVERALL CONTROL CHARACTERISTICS OF ATTENUATOR
0dB
–4.5dB
0.3V
3V
Control Signal
FIGURE 12. Piecewise Approximation to Logarithmic Control Characteristics.
VCA2613
SBOS179D
11
www.ti.com
RS
OUTPUT
Q5B
INPUT
VCM
Q1A
Q1B
Q2A
Q2B
Q3A
Q3B
Q4A
Q4B
Q5A
A1
A2
A3
A4
A5
B1
B2
PROGRAMMABLE ATTENUATOR SECTION
FIGURE 13. Programmable Attenuator Section.
comparator input threshold, the VCA output is blocked and
an appropriate fixed DC level is substituted, providing fast
and clean overload recovery. The basic architecture is shown
in Figure 14. Both high and low overrange conditions are
sensed and corrected by this circuit.
VCACNTL = 3.0V, DIFFERENTIAL, MGS = 100, (36dB)
From VCA
Output
PGA
Comparators
Gain = A
Selection
Logic
200ns/div
FIGURE 16. Overload Recovery Response For Maximum Gain.
INPUT OVERLOAD RECOVERY
One of the most important applications for the VCA2613 is
processing signals in an ultrasound system. The ultrasound
signal flow begins when a large signal is applied to a
transducer, which converts electrical energy to acoustic
energy. It is not uncommon for the amplitude of the electrical
signal that is applied to the transducer to be ±50V or greater.
To prevent damage, it is necessary to place a protection
circuit between the transducer and the VCA2613, as shown
in Figure 17. Care must be taken to prevent any signal from
turning the ESD diodes on. Turning on the ESD diodes inside
the VCA2613 could cause the input coupling capacitor (CC)
to charge to the wrong value.
E = Maximum Peak Amplitude
E
A
E
A
–
FIGURE 14. Overload Protection Circuitry.
Figures 15 and 16 show typical overload recovery wave-
forms with MGS = 100, for VCA + PGA minimum gain (0dB)
and maximum gain (36dB), respectively. LNP gain is set to
25dB in both cases.
VCACNTL = 0.2V, DIFFERENTIAL, MGS = 100, (0dB)
Output
VDD
CF
RF
Input
LNPOUTN
LNPINP
Protection
Network
LNP
200ns/div
ESD Diode
FIGURE 15. Overload Recovery Response For Minimum Gain.
FIGURE 17. VCA2613 Diode Bridge Protection Circuit.
VCA2613
12
SBOS179D
www.ti.com
PGA POST-AMPLIFIER—DETAIL
The PGA architecture consists of a differential, program-
mable-gain voltage to current converter stage followed by
transimpedance amplifiers to create and buffer each side of
the differential output. The circuitry associated with the volt-
age to current converter is similar to that previously de-
scribed for the LNP, with the addition of eight selectable PGA
gain-setting resistor combinations (controlled by the MGS
bits) in place of the fixed resistor network used in the LNP.
Low input noise is also a requirement of the PGA design due
to the large amount of signal attenuation which can be
inserted between the LNP and the PGA. At minimum VCA
attenuation (used for small input signals) the LNP noise
dominates; at maximum VCA attenuation (large input sig-
nals) the PGA noise dominates. Note that if the PGA output
is used single-ended, the apparent gain will be 6dB lower.
Figure 18 shows a simplified circuit diagram of the PGA
block. As described previously, the PGA gain is programmed
with the same MGS bits which control the VCA maximum
attenuation factor. Specifically, the PGA gain at each MGS
setting is the inverse (reciprocal) of the maximum VCA
attenuation at that setting. Therefore, the VCA + PGA overall
gain will always be 0dB (unity) when the analog VCACNTL
input is set to 0V (= maximum attenuation). For VCACNTL
=
3V (no attenuation), the VCA + PGA gain will be controlled
by the programmed PGA gain (24 to 45 dB in 3dB steps). For
clarity, the gain and attenuation factors are detailed in Table
III.
MGS
ATTENUATOR GAIN DIFFERENTIAL ATTENUATOR +
SETTING VCACNTL = 0V to 3V
PGA GAIN
DIFF. PGA GAIN
000
001
010
011
100
101
110
111
–24dB to 0dB
–27dB to 0dB
–30dB to 0dB
–33dB to 0dB
–36dB to 0dB
–39dB to 0dB
–42dB to 0dB
–45dB to 0dB
24dB
27dB
30dB
33dB
36dB
39dB
42dB
45dB
0dB to 24dB
0dB to 27dB
0dB to 30dB
0dB to 33dB
0dB to 36dB
0dB to 39dB
0dB to 42dB
0dB to 45dB
TABLE III. MGS Settings.
VDD
To Bias
Circuitry
Q1
Q11
Q12
Q9
RL
RL
VCAOUT
P
VCAOUTN
Q3
Q8
VCM
VCM
RS1
RS2
Q13
Q4
Q7
+In
–In
Q14
Q2
Q10
Q5
Q6
To Bias
Circuitry
FIGURE 18. Simplified Block Diagram of the PGA section within the VCA2613.
VCA2613
SBOS179D
13
www.ti.com
PACKAGE OPTION ADDENDUM
www.ti.com
9-Dec-2004
PACKAGING INFORMATION
Orderable Device
Status (1)
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
TQFP
TQFP
Drawing
VCA2613Y/250
VCA2613Y/2K
ACTIVE
ACTIVE
PFB
48
48
250
None
None
CU SNPB
CU SNPB
Level-2-220C-1 YEAR
Level-2-220C-1 YEAR
PFB
2000
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - May not be currently available - please check http://www.ti.com/productcontent for the latest availability information and additional
product content details.
None: Not yet available Lead (Pb-Free).
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens,
including bromine (Br) or antimony (Sb) above 0.1% of total product weight.
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
Addendum-Page 1
MECHANICAL DATA
MTQF019A – JANUARY 1995 – REVISED JANUARY 1998
PFB (S-PQFP-G48)
PLASTIC QUAD FLATPACK
0,27
0,17
0,50
M
0,08
36
25
37
24
48
13
0,13 NOM
1
12
5,50 TYP
7,20
SQ
Gage Plane
6,80
9,20
SQ
8,80
0,25
0,05 MIN
0°–7°
1,05
0,95
0,75
0,45
Seating Plane
0,08
1,20 MAX
4073176/B 10/96
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Falls within JEDEC MS-026
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
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