ADL5593 [ADI]

Correcting Imperfections in IQ Modulators to Improve RF Signal Fidelity; 校正IQ调制器缺陷，提高射频信号保真度


型号：	ADL5593
厂家：	ADI
描述：	Correcting Imperfections in IQ Modulators to Improve RF Signal Fidelity 校正IQ调制器缺陷，提高射频信号保真度射频
文件：	总8页 (文件大小：209K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

AN-1039

APPLICATION NOTE

One Technology Way • P. O. Box 9106 • Norwood, MA 02062-9106, U.S.A. • Tel: 781.329.4700 • Fax: 781.461.3113 • www.analog.com

Correcting Imperfections in IQ Modulators to Improve RF Signal Fidelity

by Eamon Nash

changes, in-factory and in-field algorithms that can reduce the

effect of these modulator imperfections is also discussed, and

particular focus is placed on the efficacy of in-factory set-and-

forget algorithms.

INTRODUCTION

The in-phase and quadrature modulator (IQ modulator) is a

key component in modern wireless transmitters. It provides a

convenient method for modulating data bits or symbols onto

an RF carrier. IQ upconversion has become the architecture

of choice for implementing transmitter signal chains for end

applications such as cellular, WiMAX, and wireless point-to-

point. IQ modulators, however, can degrade signal fidelity in

ways that are somewhat unique. These effects can degrade the

quality of the transmitted signal during the modulation process,

resulting in degraded error vector magnitude (EVM) at the

receiver, which in turn degrades bit error rate (BER). Fortu-

nately, algorithms exist that can correct these imperfections.

A TYPICAL WIRELESS TRANSMITTER

Figure 1 shows a block diagram of a direct-conversion wireless

transmitter that uses an IQ modulator to modulate a bit stream

onto a carrier. A single bit stream is split into two parallel bit

streams at half the original data rate. To limit the spectral band-

width of the final carrier, the two bit streams are low-pass filtered

in the digital domain. To do this, the original bit-streams must

be digitally oversampled by the digital signal processor or field

programmable gate array (FPGA). So, instead of two bit streams,

there are now two streams of digital words. The chosen resolu-

tion of these words depends upon multiple factors such as the

required signal-to-noise ratio of the link and the chosen mod-

ulation scheme (QPSK in this case). Word widths between 12

and 16 bits are commonly chosen.

This application note describes a typical zero-IF or direct-

conversion transmitter and provides a brief introduction to

digital modulation. Other items discussed are: the imper-

fections introduced by the modulator are examined with

particular focus on the effect of temperature and frequency

AD9788

FPGA OR DSP

AUX

DAC1

GAIN

DAC 1

ADL5375

100Ω

800

600

OVERSAMPLE

LOW-PASS

FILTER

LOW-

PASS

FILTER

DIGITAL

FILTER

PHASE

ADJUST

16-BIT

I DAC

400

200

–200

–400

–600

–800

800

600

400

200

50Ω

–200

–400

–600

–800

HPA

ADL5320

0°

800

600

400

200

90°

–200

–400

–600

–800

800

600

AUX

DAC2

LOW-

PASS

FILTER

1.0

1.5

2.0

400

TIME (msec)

200

AD8363

–200

–400

–600

–800

100Ω

50dB RMS

800

600

400

200

GAIN

DAC 2

DETECTOR

1.329

1.829

2.329

TIME (msec)

–200

–400

–600

–800

OVERSAMPLE

LOW-PASS

FILTER

DIGITAL

FILTER

PHASE

ADJUST

16-BIT

Q DAC

1.0

1.5

TIME (msec)

2.0

50Ω

SPECTRUM

ANALYZER

DIGITAL

DEMOD

AD9230

AD8352

Figure 1. A Zero IF Direct-Conversion Transmitter with Optional Loop-Back Receiver

Rev. 0 | Page 1 of 8

AN-1039

Application Note

TABLE OF CONTENTS

Introduction ...................................................................................... 1

Correcting for Quadrature and I/Q Gain Errors ......................6

Frequency Variations ....................................................................7

Post Calibration Temperature Drift............................................7

Calibration vs. Time......................................................................7

Complex Modulation....................................................................8

Conclusions....................................................................................8

A Typical Wireless Transmitter ...................................................... 1

Modulator Imperfections ............................................................ 3

Correcting Modulator Imperfections ........................................ 4

Factory Calibration ...................................................................... 4

Calibration Procedure.................................................................. 6

Rev. 0 | Page 2 of 8

Application Note

AN-1039

After low-pass filtering, the two word streams are applied to a

pair of digital-to-analog converters (DAC). The DAC outputs

drive two low-pass filters whose primary role is to remove

Nyquist images. The outputs of these filters then drive the

baseband inputs of the IQ modulator. The local oscillator (LO)

input of the modulator is driven by a relatively pure CW signal

generated by a phase-locked loop (PLL) such as the ADF4106

from Analog Devices, Inc. Now, take a closer look at the

operation of the IQ modulator.

MODULATOR IMPERFECTIONS

Contrary to the previous hypothetical situation, in a real IQ

modulator, things do not look so perfect. A series of effects in

the IQ modulator conspire to create QPSK (or QAM) vectors

that are neither equal in amplitude nor separated by exactly 45°.

Consider first what happens if for some reason the gain of the I

path is greater than that of the Q channel; this could be caused

by a DAC gain mismatch, low-pass filter insertion loss, mismatch,

or gain imbalance inside the IQ modulator. Regardless of where

this gain imbalance comes from, its effect is the same. Because

the 0°/180° vectors at the output of the I multiplier are larger

than the +90°/−90° vectors from the Q multiplier, the shape

of the constellation becomes rectangular (see Figure 2B). This

degrades signal integrity at the receiver because the receiver is

expecting a perfectly square constellation. In the QPSK example

shown in Figure 2B, a slight gain imbalance is unlikely to result

in an incorrect bit decision in the receiver unless the received

signal is very small. However, in higher order modulation

schemes such as 16 QAM or 64 QAM (see Figure 2E and

Figure 2F), the increased density of the constellation points

could easily combine with an IQ gain imbalance to produce

an incorrect symbol decision in the receiver.

The LO signal is split into two signals, equal in amplitude but

with a phase difference of exactly 90°. These two quadrature

signals drive the inputs of the two mixers that, for the purposes

of this application note, are viewed as analog multipliers. The

outputs of these two multipliers are added together (in the

Σ block of the IQ modulator) to provide the IQ modulator’s

output.

While it is apparent that the baseband data streams have

been filtered, instead briefly consider them as the original bit

streams. Instead of a stream of 1s and 0s, think of them as two

streams switching between a value of +1 and –1. So, the output

of the I multiplier consists of a vector which is flipping in-phase

between 0° and 180°as the bit stream alternates. Likewise, the

output of the Q multiplier is a vector that flips between +90°

and –90° as the bit stream modulates the original 90° vector.

Thus, if at a particular instant, both the I and Q bit streams are

equal to +1, the result at the output of the IQ modulator is the

sum of the 90° and 0° vectors, that is, a +45° vector. Likewise,

I and Q bit combinations of −1/+1, −1/−1, and +1/−1 produce

vectors (commonly called symbols) all of equal amplitude at

+135°, −135°, and −45°, respectively. If these vectors were

plotted, observe the constellation of the modulated carrier

(see Figure 2A).

In most IQ modulators, the 90° phase split of the LO is achieved

using either a polyphase filter or a divide-by-two flip-flop circuit

(which requires an external LO that is twice the desired output

frequency). In either circuit, the 90° phase split or quadrature is

never perfect. For example, if there is a 1° quadrature error, the

shape of the resulting constellation is slightly trapezoidal (see

Figure 2C). Just like IQ gain imbalance, this can result in

incorrect bit decisions in the receiver.

Now consider what happens if either the I or Q paths have

unwanted dc offset errors. This results in the +1/−1 multipli-

cation being skewed. For example, an offset that is equal to 1%

of the baseband signal amplitude causes the +1/−1 multipliers

to be modified to +1.01/−0.99. This has the effect of shifting

the center of the constellation off the origin, on either the I or

Q axis, most likely in both (see Figure 2D). In the frequency

domain, this manifests itself as a small portion of the unmodu-

lated carrier appearing at the output of the modulator. In the

frequency domain, this LO leakage (also referred to as LO

feedthrough) appears at the center of the modulated spectrum.

(A)

(B)

(C)

Because of parasitic capacitances within the silicon die and

bond-wire to bond-wire coupling, the signal that is applied

to the LO port of the IQ modulator may also couple directly

to the RF output. This leakage is independent of the offset

multiplication effect that was described previously. However,

its manifestation, that is, the presence of the unmodulated

carrier in the output spectrum, is exactly the same. Thus, the

net LO leakage seen at the output of the IQ modulator is the

vector sum of these two components. Fortunately, as discussed

in the Correcting Modulator Imperfections section, the com-

posite LO leakage at the output can be mitigated by a single

compensation technique.

(D)

(E)

(F)

Figure 2. Error Vector Magnitude Constellations that Result from Various

Modulator Imperfections

Rev. 0 | Page 3 of 8

AN-1039

Application Note

Table 1. IQ Modulator Selection Table Showing Uncompensated Gain and Phase Imbalance

IQ 3dB

CARRIER

SIDEBAND

GAIN

PHASE

NOISE

SUPPLY

PART

NUMBER

FREQUENCY BANDWIDTH SUPPRESS SUPPRESSION IMBALANCE IMBALANCE

FLOOR

P1dB

OUTPUT

VOLTAGE

(V)

CURRENT SPECS @

(MHz)

(dBm)

(dBc)

(dB)

(°)

(dBm/Hz) (dBm) IP3 (dBm)

(mA)

(MHz)

AD8345

AD8346

AD8349

140 TO 1000

800 TO 2500

700 TO 2700

0.50

1.00

2.70 TO 5.50

4.75 TO 5.50

4.75 TO 5.25

4.75 TO 5.50

800

1900

900

–42

–45

–50

–45

–32

–33

–46

–38

–44

–39

–42

–36

–35

–41

–55

–45

–57

–50

–52

–50

–46

–50

–48

0.200

0.100

0.030

0.100

0.090

0.010

0.015

–0.050

–0.030

0.050

–155.0

–147.0

–155.0

–160.0

–158.6

–158.0

–157.0

–160.0

–159.0

–160.0

–157.0

2.5

–3.0

7.6

N/A

1.90

21.0

24.0

27.0

26.0

22.8

27.0

26.0

25.0

29.0

30.0

135

205

175

165

174

173

200

215

230

170

160

500

750

700

250

ADL5370 300 TO 1000

ADL5371 500 TO 1500

ADL5372 1500 TO 2500

ADL5373 2300 TO 3000

ADL5374 3000 TO 4000

ADL5375 400 TO 6000

ADL5385 50 TO 2200

ADL5386 50 TO 2200

ADL5590 869 TO 960

ADL5591 1805 TO 1990

11.0

14.4

14.2

13.8

12.0

9.4

450

0.76

900

–0.03

0.21

1900

2500

3500

900

0.10

0.25

–0.29

–0.39

–0.50

0.20

11.0

11.1

16.0

350

940

1960

0.30

CORRECTING MODULATOR IMPERFECTIONS

FACTORY CALIBRATION

Note that in Figure 1, in addition to the direct conversion signal

chain, an optional loop-back or transmit observation receiver

has also been incorporated into the radio. The primary function

of this receiver is to analyze the adjacent channel power ratio

(ACPR) of the transmitter that is primarily caused by distortion

in the high power amplifier (HPA). By continually observing

the ACPR of the transmitter, digital predistortion of the

baseband signal can be employed to partially correct HPA

nonlinearities while allowing the HPA to operate closer to its

compression point.

If a wireless transmitter does not use digital predistortion, it

would be difficult to justify the cost of a loop-back receiver

purely for the sake of the IQ modulator. In such situations,

the two options that remain are:

•

Do not perform any correction of the IQ modulator’s

imperfections.

•

Complete a one-time factory calibration and store the

correction coefficients in nonvolatile memory.

In recent years, the performance of IQ modulators has

improved to such a degree that it is now feasible (depending

on the modulation scheme) to design a transmitter without any

need to provide correction for imperfections. For example, the

ADL5375 from Analog Devices has gain and quadrature imbal-

ances of 0.05 dB and 0.29°, respectively, at 900 MHz, with little

or no degradation over temperature. As a result, in many appli-

cations, it may be adequate to dispense with any correction

algorithms. Table 1 shows the performance of this and other

members of the Analog Devices IQ modulator family.

The presence of a loop-back receiver can be opportunistically

used to also correct for modulator imperfections.

A detailed discussion of the IQ modulator correction tech-

niques used when a loop-back receiver is present is beyond the

scope of this application note. However, the general procedure

involves using the loop-back receiver to demodulate the I and

Q bit streams. The demodulated constellation is then examined

for evidence of IQ gain imbalance, imperfect quadrature, and/

or LO leakage. Once these imperfections have been identified,

the I and Q data streams can be preprocessed so that the IQ

modulator imperfections cancel out. For example, if the

demodulated constellation from the loop-back receiver shows

a rectangular constellation with the width of I being larger than

the height of Q (see Figure 2B), gain adjust registers in the DAC

can be used to either decrease the size of the I data stream or

increase the size of the Q data stream. Likewise, the phase

adjustment registers of the DAC can be used to slightly skew

the I and Q channels so that the imperfect quadrature of the

IQ modulator’s phase splitter is compensated.

Rev. 0 | Page 4 of 8

Application Note

AN-1039

The second alternative presented previously is to perform

factory calibration. To do this, the popular single sideband

spectrum can be used as a simple but valuable diagnosis

tool during factory calibration. To create a single sideband

spectrum, the I and Q inputs are driven by low frequency

(typically 1 MHz) sine and cosine signals, that is, the base-

band signals are in quadrature. Figure 3 shows the spectrum

that results when these baseband signals are mixed with the LO.

The primary components of the single sideband spectrum are:

Figure 4 shows a plot that can be used to relate sideband

suppression to I/Q gain mismatch and quadrature mismatch.

From the plot, it can be noted that a quadrature phase error of

1°, coupled with an I/Q gain mismatch of 0.5 dB, results in

−30 dB of sideband suppression. It is notable in this example

that improving the quadrature phase mismatch has no effect

on the sideband suppression unless the gain mismatch is also

improved.

–10

•

Lower sideband: If the IQ modulator has no imperfections,

this is the only spectral component observed, the result of

multiplication and summing of the baseband sine and

cosine signals with the two in-quadrature LO signals.

Undesired upper sideband: This undesired component

results from gain and phase imbalances between the I and

Q signal paths along with LO quadrature imbalance.

Undesired LO leakage: As discussed earlier, the LO leakage

results from I and Q offsets and/or parasitic leakage of the

LO directly to the IQ modulator’s output.

2.5dB

–20

1.25dB

0.5dB

–30

–40

–50

–60

–70

–80

0.25dB

0.125dB

•

0.05dB

0.025dB

0.0125dB

0dB

–90

–10

–20

–30

–40

–50

–60

–70

–80

–90

0.01

0.1

100

PHASE ERROR (Degrees)

Figure 4. Plot Showing the Relationship Between Modulator Errors and

Sideband Suppression

By using a directional coupler and a power splitter (as shown

in Figure 1), it is quite simple to add an auxiliary output to

the transmitter that can be used during factory calibration. A

spectrum analyzer is connected to this port. Another alternative

would be to connect the spectrum analyzer at the antenna

connector (after the signal has been adequately attenuated).

CENTER 899.9334MHz

333kHz/

SPAN 3.33MHz

Figure 3. Single Sideband Spectrum

Rev. 0 | Page 5 of 8

AN-1039

Application Note

CALIBRATION PROCEDURE

CORRECTING FOR QUADRATURE AND I/Q GAIN

ERRORS

Correcting all of the modulator’s imperfections is a multistep

process. Start by looking at the procedure for LO leakage cor-

rection which results in a constellation that is offset from the

origin. A single sideband spectrum is applied to the transmitter

and is monitored on the spectrum analyzer. Next, small diffe-

rential offset voltages are applied to the I and Q inputs. Applying

differential offset voltages to the I and Q inputs should not be

confused with changing the dc bias levels (also referred to as the

common-mode level) on these pins, which has no effect. This is

done as an I offset sweep followed by a Q offset sweep (or vice

versa). Returning briefly to Figure 1, note that the AD9788 (a

16-bit, 800 MSPS dual DAC) conveniently includes two aux-

iliary DACs that can be used to couple differential dc offset

voltages on I and Q lines. This coupling is performed externally

using resistor dividers.

A similar procedure can be used to correct quadrature and I/Q

gain mismatch. IQ modulator family data sheets typically specify

the quadrature phase mismatch and I/Q gain imbalance in

degrees and decibels, respectively, along with the sideband

suppression (also in decibels). Using this information, it is

advisable to perform the first optimization pass on the weaker

of the two specifications, that is, the specification which most

contributes to the sideband suppression. For example, assume

that the device data sheet specifies a sideband suppression of

−40 dBc, comprising of 1 degree of phase imbalance and 0.1 dB

of gain imbalance amplitude. In this case, it is advisable to first

try to adjust phase because making a gain adjustment has

almost no effect as long as the 1 degree of phase error is present

(see Figure 4).

Figure 5 shows how sweeping the I and Q offset voltages alters

the LO leakage. Start by sweeping the I offset voltage around 0 V

while holding the Q offset voltage at 0 V. With modern IQ

modulators exhibiting unadjusted LO leakage in the −40 dBm

range and having voltage gains in the −5 dB to +5 dB range, an

offset voltage sweep range of 5 mV is more than adequate to

identify the location of the null (in this example, 2 mV is

adequate to identify a nulling voltage somewhere between

100 μV and 200 μV). Note, however, that the first pass (black

trace) only manages to reduce the LO leakage to just under

−40 dBm. This clearly indicates that the Q offset needs

correction. The second pass (blue trace) involves sweeping

the Q offset around 0 V with the I offset held at the value that

yielded the first I null. Note that a Q offset of 400 μV reduces

the LO leakage a further 10 dB to around −50 dBm. However,

a third pass is required. The trough from the first pass is quite

shallow because the Q channel had not yet been adjusted. This

makes it difficult to identify the ideal I nulling voltage. A third

pass (red trace) that involves again sweeping the I offset while

holding the Q offset at 400 μV, identifies the optimum I nulling

voltage to be 150 μV.

Figure 6 shows the results of a gain sweep followed by a

phase sweep. In the first pass, the gain delta between I and Q

is adjusted over a range of approximately 2 dB. The TxDAC®

in Figure 1 facilitates this adjustment by providing internal

gain adjust auxiliary DACs. The sweep yields a null of around

−57 dBc for a gain difference of approximately −0.1 dB (gain is

scaled on the top axis). Next, adjust the skew between I and Q.

This drives the null down further to −60 dBc for a phase adjust

of −0.05°.

GAIN ADJUST (dB)

–2

–10

–1

–20

–30

–40

–50

–60

–70

FIRST PASS (GAIN ADJUST)

SECOND PASS (PHASE ADJUST)

–10

–20

–30

–40

–50

–60

–70

–80

FIRST PASS – I OFFSET ADJUST

SECOND PASS – Q OFFSET ADJUST

THIRD PASS – I OFFSET ADJUST

–0.4

–0.3

–0.2

–0.1

0.1

0.2

0.3

0.4

PHASE ADJUST (Degrees)

Figure 6. Multipass Sideband Suppression Compensation Sweeps

In this case, a third pass is not necessary and does not yield

further improvement. This stems from the fact that the

unadjusted phase error is very close to the optimized value

(~0.05°). Thus, the first-pass gain adjust yields a deep trough

that is only slightly improved during the phase sweep. This

contrasts to the LO leakage nulling where a third pass yielded

further improvement.

–2

–1

I AND Q DIFFERENTIAL OFFSET VOLTAGES (mV)

Figure 5. Multipass LO Leakage Compensation Sweeps

Rev. 0 | Page 6 of 8

Application Note

AN-1039

Once the LO leakage and quadrature error have been calibrated,

all that remains is for the calibration coefficients to be stored in

nonvolatile RAM so that they are available when the equipment

is turned on in the field. To recap, the four calibration

coefficients are

POST CALIBRATION TEMPERATURE DRIFT

Factory calibration at multiple temperatures is even more

difficult and expensive than calibration at multiple frequencies.

As a result, it is generally only practical to perform factory

adjustment of LO leakage and sideband suppression at an

ambient temperature. Thus, what happens to post-calibration

performance as temperature varies?

•

I channel offset voltage

Q channel offset voltage

I channel vs. Q channel gain imbalance

Quadrature phase imbalance

In Figure 8, the LO leakage and sideband suppression have

again been nulled midband. After nulling, the device is cycled

over temperature. This again has the effect of moving sideband

suppression and LO leakage off their nulled levels. However,

notice that the performance at temperature is quite flat across

frequency and it is no longer clear at which frequency the

nulling was performed. The net improvement over temperature

is approximately 15 dB compared to the unadjusted LO leakage.

FREQUENCY VARIATIONS

Calibrating at multiple frequencies within a band adds time to

the factory calibration, requires more nonvolatile memory for

the larger look up table, and is more cumbersome during field

operation as calibration coefficients have to be swapped out as

the frequency changes.

CALIBRATION VS. TIME

Now, consider what happens to the quality of calibration as the

frequency changes. In Figure 7, sideband suppression and LO

leakage have been nulled to −60 dBc and −74 dBm, respectively,

at 1900 MHz. Figure 7 also shows how the uncompensated

sideband suppression and LO leakage vary with frequency (the

flatter green and red traces a the top of the plot). Next, adjust

the frequency over a range of 30 MHz (the typical width of a

cellular telephony band) without recalibration. The LO leakage

quickly loses its null and at some frequencies is only around

8 dB better than the uncompensated value. In the case of the

sideband suppression, the difference between the compensated

and uncompensated values becomes as low as around 1 dB.

Figure 7 suggests that factory calibration be performed at

multiple frequencies within a band to maintain nulled

performance across the band.

In the set-and-forget factory calibration scheme that has just

been described, the question of long-term drift arises because

the equipment may never be recalibrated in the field. Experi-

ments have shown that it is very difficult, if not impossible, to

measure the degradation of nulled sideband suppression and

LO leakage over time. Very mild changes in environmental

conditions tend to quickly move the device off its null. This

makes it impossible to determine whether the environment and

the test equipment are altering the experiment or if genuine

device drift over time is taking place.

However, Figure 8 shows that the question of drift over time

is less important. This is because the effect of temperature drift

is much more significant. Thus, in a system that experiences

reasonable temperature fluctuations, whatever drift over time

takes place is completely masked by the temperature drift.

–30

UNADJUSTED SIDEBAND SUPPRESSION (dBc)

–35

UNADJUSTED LO LEAKAGE (dBm)

–35

–40

–45

–50

–55

–60

–40

–45

–50

–55

–60

–65

POST (MIDBAND)

NULLING SIDEBAND

SUPPRESSION (dBc)

–70

–75

–80

+25°C UNADJUSTED

+25°C NULLED

+85°C UNADJUSTED

+85°C NULLED

–40°C UNADJUSTED

–70

POST (MIDBAND) NULLING LO LEAKAGE (dBm)

–75

–40°C NULLED

1870

1880

1890

1900

1910

1920

1930

–80

1802

OUTPUT FREQUENCY (MHz)

1812

1822

1832

1842

1852

1862

1872

1882

LO FREQUENCY (MHz)

Figure 7. Variation of LO Leakage and Sideband Suppression vs. Frequency

after Nulling Midband

Figure 8. Variation of LO Leakage vs. Frequency and Temperature After

Nulling Midband

Rev. 0 | Page 7 of 8

AN-1039

Application Note

COMPLEX MODULATION

CONCLUSIONS

While a detailed discussion is beyond the scope of this article,

it is worth mentioning that all of the issues associated with

modulator imperfections can be avoided with a slightly differ-

ent transmit architecture. Many modern DACs incorporate

complex modulators, that is, digital engines that convert

baseband I and Q data up to a low intermediate frequency (IF).

These signals, which are still in Cartesian I and Q format, drive

the IQ modulator. Because modern IQ modulators, such as

the ADL5375, have baseband input bandwidths of as high as

750 MHz, low IFs in the 100 MHz to 250 MHz range can be

easily accommodated. When an IQ modulator is driven by

such a signal, the output spectrum is essentially a single

sideband spectrum similar to what is shown in Figure 3.

While modern IQ modulators offer excellent out-of-the-box

quadrature accuracy, IQ gain imbalance, and LO leakage, their

performance can be improved further using calibration. If the

transmitter incorporates a loop-back receiver as part of a digital

predistortion scheme, the receiver can also be used to conti-

nuously monitor and correct the imperfections of the IQ

modulator. The post-calibration performance is only limited

by the available compensation step sizes and the ability of the

receiver to precisely measure the constellation degradation.

In transmitters that do not contain a loop-back receiver, factory

calibration is a reasonable alternative. A single calibration in the

middle of a operating band most likely causes degradation at

the band edges. As a result, calibration at multiple frequencies

within a band is more effective. When temperature drift is

factored in, factory calibration at the ambient temperature

typically improves LO leakage and sideband suppression by

around 10 dB to 15 dB.

The lower sideband becomes the modulated carrier and is

displaced from the LO by a frequency offset equal to the

intermediate frequency. The imperfections of the IQ modulator

now manifest themselves as out-of-band effects, which can be

filtered away, resulting in in-band EVM, which is not affected

by the IQ modulator’s imperfections.

However, this approach comes at some cost. Care must be taken

to filter out the LO leakage along with the undesired upper

sideband. In contrast, a Nyquist filtered zero IF spectrum is

completely free of spurious components apart from harmonics

of the LO. In addition, as the frequency of the low IF increases,

the distortion of the DAC and IQ modulator increases slightly.

registered trademarks are the property of their respective owners.

AN08383-0-10/09(0)

Rev. 0 | Page 8 of 8

ADL5593 [ADI]

相关型号：

ADL5601

ADL5601-EVALZ

ADL5601ARKZ

ADL5601ARKZ-R7

ADL5602

ADL5602-EVALZ

ADL5602ARKZ-R7

ADL5604

ADL5604-EVALZ

ADL5604ACPZ-R7

ADL5605

ADL5605-EVALZ