UC3855A/B [TI]
ADVANCED PFC/PWW COMBINATION CONTROLLER WITH TRAILING-EDGE/TRAILING-EDGE MODULATION; 后缘/后缘调制高级PFC / PWW组合控制器
| 型号： | UC3855A/B |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | ADVANCED PFC/PWW COMBINATION CONTROLLER WITH TRAILING-EDGE/TRAILING-EDGE MODULATION |
| 文件： | 总32页 (文件大小：553K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
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FEATURES
DESCRIPTION
The UCC28521 and UCC28528 combination
PFC/PWM controllers provide complete control
functionality for any off-line power system
requiring compliance with the IEC1000−3−2
harmonic reduction requirements. By combining
the control and drive signals for the PFC and the
PWM stages into a single device, significant
performance and cost benefits are gained.
D
D
D
D
D
Provides Control of PFC and PWM Power
Stages In One Device
Trailing-Edge PFC, Trailing-Edge PWM
Modulation
Built-In Sequencing of PFC and PWM
Turn-On
Based on the UCC28511, the new devices employ
an average current mode control architecture with
input voltage feedforward. The major difference of
2-A Source and 3-A Sink Gate Drive for Both
PFC and PWM Stages
the UCC28521/28 is the trailing-edge
/
Typical 16-ns Rise Time and 7-ns Fall Time
into 1-nF Loads
trailing-edge (TEM/TEM) modulation scheme for
the PFC/PWM stages. The UCC28528 differs
from the UCC28521 as its PWM stage does not
turn off with the falling bulk voltage. The
UCC28528 PWM was designed for low power
auxiliary supplies.
PFC Features
− Average-Current-Mode Control for
Continuous Conduction Mode Operation
− Highly-Linear Multiplier for Near-Unity
Power Factor
− Input Voltage Feedforward Implementation
− Improved Load Transient Response
− Accurate Power Limiting
These devices offer performance advantages
over earlier generation combination controllers. A
key PWM feature is programmable maximum duty
cycle. For the PFC stage, the devices feature an
improved multiplier and the use of
a
− Zero Power Detect
transconductance amplifier for enhanced
transient response.
PWM Features
The core of the PFC section is in a three-input
multiplier that generates the reference signal for
the line current. The UCC28521/28 features a
highly linearized multiplier circuit capable of
producing a low distortion reference for the line
current over the full range of line and load
conditions. A low-offset, high-bandwidth current
error amplifier ensures that the actual inductor
current (sensed through a resistor in the return
path) follows the multiplier output command
signal. The output voltage error is processed
through a transconductance voltage amplifier.
− Peak-Current-Mode Control Operation
− 1:1 PFC:PWM Frequency Option
− Programmable maximum Duty Cycle Limit
Up to 90%
− Programmable Soft-Start
APPLICATIONS
D
D
High-Efficiency Server and DesktopSupplies
High-Efficiency Telecom AC-DC Converter
AVAILABLE APPLICATION OPTIONS
PART NUMBER
PWM STAGE
APPLICATION
UCC28521
UCC28528
PWM shuts off at 71% of the nomonal bulk voltage
PWM does not turn off with falling bulk voltage
ac−dc and main dc−dc converter
ac−dc and standby converter
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Copyright 2005, Texas Instruments Incorporated
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ꢡ ꢥ ꢢ ꢡꢚ ꢛꢯ ꢝꢜ ꢠ ꢨꢨ ꢦꢠ ꢞ ꢠ ꢟ ꢥ ꢡ ꢥ ꢞ ꢢ ꢪ
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1
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
DESCRIPTION (CONTINUED)
The transient response of the circuit is enhanced by allowing a much faster charge/discharge of the voltage
amplifier output capacitance when the output voltage falls outside a certain regulation window. A number of
additional features such as UVLO circuit with selectable hysteresis levels, an accurate reference voltage for the
voltage amplifier, zero power detect, OVP/enable, peak current limit, power limiting, high-current output gate
driver characterize the PFC section.
The PWM section features peak current mode control (with a ramp signal available to add slope compensation),
programmable soft-start, accurate maximum duty cycle clamp, peak current limit and high-current output gate
driver. The PWM stage is suppressed until the PFC output has reached 90% of its programmed value during
startup. During line dropout and turn off, the UCC28521 allows the PWM stage to operate until the PFC output
has dropped to 71% of its nominal bulk output value. The UCC28528 on the other hand allows the PWM stage
to operate continuously until the bias falls below the UVLO turn-off threhold.
Both devices are available in 20-pin DW package.
SIMPLIFIED APPLICATION DIAGRAM
PRIMARY
SECONDARY
RECT
+
VOUT
−
D1
−
VAC
+
BIAS
−
UCC28521/28
11 PWRGND GT2 10
REF
12 GT1
VCC
ISENSE2
VERR
9
8
7
6
5
4
3
2
1
13 SS2
14 PKLMT
15 CAOUT
16 ISENSE1
17 MOUT
18 IAC
GND
Z
CT_BUFF
D_MAX
VSENSE
RT
PWM
V−LOOP
19 VFF
20 VREF
VAOUT
Z
REF
Z
2
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
ABSOLUTE MAXIMUM RATINGS
†}
over operating free-air temperature (unless otherwise noted)
Supply voltage VCC
Idle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 V
Operating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V
Gate drive current (GT1, GT2)
Continuous . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.4 A
Pulsed
Sourcing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −2.5 A
Sinking . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.5 A
Maximum GT1, GT2 voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.5 V to VCC+0.3 V
Input voltage
VSENSE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 V to 11 V
D_MAX, SS2, CAOUT, ISENSE1, MOUT, VFF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.5 V to VREF+0.3 V
VAOUT, CT_BUFF, ISENSE2, PKLMT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.5 V to 6 V
Pin Current
RT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.5 mA
VFF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −5 mA
CT_BUFF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 mA
VAOUT, VERR, ISENSE2, SS2, CAOUT, IAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 mA
Maximum pin capacitance
ISENSE2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220 pF
0
0
0
0
0
Operating junction temperature range, T . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −55 C to 150 C
J
Storage temperature range, T . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65 C to 150 C
stg
Lead temperature 1.6mm (1/16 inch from case for 10 seconds) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300 C
†
‡
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied.
Exposure to absolute−maximum−rated conditions for extended periods may affect reliability.
Currents are positive into, negative out of the specified terminal. All voltages are referenced to GND.
ELECTROSTATIC DISCHARGE (ESD) PROTECTION
PARAMETER
MAX
2.5
0.5
UNITS
Human body model
CDM
kV
AVAILABLE OPTIONS{}
PACKAGED
DEVICES
OPTIONS
PFC:PWM
FREQUENCY
RATIO
UVLO
TURN-ON (V)
UVLO
HYSTERESIS (V)
PWM UVLO2
TURN-OFF (V)
PWM UVLO2
HYSTERESIS (V)
SOIC W−20
(DW)
1:1
1:1
10.2
10.2
0.5
0.5
5.30
−
1.45
−
UCC28521DW
UCC28528DW
†
‡
The DW package is available taped and reeled. Add R suffix to device type (e.g. UCC28512DWR) to order quantities of 2,000 devices per reel.
All devices are rated from −40°C to +105°C.
AVAILABLE OPTIONS{
THERMAL
THERMAL
IMPEDANCE
JUNCTION TO
CASE
T
< 25°C
A
T
< 25°C
T < 105°C
A
IMPEDANCE
JUNCTION TO
AMBIENT
A
DERATING
FACTOR
PACKAGE
POWER RATING
POWER RATING
8-pin plastic SOIC
(DW)
84.3 °C/W
16.4 °C/W
1.5 W
84.3 mW/°C
0.5 W
†
Based on low−K PCB ∅ LFM.
3
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
ELECTRICAL CHARACTERISTICS
T = –40°C to 105°C for the UCC2851x, T = T , VCC = 12 V, R = 156 kΩ, R
= 10 kΩ
A
A
J
T
CT_BUFF
(unless otherwise noted)
supply current
PARAMETER
TEST CONDITIONS
MIN
TYP
100
MAX
150
6
UNITS
µA
Supply current, off
Supply current, on
VCC turn-on threshold −300 mV
no load on GT1 or GT2
4
mA
undervoltage lockout (UVLO)
PARAMETER
TEST CONDITIONS
MIN
9.7
TYP
MAX
10.8
10.6
0.8
UNITS
VCC turn-on threshold
VCC turn-off threshold
UVLO hysteresis
UCC28521
10.2
9.7
UCC28521
UCC28521
9.1
0.3
V
0.5
voltage amplifier
PARAMETER
TEST CONDITIONS
25°C
MIN
TYP
MAX
7.61
7.65
300
UNITS
7.39
7.35
7.50
7.50
100
60
Input voltage
bias current
V
Over temperature
V
V
= V
REF
nA
dB
SENSE
SENSE
2 V ≤ VAOUT ≤ 4 V
Open loop gain
50
5.3
0.00
70
High-level output voltage
Low-level output voltage
I
I
I
= –150 µA
= 150 µA
5.5
5.6
0.15
130
LOAD
LOAD
VAOUT
V
0.05
100
−3.5
3.5
g
M
conductance
= −20 µA to 20 µA
µS
Maximum source current
Maximum sink current
−1
mA
1
PFC stage overvoltage protection and enable
PARAMETER
TEST CONDITIONS
MIN
VREF
TYP
VREF
MAX
UNITS
V
VREF
Overvoltage reference window
+ 0.440 + 0.490 + 0.540
Hysteresis
300
1.7
500
1.9
0.2
600
2.1
0.3
mV
Enable threshold
Enable hysteresis
V
0.08
4
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
ELECTRICAL CHARACTERISTICS
T = –40°C to 105°C for the UCC2851x, T = T , VCC = 12 V, R = 156 kΩ, R
= 10 kΩ
A
A
J
T
CT_BUFF
(unless otherwise noted)
current amplifier
PARAMETER
Input offset voltage
TEST CONDITIONS
MIN
TYP
MAX
5
UNITS
V
V
V
V
= 0 V,
= 0 V,
= 0 V,
= 0 V,
V
V
V
= 3 V
= 3 V
= 3 V
–5
0
mV
CM
CM
CM
CM
CAOUT
CAOUT
CAOUT
Input bias current
−50
25
−100
100
nA
dB
Input offset current
Open loop gain
2 V ≤ V
≤ 5 V
90
80
5.6
0
CAOUT
Common−mode rejection ratio
High-level output voltage
Low-level output voltage
0 V ≤ V
CM
≤ 1.5 V, V = 3 V
CAOUT
I
= –500 µA
= 500 µA
6.3
0.2
2.0
7.0
0.5
LOAD
LOAD
V
I
(1)
Gain bandwidth product
See Note 1
MHz
oscillator
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
230
1%
UNITS
f
, PWM frequency, initial accuracy
T
= 25°C
170
−1%
160
4.5
200
kHz
PWM
A
Frequency, voltage stability
Frequency, total variation
dc-to-dc ramp peak voltage
10.8 V ≤ V
CC
≤ 15 V
Line, Temp
240
5.5
kHz
V
5.0
4.0
(1)
dc-to-dc ramp amplitude voltage
(peak-to-peak)
PFC ramp peak voltage
4.5
3.5
5.0
4.0
5.5
4.5
PFC ramp amplitude voltage (peak-to-peak)
voltage reference
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
7.61
7.65
15
UNITS
25°C
7.39
7.35
7.50
7.50
5
V
V
Input voltage
Over temperature
I = −1 mA to −6 mA
Load regulation
Line regulation
REF
10.8 V ≤ V
mV
mA
≤ 15 V
1
10
CC
VREF = 0V
Short circuit current
−20
–25
−50
peak current limit
PARAMETER
PKLMT reference voltage
PKLMT propagation delay
TEST CONDITIONS
MIN
TYP
MAX
20
UNITS
mV
–20
150
0
PKLMT to GT1
300
500
ns
multiplier
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
−9
UNITS
I
I
I
I
I
, high-line low-power output current
, high-line high-power output current
, low-line low-power output current
, low-line high-power output current
, IAC-limited output current
I
I
I
I
I
I
I
I
I
I
= 500 µA, VFF = 4.7 V, VAOUT = 1.25 V
= 500 µA, VFF = 4.7 V, VAOUT = 5 V
= 150 µA, VFF = 1.4 V, VAOUT = 1.25 V
= 150 µA, VFF = 1.4 V, VAOUT = 5 V
= 150 µA, VFF = 1.3 V, VAOUT = 5 V
= 300 µA, VFF = 2.8 V, VAOUT = 2.5 V
= 150 µA, VFF = 1.4 V, VAOUT = 0.25 V
= 500 µA, VFF = 4.7 V, VAOUT = 0.25 V
= 500 µA, VFF = 4.7 V, VAOUT = 0.5 V
= 150 µA, VFF = 1.4 V, VAOUT = 5 V
−3
−75
–6
–90
–15
–290
–290
1
MOUT
MOUT
MOUT
MOUT
MOUT
AC
AC
AC
AC
AC
AC
AC
AC
AC
AC
−110
−50
−10
µA
−245
−245
0.8
−330
−330
1.2
Gain constant (k)
1/V
µA
µA
µA
µW
0
–0.2
–0.2
–0.2
−462
0
I
, zero current
MOUT
0
Power limit (I
MOUT
× V )
FF
−343
–406
1. Ensured by design. Not 100% tested in production.
5
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
ELECTRICAL CHARACTERISTICS
T = –40°C to 105°C for the UCC2851x, T = T , VCC = 12 V, R = 156 kΩ, R
= 10 kΩ
A
A
J
T
CT_BUFF
(unless otherwise noted)
zero power
PARAMETER
TEST CONDITIONS
MIN
0.20
40
TYP
0.33
90
MAX
0.50
140
UNITS
V
Zero power comparator threshold
Zero power comparator hysteresis
Measured on VAOUT,
Measured on VAOUT,
falling edge
rising edge
mV
PFC gate driver
PARAMETER
TEST CONDITIONS
≤ −200 mA
MIN
TYP
5
MAX
12
UNITS
GT1 pull-up resistance
GT1 pull-down resistance
GT1 output rise time
GT1 output fall time
−100 mA ≤ ∆I
OUT
Ω
I
= 100 mA
2
10
OUT
16
7
25
C
= 1 nF,
R
= 10 Ω
LOAD
ns
LOAD
15
Maximum duty cycle
Minimum duty cycle (trailing edge)
93%
1%
95%
−
100%
7%
See Note 1
PWM stage undervoltage lockout (UVLO2)
PARAMETER
TEST CONDITIONS
MIN
TYP
6.75
5.3
MAX
UNITS
PWM turn-on reference
PWM turn-off threshold
Hysteresis
UCC28521
UCC28521
UCC28521
6.30
7.30
V
1.16
1.45
1.74
PWM stage soft-start
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
SS2 charge current
V
V
= 7.5 V,
SS2 = 0 V
SS2 = 2.5 V,
–7.0
–10.5
–14.0
µA
SENSE
= 2.5 V,
SENSE
SS2 discharge current
Input voltage (VERR)
UCC28521
UCC28521
6
10
14
mA
mV
(UVLO2 = Low, ENABLE = High)
I
= 2 mA,UVLO2 = Low
300
VERR
PWM stage duty cycle clamp
PARAMETER
TEST CONDITIONS
MIN
70%
TYP
MAX
UNITS
Maximum duty cycle
D_MAX = 4.15 V
75%
80%
PWM stage pulse-by-pulse current sense
PARAMETER
TEST CONDITIONS
= 0 V, measured on VERR
MIN
1.35
TYP
MAX
UNITS
Current sense comparator offset voltage
I
1.50
1.65
V
SENSE2
1. Ensured by design. Not 100% tested in production.
6
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ELECTRICAL CHARACTERISTICS
T = –40°C to 105°C for the UCC2851x, T = T , VCC = 12 V, R = 156 kΩ, R
= 10 kΩ
A
A
J
T
CT_BUFF
(unless otherwise noted)
PWM stage overcurrent limit
PARAMETER
TEST CONDITIONS
MIN
1.15
TYP
1.30
50
MAX
UNITS
V
Peak current comparator threshold voltage
1.45
(1)
Input bias current
nA
PWM stage gate driver
PARAMETER
TEST CONDITIONS
MIN
TYP
5
MAX
12
UNITS
Ω
GT2 pull-up resistance
GT2 pull-down resistance
GT2 output rise time
GT2 output fall time
−100 mA ≤ ∆I
≤ −200 mA
OUT
I
= 100 mA
2
10
Ω
OUT
16
7
25
ns
C
= 1 nF,
R
= 10 Ω
LOAD
LOAD
15
ns
1. Ensured by design. Not 100% tested in production.
TERMINAL FUNCTIONS
TERMINAL
I/O
DESCRIPTION
NAME
NO.
Stage
Output of the current control amplifier of the PFC stage. CAOUT is internally connected
to the PWM comparator input in the PFC stage
CAOUT
15
PFC
O
O
I
Internally buffered PWM stage oscillator ramp output, typically used to program slope
compensation with a single resistor
CT_BUFF
D_MAX
5
4
PWM
PWM
Positive input to set the maximum duty cycle clamp level of the PWM stage duty ratio
can be between 0.09 and 0.90.
GND
6
−
−
O
O
I
Analog ground
GT1
12
10
18
16
8
PFC
PWM
PFC
PFC
PWM
PFC stage gate drive output
GT2
PWM stage gate drive output
IAC
Multiplier current input that is proportional to the instantaneous rectified line voltage
Non-inverting input to the PFC stage current amplifier
Input for PWM stage current sense and peak current limit
ISENSE1
ISENSE2
I
I
PFC multiplier high−impedance current output, internally connected to the current am-
plifier inverting input
MOUT
17
PFC
I/O
PKLMT
PWRGND
RT
14
11
2
PFC
−
I
−
I
Voltage input to the PFC peak current limit comparator
Power ground for GT1, GT2 and high current return paths
Oscillator programming pin that is set with a single resistor to GND
Soft start for the PWM stage
−
SS2
13
PWM
I
Output of the PFC transconductance voltage amplifier and it is internally connected to
the Zero Power Detect comparator input and the multiplier input
VAOUT
VCC
1
9
7
PFC
−
I/O
I
I
Positive supply voltage pin
Feedback error voltage input for the PWM stage, typically connected to an optocoupler
output
VERR
PWM
Voltage feedforward pin for the PFC stage, sources an IAC/2 current that should be
externally filtered
VFF
19
20
3
PFC
−
I
O
I
VREF
VSENSE
Precision 7.5-V reference output
Inverting input to the PFC transconductance voltage amplifier, and input to the OVP,
ENABLE and UVLO2 comparators
PFC
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BLOCK DIAGRAM
OSC
CLK2
CT_BUFF
D_MAX
4
VERR ISENSE2
SS2
VCC
9
5
7
8
13
6.75 V
UVLO2
1.9 V
PWM STAGE
SOFT START
7.5 V
REFERENCE
+
ENABLE
20 VREF
+
RT
2
3 V
UVLO
PFC:PWM
Frequency
10.2 V, 9.7 V
+
VCC
1:1 = I
1:2 = 0.5I
UCC28521 only
RT
I
RT
RT
D_MAX
COMP
I
PWM
1.3 V
LIMIT
10 GT2
R
+
Q
R
1.5 V
S
+
PWM
+
CLK
CLK
CLK
PWM
PFC
PWM
PFC
PFCOVP
8.0 V
+
ZERO
POWER
VAOUT
1
3
g
VOLTAGE
0.33 V
MULT
M
ERROR AMP
+
VCC
VSENSE
X
+
÷
X
7.5 V
+
12 GT1
2
)
VFF 19
(V
FF
CURRENT
AMP
Q
CLK
S
MIRROR
2:1
11 PWRGND
14 PKLMT
R
R
PFC
I
+
PWM
LIMIT
IAC 18
MOUT 17
+
16
ISENSE1
15
6
GND
CAOUT
DETAILED PIN DESCRIPTIONS
CAOUT (Pin 15): This is the output of a wide-bandwidth operational amplifier that senses line current and
commands the PFC stage PWM comparator to force the correct duty cycle. This output can swing above the
PFC ramp peak voltage to command maximum duty cycle and below the PFC ramp voltage to force zero duty
cycle when necessary. Connect current loop compensation components between CAOUT and MOUT.
CT_BUFF (Pin 5): The 4-V amplitude oscillator ramp is internally buffered at this pin to allow a resistor to be
connected directly from this pin to ISENSE2 for slope compensation. The internal buffer can drive a typical
500-µA resistive load at this pin.
D_MAX (Pin 4): Program the maximum duty cycle at GT2 by applying a dc voltage to this pin. Between 0.09
and 0.90, the maximum duty ratio is linearly related to D_MAX. Usually, this voltage is set with a precision
resistor divider powered by VREF. A first order approximation, with the CT_BUFF frequency near 200 kHz, is
estimated by:
V
* 1
DX
4
D
^
MAX
where, D
V
is a dimensionless ratio
is the voltage at D_MAX in volts
MAX
DX
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DETAILED PIN DESCRIPTIONS (CONTINUED)
The maximum duty ratio is modestly dependent on the switching frequency. A more accurate estimate of the
maximumǒduty cycle that is valid over the full range of switching frequencies (65 kHz to 600 kHz) is given by:
ǓV
*8
*8
+ 0.26 * ǒ4.4 10
Ǔ
D
f
) 6.9 10
f * 0.31
MAX
S
DX
S
where, f is the oscillator frequency measured at CT_BUFF in Hz
S
This pin can also be used to set D
to 0 by setting V
less than 0.7 V.
MAX
DX
GND (Pin 6): Signal ground for the integrated circuit. All voltages measured with respect to ground are
referenced to this pin. The bypass capacitors for VCC and VREF should connect to this pin with as little lead
length as possible. PWRGND must be externally connected to this pin. For best results, use a single small circuit
trace to electrically connect between the circuits that use the GND return path and the circuits that use the
PWRGND return path.
GT1 (Pin 12): A 2-A peak source and 3-A peak sink current totem pole MOSFET gate driver for the PFC stage.
Some overshoot at GT1 can be expected when driving a capacitive load, but adding a minimal series resistor
of about 2 Ω between GT1 and the external MOSFET gate can reduce this overshoot. GT1 is disabled unless
VCC is outside the UVLO region and VREF is on.
GT2 (Pin 10): A 2-A peak source and 3-A peak sink current totem pole MOSFET gate driver for the PWM stage,
identical to the driver at GT1.
IAC (Pin 18): This multiplier input senses the rectified ac line voltage. A resistor between IAC and the line
voltage converts the instantaneous line voltage waveform into a current input for the analog multiplier. The
recommended maximum IAC current is 500 µA.
ISENSE1 (Pin 16): This pin is the non-inverting input terminal of the current amplifier. Connect a resistor
between this pin and the grounded side of the PFC stage current sensing resistor. The resistor connected to
this pin should have a value that equals the value of the resistor that is connected between the MOUT pin and
the ungrounded side of the PFC current sense resistor.
ISENSE2 (Pin 8): A voltage across the PWM stage external current sense resistor generates the input signal
to this pin, with the peak limit threshold set to 1.3 V for peak current mode control. An internal 1.5-V level shift
between ISENSE2 and the input to the PWM comparator provides greater noise immunity. The oscillator ramp
can also be summed into this pin for slope compensation. Figure 36 shows the typical relationship of the
capacitance on the ISENSE2 pin and the minimum controllable limit of the pulse width on the gate2 output. If
the V
is at the voltage that corresponds to a minimum controllable duty cycle and then is reduced further
ERR
the pulse width collapses to near zero.
MOUT (Pin 17): The output of the multiplier and the input to the current amplifier in the PFC stage are internally
connected at this pin. Set the power range of the PFC stage with a resistor tied between the MOUT pin and the
non-grounded end of the PFC current sense resistor. Connect impedance between the MOUT pin and the
CAOUT pin to compensate the PFC current control loop. The multiplier output is a current and the current
amplifier input is high impedance. The multiplier output current is given by:
ǒV
* 1.0Ǔ I
VAOUT
IAC
I
+
MOUT
2
K ǒV
Ǔ
VFF
−1
where, K is the multiplier gain constant, in volts
.
I
is limited to two times I
for power limiting.
AC
MOUT
PKLMT (Pin 14): Program the peak current limit of the PFC stage using this pin. The threshold for peak limit
is 0 V. Use a resistor divider between VREF and the non-grounded side of the PFC current sense resistor in
order to shift the level of this signal to a voltage that corresponds to the desired overcurrent threshold voltage,
measured across the PFC current sense resistor.
PWRGND (Pin 11): Ground for the output drivers at GT1 and GT2. This ground should be tied to GND externally
via a single Kelvin connection.
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DETAILED PIN DESCRIPTIONS (CONTINUED)
RT (Pin 2): A resistor between RT and GND programs the oscillator frequency, measured at CT_BUFF. In all
options, the PWM stage operates at the frequency that is measured at CT_BUFF. In the UCC28521 the PFC
stage operates at the same frequency as the PWM stage. The voltage is dc (nominally 3 V); do not connect a
capacitor to this pin in an attempt to stabilize the voltage. Instead, connect the GND side of the
oscillator-programming resistor closer to the GND pin. The recommended range of resistors is 45 kΩ to 500
kΩ for a frequency range of 600 kHz to 65 kHz, respectively. Resistor R programs the oscillator frequency f ,
T
S
as measured at CT_BUFF, according to the following equation:
1
1
*7
ǒ
Ǔ
R +
* 2.0 10
T
*12
f
31 10
S
where, R is in Ω
T
S
f is in Hz
SS2 (pin 13): A capacitor between SS2 and GND programs the softstart duration of the PWM stage gate drive.
When the UVLO2 comparator enables the PWM stage, an internal 10.5-µA current source charges the external
capacitor at SS2 to 3 V to ramp the voltage at VERR during startup. This allows the GT2 duty cycle to increase
from 0% to the maximum clamped by the duty cycle comparator over a controlled time delay t given by:
SS
*6
t
10.5 10
SS
C
+
,
Farads
SS2
3
In the event of a disable command or a UVLO2 dropout, SS2 quickly discharges to ground to disable the PWM
stage gate drive.
VAOUT (Pin 1): This transconductance voltage amplifier output regulates the PFC stage output voltage and
operates between GND and 5.5 V maximum to prevent overshoot. Connect the voltage compensation
components between VAOUT and GND. When this output goes below 1 V, the multiplier output current goes
to zero. When this output falls below 0.33 V, the zero power detect comparator ensures the PFC stage gate drive
is turned off. In the linear range, this pin sources or sinks up to 30 µA. A slew rate enhancement feature enables
VAOUT to sink or source up to 3.3 mA, when operating outside the linear range.
VCC (Pin 9): Chip positive supply voltage that should be connected to a stable source of at least 20 mA between
12 V and 17 V for normal operation. Bypass VCC directly to GND with a 0.1 µF or larger ceramic capacitor to
absorb supply current spikes caused by the fast charging of the external MOSFET gate capacitances.
VERR (Pin 7): The voltage at this pin controls the GT2 duty cycle and is connected to the feedback error signal
from an external amplifier in the PWM stage. This pin is clamped to a maximum of 3 V and can demand 100%
duty cycle at GT2. The typical pull-up current flowing out of this pin is 10 µA.
VFF (Pin 19): The output current from this pin comes from an internal current mirror that divides the IAC input
current by 2. The input voltage feedforward signal for the multiplier is then generated across an external
single-pole R/C filter connected between VFF and GND. At low line, the VFF voltage should be set to 1.4 V.
VREF (Pin 20): This is the output of an accurate 7.5-V reference that powers most of the internal circuitry and
can deliver over 10 mA, with a typical load regulation of 5 mV ensured for an external load of up to 6 mA. The
internal reference is current limited to 25 mA, which protects the part if VREF is short-circuited to ground. VREF
should be bypassed directly to GND with a ceramic capacitor between 0.1 µF and 10 µF for stability. VREF is
disabled and remains at 0 V when VCC is below the 9.7-V UVLO threshold.
VSENSE (Pin 3): Inverting input to the PFC transconductance voltage amplifier, which serves as the PFC
feedback connection point. When VSENSE operates within +/− 0.35 V of its steady-state value, the current at
VAOUT is proportional to the difference between the VREF and VSENSE voltages by a factor of g . Outside
M
this range, the magnitude of the current of VAOUT is increased in order to enhance the slew rate for rapid voltage
control recovery in the PFC stage. Decisive activation and deactivation of the voltage control recovery is
internally implemented with about 120 mV of hysteresis at VSENSE. VSENSE is internally connected to the PFC
OVP, Enable and UVLO2 comparators as well.
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APPLICATION INFORMATION
D2
L1
D3
PRIMARY
SECONDARY
D4
+
R3
R4
R6
T1
C2
VOUT
D1
C1
R1
−
Q2
Q1
R2
R5
GND2
VAC
PGND
R11
R9
D5
C3
R10
R8
R7
U1
UCC28521/28
AGND
REF
11 PWRGND
12 GT1
GT2 10
VCC
ISENSE2
VERR
9
8
PGND
C5
R16 R17
C4
13 SS2
R22
D6
14 PKLMT
15 CAOUT
7
AGND
R12
R18
GND
6
GND2
C6
R14
16 ISENSE1 CT_BUFF
5
C7
C14
R25
R13
AGND PGND
17 MOUT
18 IAC
D_MAX
VSENSE
RT
4
R19
U2
R23
3
R20
C13
R25
19 VFF
2
20 VREF
VAOUT
1
C12
R15
C8
C10
REF
C9
C11
U3
TL431
R24
R21
AGND
GND2
Figure 1. Typical Application Circuit: Boost PFC and Flyback PWM Power System
The UCC28521 combination controller includes a power factor correction (PFC) controller that is synchronized
with a pulse width modulator (PWM) controller integrated into one chip. The PFC controller has all of the features
for an average current mode controlled PFC. The PWM controller has all of the features for an isolated peak
current program mode controlled converter. The two controllers are internally synchronized at a fixed frequency
with both the PFC controller and the PWM controller are trailing edge modulated (TEM).
11
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APPLICATION INFORMATION
design procedure
The following discussion steps through the typical design process of a PFC/PWM converter system. The design
process begins with the power stage elements, then the control elements for the PFC stage, then the control
elements for the PWM stage. Keep in mind that a general design process is often iterative. Iteration typically
begins after either simulating and/or testing the completed PFC/PWM system. This design procedure refers to
the typical application in Figure 1.
A design begins with a list of requirements for output voltage, output power and ac line voltage range. Other
details, such as efficiency and permissible current harmonics could be given at the onset, or developed
throughout the product design cycle. The need for power factor correction arises from either an agency
requirement, such as EN−61000, or if the available line power is nearly equal to the output power of the power
system. Hold-up time requirements are also necessary at the early stages of design. Typically, the hold-up time,
t
, is at least the period of 1.5 line cycles.
HU
The general structure of the PFC/PWM stage power system is two switched-mode converters connected in
cascade. Each stage has an associated efficiency and each stage has its own set of fault limiting controls that
must be properly set in order to achieve the desired line harmonic and load regulation performance,
simultaneously. The PFC stage must always be designed to supply sufficient average power to the PWM stage.
The cycle-by-cycle current limit of the PFC stage should be programmed to activate at a slightly larger power
level at low ac line voltage than the average power clamp in order to allow for PFC current sense tolerances.
This will allow power factor correction for the full range of maximum rated load. If the instantaneous load nearly
equals the average load, then the fault clamps for the PWM stage can be programmed to limit power at a level
that is slightly less than or equal to the average power clamp of the PFC stage. The margin for the clamping
action should allow for measurement tolerances and efficiency. Conversely, if the instantaneous load has high
peaks that are much shorter than the hold-up time, the current limit and duty ratio limits of the PWM stage can
clamp at a higher level than the average power clamp in the PFC stage. In order to simplify the design procedure,
the average and the peak loads of the PWM stage are assumed to be equal. Thus, all of the current limits and
duty cycle limits are programmed to clamp power at a slightly lower level (10%) than the average power clamp
on the PFC stage.
developing the internal parameters
Select the energy storage voltage V . Since the PFC stage is a boost converter, the voltage across C1 must
C1
be larger than the peak ac line voltage by enough to permit controllability in the event of load transients. Typically,
this will be around 5% which is about 400 V for a universal ac line application of 85 V to 265 V
.
AC
AC
Once the energy storage voltage, V , is determined, the range of the PFC stage duty ratio, D , is set. For CCM
C1
1
operation of the PFC stage, the minimum PFC duty ratio is given by:
Ǹ
2 VAC
MIN
D
+ 1 *
1(min)
V
C1
(1)
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APPLICATION INFORMATION
Select an approximate switching frequency for the PFC stage. A good starting frequency for a MOSFET based
PFC stage is in the range of 100 kHz to 200 kHz, depending on maximum line voltage and maximum line current.
Adjustments in switching frequency may result from meeting switching loss requirements in Q1 and D3, or in
order to optimize the design of L1.
Select an appropriate topology for the PWM stage using the information about the power requirements and the
magnitude of V . For simplicity, the typical application in Figure 1 shows a flyback converter in the PWM stage.
C1
In most cases, the PWM stage topology must have transformer isolation and the topology must require only one
pulse-width signal. Topologies that have these features include:
• single-transistor forward converter
• single-transistor flyback converter
• two-transistor forward converter
• two-transistor flyback converter
Estimate the nominal and the maximum duty ratios of the PWM stage (D
, D
and the associated peak
2(nom) 2(max)
Q2 drain current, i
), based on the topology, PWM hysteresis option and output voltage requirements of
Q2(peak)
the PWM stage. Also estimate whether or not it is appropriate to operate the PWM stage at the same switching
frequency as the PFC stage or if the PWM stage can operate at twice the switching frequency of the PFC stage.
Base the estimation for the switching frequency of the PWM stage on the maximum voltages and currents of
the power MOSFETs and power diodes. Program the oscillator frequency of the PWM stage with the value of
R20.
1
1
*7
R20 +
ǒ
* 2.0 10
Ǔ
,
W
*12
f
31 10
S(pwm)
(2)
Most applications require that the PWM stage regulates at the minimum energy storage capacitance voltage.
Maximum duty ratio D
to estimate the peak current stresses for transformer T1 and any other inductive element in the PWM stage.
and i
should be calculated for the minimum energy storage voltage in order
2(max)
Q2(peak)
At this point, enough information is available to estimate which member of the UCC28510 family should be
selected.
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APPLICATION INFORMATION
power stage elements
The power stage elements include the following elements: C1−3, D1−5, L1, R2, R5, Q1, Q2, T1. Details
concerning the PWM stage elements C2, D4, D5, Q2 and T1 will not be discussed in detail here, due to their
dependence on the choice PWM stage topology. The PWM stage is an isolated dc-to-dc topology with the same
stresses and loss mechanisms that are typical for the selected topology. An estimation of the average steady
state duty ratio of the PWM stage and the Q2 switch current will be needed for stress estimations in the PFC
stage. Also, the natural step response of the PWM stage is required to estimate the soft start capacitor, C5, and
the bias supply capacitor, C3.
The selection process of the PFC stage elements C1, C3, D1−3 and Q1 are discussed in detail here. In general,
the selection process for the PFC stage elements is the same as for a typical fixed switching frequency PFC
design, except for capacitor C1 due to PFC/PWM stage synchronization.
Diode bridge D1 is selected to withstand the rms line current and the peak ac line voltage. Diode D2 allows
capacitor C1 to charge during initial power up without saturating L1 and it is selected to withstand the peak inrush
current and peak of the maximum ac line voltage. Additional inrush current limiting circuitry in series with the
ac line could be required, depending on agencies or situations.
The PFC stage inductor, L1, is selected to have a maximum current ripple at the minimum ac line voltage.
Typically a ripple factor, k , is chosen to be about 0.2. If the line current has excessive crossover distortion,
RF
a larger ripple factor (perhaps 0.3) will reduce the distortion but the line current will have more switching ripple.
Initially, the inductance can be estimated by approximating the input power equal to the output power.
2
V
D
T
AC(min)
1(min)
S(pfc)
L1 +
k
P
RF
IN
(3)
Di
L1(p−p)
1
where, k
+
and T
is
S(pfc)
RF
i
switching frequency of the PFC
L1(max)
Inductor L1 must be designed to withstand the maximum ac rms line current without saturation at the peak ac
line current.
Select power MOSFET Q1 and diode D3 with the same criteria that is normally used for fixed switching
frequency PFC design. They must have sufficient voltage rating to withstand the energy storage voltage, V
and they must have sufficient current ratings. Gate drive resistor R9 is necessary to limit the source and sink
C1
current from the GT1 pin. Some circumstances require additional gate drive components for improved
[10]
protection and performance.
of Q2 for the same reason.
A similar gate drive resistor, R10, is required between the GT2 pin and the gate
14
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
APPLICATION INFORMATION
The current sense resistor for the PFC stage, R2, is selected to operate over a 1-V dynamic range (V
).
DYNAMIC
The sense resistor must also have a large enough power rating to permit safe operation with the maximum RMS
line current.
V
DYNAMIC
) 0.5 Di
R2 +
i
L1(max)
L1(p−p)
IN
(4)
Ǹ
2 P
where, i
+
L1(max)
V
AC(min)
The PFC I
comparator threshold is at the ground reference for the controller device. So, the PFC current
LIMIT
sense voltage, measured at PKLMT must be biased with a positive voltage to cross 0.0 V when the
instantaneous PFC current is at its maximum. The bias voltage is established with R14 and R7, as shown in
equation 7, and resistor R14 is arbitrarily chosen around 10 kΩ.
R7
R14
1
+
V
REF
* 1
i
R2
L1(max)
(5)
The capacitance value of the energy storage capacitor, C1 is selected to meet hold-up time requirements (t
by the equation:
)
HU
2P
t
OUT
HU
C1 +
2
2
ǒ
Ǔ
V
* V * 0.7V
C1
C1
C1
(6)
Capacitor C1 must be rated for the selected energy storage voltage and it must be able to withstand the rms
ripple current, I
, that is produced by the combined action of the PFC stage and the PWM stage. The
C1(rms)
average Q2 drain current during the interval that GT2 activates MOSFET Q2 is used to find I
estimate can be made using the inequality in equation 9, then consult Figure 2 for better accuracy.
. An initial
C1(rms)
16 V
BOOST
I
+ P
* 1
Ǹ
C1(rms)
OUT
3 * p V
AC(max)
(7)
The ratio of I
to I can be found by using the appropriate graph, Figure 3 for the 1X:1X oscillator option
or Figure 4 for the 1X:2X oscillator option. To use the graphs, locate the ratio of VAC to V along the horizontal
axis then, draw a vertical line to the intersection of the curve for the duty ratio of the PWM stage. Draw a
C1(rms)
Q2
C1
horizontal line from the intersection to the vertical axis and read the ratio of I
to I
.
C1(rms)
Q2
15
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
APPLICATION INFORMATION
The current sense resistor for the PWM stage, R5, is selected so that at maximum current, its voltage is the
ǒPWM stage I
threshold voltage of the peak current Ǔcomparator (nominally 1.3 V).
V
TH
LIMIT
R5 +
i
Q2(peak)
(8)
In many cases, an input line filter will be necessary in order to meet the requirements of an agency or application.
The input line filter design has been omitted from this procedure due to the vast array of requirements and
circumstances. We urge you to refer to Reference [11] for details.
PFC stage control
The PFC stage is designed in a three-step process. First, set the dynamic range of the multiplier, second,
stabilize the average current control loop and third, stabilize the voltage loop that controls the energy storage
capacitor voltage. Use as much of the dynamic range of the multiplier as possible. The current control loop must
have wide bandwidth in order to follow the instantaneous rectified line voltage. The voltage loop must be slower
than twice the ac line frequency so that it will not compromise the power factor.
multiplier
The dynamic range of the multiplier is a function of the currents and/or voltages of the IAC, VAOUT and VFF
pins. Coordinate the selection process to use the full range of the multiplier and obtain the desired power limiting
features. Select the components R1 and R15 to use the i (t) range and the V
range under the condition
IAC
VFF
that the maximum of the V
range, described in equation 11. The selection process is similar to the
VAOUT
[12]
selection process for UC3854, except for the VFF voltage and MOUT current limitations.
In this product
series, the divide-by-square function is internally implemented so that it divides by the greater of 1.4 V or V
.
VFF
If the 1.4-V level controls the divider, power factor correction may still occur if the VAOUT level is within the
functioning range of the multiplier. Power factor correction occurs during that condition because the multiplier
section functions as a two-input multiplier, rather than a three-input multiplier. Notice that the voltage at the VFF
pin will be proportional to the average of the IAC current. Typically, V
=1.4 V at low ac line voltage is set as
VFF
the design boundary; the upper boundary of V
range varies by less than 4.3:1.
will remain within the range if the functional ac line voltage
VFF
( )
0 v i
t v 500 mA,
IAC
(9)
( )
0 v V
t v 5 V,
VAOUT
1.4 V v V
v V
* 1.4 V
VFF
VREF
16
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
APPLICATION INFORMATION
The selection process begins with the selection of R1 so that the peak I current at high ac line is about 500 µA,
AC
see Table 2. Second, select R15 for the minimum VFF voltage, also shown in Table 2. Third, select C8, in Table
2, to average the VFF voltage with sufficiently low ripple to meet a third harmonic distortion budget. For a system
with a 3% THD target, it is typical to allow the feedforward circuit to contribute 1.5% third harmonic distortion
to the input waveform [4]. An attenuation factor of 0.022 will meet the criteria. Finally, select the MOUT resistor
in Table 2, R12, so that the voltage across R12 equals the voltage across sense resistor R2 under the condition
of maximum power, minimum ac line voltage (V
), and VAOUT at its maximum level of 5 V. Experimentally,
VFF,MIN
the multiplier output resistor, R12, may need to be increased slightly if the energy storage capacitor voltage sags
under maximum load. This would be due to tolerances in the components and the multiplier. In order to minimize
current amplifier offsets, set the value of the resistor on the ISENSE1 pin, R8, equal to the value of R12 as shown
in in Table 2.
Table 1.
REFERENCE
DESIGNATOR
EQUATION
NOTES
Ǹ
I
2 V
AC(max)
set i
+ 500 mA
R1
IAC(peak)
IAC(peak)
V
VFF(avgmin)
set V
+ 1.4 V
R15
C8
2 R1
VFF(avgmin)
V
AC(min) 0.9
1
A
A
+ 0.022 for 3% THD
FF(2)
2 p f
R15
AC
FF(2)
2
k = 1
V
V
(
)(
)
P
R1 R2 k
VFF(min)
IN
= 1.4V
R12
R8
VFF(min)
ǒ Ǔ
V
* 1
V
V
= 5.0V
VAOUT(max)
VAOUT(max)
R12
AC(min)
Always change R8 if R12 is changed
PFC current loop control
This controller uses average current loop control for the PFC stage. The current control loop must typically be
fast enough to track the rectified sinusoidal ac line voltage. There are many ways to design a controller that will
[5]
stabilize the PFC current loop. The method that is described here achieves good results for most applications.
This method assumes that both the natural frequency of the system and the zero of the linearized boost PFC
are much lower than both the switching frequency and the desired crossover frequency, f
, as described
CO(pfc)
in equation 12. The left side of the inequality in equation 12 will usually be true since the capacitance of C1 is
quite large.
1 * D
2P
PWM(min)
IN
and
tt 2 p f
tt 2 p f
CO(pfc)
S(pfc)
2
Ǹ
L1 C1
C1 V
C1
(10)
17
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
APPLICATION INFORMATION
The left side of the inequality should be at least a factor of 10 lower than the middle term; the right side of the
inequality should be at least five times larger than the middle term. For the purposes of 50 Hz to 60 Hz power
factor correction, good results can be achieved with the crossover frequency set to about 10 kHz. A lower
crossover might be necessary if the switching frequency of the PWM stage is below 100 kHz, or if the
compensator gain at the crossover frequency is large (over ~40 dB).
Upon selecting the crossover frequency, select R13 to set the gain at the crossover frequency, then select C6
to place a zero at the crossover frequency and select C7 to provide a pole at half of the switching frequency.
The equations are in Table 3.
Table 2.
REFERENCE
DESIGNATOR
EQUATION
NOTES
2 p f
L1 V
CO(pfc)
CT_BUFF(p−p)
V
+ 4 V
R13
R12
CT_BUFF(p−p)
V
C1
1
C6
C7
R13 2 p f
CO(pfc)
1
p f
S(pfc)
PFC voltage loop
The voltage loop must crossover at a lower frequency than twice the ac line frequency so that voltage
corrections will not interfere with power factor correction. Second harmonic ripple from the sensed V voltage
C1
directly results in third harmonic distortion on the ac line, similar to ripple on the VFF voltage.
PWM stage control
The control elements of the PWM stage are the same as a typical isolated current program mode converter.
The secondary elements include C12 to C14, D6, R22 to R25, U2 and U3, which perform the error amplifier,
compensation and isolation functions. On the primary side, VERR is connected to the node between the
opto-isolator output, U2, and a pull-up resistor, R17. Resistor R17 represents the gain in the conversion from
the output current of opto-isolator U2 and the VERR input.
Slope compensation is programmed using resistors R18 and R11, which form a summing node at ISENSE2.
The voltage at CT_BUFF is a saw-tooth waveform that swings between 1 V and 5 V.
Many applications require a duty ratio limit for the PWM stage in order to prevent transformer saturation.
Program the maximum duty ratio using the following ratio of resistors R16 to R19.
V
R16
R19
VREF
+
* 1
1 ) 4 D
PWM(max)
(11)
Soft-start
The soft-start capacitor, C5, which is connected to SS2, controls the soft-start ramp of the PWM stage. The
soft-start ramp begins when the VSENSE voltage exceeds 6.75 V. In order to avoid loop saturation, the soft-start
ramp rate must be less than or equal to the open loop response of the PWM stage converter.
18
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
REFERENCE DESIGN
Universal line input 100-W PFC output with 12 V, 8-W bias rail supply design is discussed in UCC28517EVM,
TI literature number SLUU117. UCC28517 is a closely related combo device to the UCC28521. The UCC28517
has 1:2 PFC:PWM frequency option. The schematic is shown in Figures 2, 3, 4. Please refer to the SLUU117
document on http://www.ti.com for further details.
+
Figure 2. Section A
+
Figure 3. Section B
19
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
REFERENCE DESIGN
Note: D10 and D9 are Schottky diodes
from Vishay, part no. BYS10−25
Figure 4. Section C
20
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
TYPICAL CHARACTERISTICS
STARTUP CURRENT
vs
SUPPLY CURRENT
vs
TEMPERATURE
TEMPERATURE
5.0
4.8
140
130
120
4.6
4.4
4.2
4.0
110
100
90
3.8
3.6
UCC 28521
3.4
3.2
3.0
80
−50
−25
0
25
50
75
100
125
−50
−25
0
25
50
75
100
125
Temperature − °C
Temperature − °C
Figure 5
Figure 6
SUPPLY CURRENT
vs
REFERENCE VOLTAGE
vs
SUPPLY VOLTAGE
TEMPERATURE
4.5
7.60
7.58
7.56
7.54
7.52
7.50
7.48
7.46
7.44
7.42
10.2 V UVLO
Turn-On Threshold
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
7.40
4.0
6.0
8.0
10.0 12.0 14.0
16.0 18.0
−50
−25
0
25
50
75
100
125
V
CC
− Supply Voltage − V
Temperature − °C
Figure 7
Figure 8
21
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
TYPICAL CHARACTERISTICS
VREF
vs
LOAD CURRENT
VREF CURRENT LIMIT
8.0
7.500
7.495
7.490
7.485
7.480
7.475
7.470
7.465
7.460
V
CC
= 10 V
7.0
6.0
5.0
4.0
3.0
2.0
1.0
0.0
V
= 12 V
CC
V
CC
= 15 V
0
10
20
30
40
0.0
5.0
10.0
15.0
20.0
V
REF
− External Load Current − mA
I
− External Load Current − mA
REF
Figure 9
Figure 10
PFC UVLO THRESHOLDS
vs
TEMPERATURE (UCC28521)
PWM UVLO2 THRESHOLDS
vs
TEMPERATURE (UCC28521)
12
8
UVLO On
UVLO2 On
7
6
10
8
UVLO Off
UVLO2 Off
5
4
3
6
4
2
1
0
UVLO2 Hysteresis
2
0
UVLO Hysteresis
−50
−25
0
25
50
75
100
125
−50
−25
0
25
50
75
100
125
Temperature − °C
Temperature − °C
Figure 11
Figure 12
22
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
TYPICAL CHARACTERISTICS
OSCILLATOR FREQUENCY
OSCILLATOR FREQUENCY
vs
vs
RT
RT OVER VCC (11 V TO 15 V) (−40°C TO 105°C)
800
700
600
500
1000
100
0
400
300
200
100
0
10
100
1000
10
100
1000
RT − Timing Resistor − kΩ
RT − Timing Resistor − kΩ
Figure 13
Figure 14
GT1 MAXIMUM DUTY CYCLE
vs
GT1, GT2 PULL-UP, PULL-DOWN RESISTANCE
vs
PFC SWITCHING FREQUENCY (C
= 0.85 V)
TEMPERATURE
AOUT
8
7
6
5
100
99
98
Pull-up
4
3
97
96
2
1
0
Pull-down
95
94
100
300
−50
−25
0
25
50
75
100
125
200
400
0
500
Temperature − °C
GT1 Switching Frequency − kHz
Figure 15
Figure 16
23
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
TYPICAL CHARACTERISTICS
GT1 RISE/FALL TIME
vs
GT1, GT2 RISE AND FALL TIMES
vs
C
AND R
(V = 12 V)
TEMPERATURE
LOAD
SERIES CC
18
16
14
12
10
45
40
35
30
25
20
t : R
= 2 Ω
R
SERIES
t
R
t : R
F SERIES
= 2 Ω
t
F
t : R
R
= 10 Ω
SERIES
8
6
4
2
t : R
F
= 10 Ω
SERIES
12
10
5
0
0
−50
−25
0
25
50
75
100
125
0
2
4
6
C
− nF
Temperature − °C
LOAD
Figure 17
Figure 18
GT2 MAXIMUM DUTY CYCLE
MULTIPLIER OUTPUT CURRENT
vs
vs
D_MAX VOLTAGE
VOLTAGE ERROR AMPLIFIER OUTPUT
100
350
100 kHz
200 kHz
90
80
300
350
I
= 150 µA
AC
V
FF
= 1.4 V
70
60
50
500 kHz
I
= 300 µA
AC
200
150
100
V
FF
= 2.8 V
800 kHz
40
30
20
10
50
0
I
= 500 µA
AC
V
FF
= 4.7 V
0
1.0 1.5
2
2.5 3.0 3.5 4.0 4.5
D_MAX Voltage − V
Figure 19
5.0 5.5
1.0
2.0
4.0
6.0
V
− Voltage Error Amplifier Output − V
AOUT
Figure 20
24
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
TYPICAL CHARACTERISTICS
MULTIPLIER
CONSTANT POWER PERFORMANCE
MULTIPLIER GAIN
vs
VOLTAGE ERROR AMPLIFIER OUTPUT
500
2.0
1.8
1.6
1.4
1.2
V
= 5 V
AOUT
450
400
V
= 4 V
= 3 V
= 2 V
350
AOUT
300
350
V
AOUT
200
150
100
50
I
= 150 µA, V = 1.4 V
FF
AC
V
AOUT
I
= 300 µA, V = 2.8 V
FF
AC
1.0
0.8
I
= 500 µA, V = 4.7 V
FF
AC
0
0
1
2
3
4
5
1.0 1.5 2.0 2.5 3.0 3.5 4.0
4.5 5.0 5.5
V
− Voltage Error Amplifier Output − V
V
FF
− Feedforward Voltage − V
AOUT
Figure 21
Figure 22
CURRENT AMPLIFIER OPEN LOOP
GAIN AND PHASE
VOLTAGE AMPLIFIER TRANSCONDUCTANCE
vs
TEMPERATURE
120
150
120
90
100
80
Phase
60
Gain
60
40
30
0
−30
−60
−90
20
0
8
10
4
10
6
10
2
10
−50
−25
0
25
50
75
100
125
1
Temperature − °C
Frequency − Hz
Figure 23
Figure 24
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
TYPICAL CHARACTERISTICS
VOLTAGE AMPLIFIER
OUTPUT CURRENT CAPABILITY
VOLTAGE AMPLIFIER OUTPUT CURRENT IN
LINEAR REGION OF OPERATION
4
3
2
1
40
30
20
10
0
0
−1
−10
−2
−3
−4
−20
−30
−40
7.0
7.2
7.4
7.6
7.8
8.0
7.0
7.2
7.4
7.6
7.8
8.0
V
− Voltage Normalized to V
− V
V
− Voltage Normalized to V
− V
SENSE
REF
SENSE
REF
Figure 25
Figure 26
VOLTAGE AMPLIFIER V
BIAS CURRENT
SENSE
VOLTAGE AMPLIFIER
OPEN LOOP GAIN AND PHASE
vs
TEMPERATURE
300
250
180
V
Load = 15 pF
AOUT
Phase
150
120
200
150
90
60
Gain
100
30
0
50
0
−30
5
10
7
10
10
3
10
−50
−25
0
25
50
75
100
125
Temperature − °C
Frequency − Hz
Figure 27
Figure 28
26
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
TYPICAL CHARACTERISTICS
VOLTAGE AMPLIFIER SLEW CURRENTS
SOFTSTART CURRENTS
vs
vs
TEMPERATURE
TEMPERATURE
15
5
4
I
(mA)
DISCHG
10
5
3
2
I
SINK
1
0
0
−1
−2
−5
I
(µA)
CHARGE
−3
−4
−5
I
SOURCE
−10
−15
−50
−25
0
25
50
75
100
125
−50
−25
0
25
50
75
100
125
Temperature − °C
Temperature − °C
Figure 29
Figure 30
TYPICAL MINIMUM ON TIME
vs
CAPACITANCE
60
Absolute Maximum Limit
of Capacitance Allowed
50
40
30
20
10
0
0
100
200
300
400
500
Capacitance − pF
Figure 31
27
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
REFERENCES
1. Evaluation Module and associated User’s Guide, UCC28521EVM, Texas Instruments Literature Number
SLUU218
2. Datasheet, UCC38500/1/2/3 BiCMOS PFC/PWM Combination Controller, Texas Instruments Literature
Number SLUS419C
3. Power Supply Seminar SEM−600, High Power Factor Preregulator for Off-line Power Supplies, L.H. Dixon,
Texas Instruments Literature Number SLUP087
4. Power Supply Seminar SEM−700, Optimizing the Design of a High Power Factor Switching Preregulator,
L.H. Dixon, Texas Instruments Literature Number SLUP093
5. Power Supply Seminar SEM−1500 Topic 2, Designing High-Power Factor Off−Line Power Supplies, by
James P. Noon
6. Application Note, UC3854 Controlled Power Factor Correction Circuit Design ,Texas Instruments Literature
Number SLUA144
7. Design Note, Optimizing Performance in UC3854 Power Factor Correction, Texas Instruments Literature
Number SLUA172
8. Design Note, UC3854A and UC3854B Advanced Power Factor Correction Control ICs, Texas Instruments
Literature Number SLUA177
9. Design Note, UC3854A/B and UC3855A/B Provide Power Limiting with Sinusoidal Input Current for PFC
Front Ends, Texas Instruments Literature Number SLUA196
10. Laszlo Balogh, A Design and Application Guide for High Speed Power MOSFET Gate Drive Circuits, 2001
Power Supply Design Seminar Manual SEM1400, 2001
11. Bob Mammano and Bruce Carsten, Understanding and Optimizing Electromagnetic Compatibility in
Switchmode Power Supplies, 2002 Power Supply Design Seminar Manual SEM1500, 2002
RELATED PRODUCTS
PART NUMBER
DESCRIPTION
COMMENTS
Advanced BiCMOS PFC/PWM combination
controllers
1:1 or 1:2 frequency option; leading edge, trailing edge modulation,
programmable PWM Max dc
UCC28510-17
UCC38500/1/2/3 BiCMOS PFC/PWM combination controller
1:1 leading edge, trailing edge modulation, 50% PWM Max dc
High PF, UC3854 compatible, leading edge trailing edge modulation
UCC3817/18
UCC3819
BiCMOS power factor preregulator
Programmable output power factor preregulator Tracking boost topology for dynamic output voltage adjustments
UC3854
High Power Factor Preregulator
High PF, industry standard PFC controller; 35 V max
CC
UC3854A/B
UC3855A/B
Enhanced high power factor preregulator
High performance power factor preregulator
Improved high PF, industry standard PFC controller; 22 V
max
CC
ZVT output for lower EMI emission & higher efficiencies
8-Pin package; simplified architecture to minimized external compo-
nents
UC3853
High power factor preregulator
UCC38050
UC3852
Transition mode PFC controller
High power factor preregulator
Constant on-time transition mode PFC controller
Constant off-time transition mode PFC controller; 30 V
max
CC
REVISION HISTORY
REVISION
DATE
CHANGE
SLUS608B
6/27/05
Corrected CAOUT pin description.
28
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SLUS608B − JANUARY 2005 REVISED JUNE 2005
MECHANICAL DATA
DW (R-PDSO-G**)
PLASTIC SMALL-OUTLINE PACKAGE
16 PINS SHOWN
0.020 (0,51)
0.050 (1,27)
16
0.010 (0,25)
0.014 (0,35)
9
0.419 (10,65)
0.400 (10,15)
0.010 (0,25) NOM
0.299 (7,59)
0.291 (7,39)
Gage Plane
0.010 (0,25)
1
8
0°− 8°
0.050 (1,27)
0.016 (0,40)
A
Seating Plane
0.012 (0,30)
0.004 (0,10)
0.004 (0,10)
0.104 (2,65) MAX
PINS **
16
18
20
24
0.610
28
DIM
0.410
0.462
0.510
0.710
(18,03)
A MAX
(10,41) (11,73) (12,95) (15,49)
0.400
0.453
0.500
0.600
0.700
(17,78)
A MIN
(10,16) (11,51) (12,70) (15,24)
4040000/E 08/01
NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. Body dimensions do not include mold flash or protrusion not to exceed 0.006 (0,15).
D. Falls within JEDEC MS-013
29
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PACKAGE OPTION ADDENDUM
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27-Jun-2005
PACKAGING INFORMATION
Orderable Device
Status (1)
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
SOIC
SOIC
SOIC
SOIC
Drawing
UCC28521DW
UCC28521DWR
UCC28528DW
UCC28528DWR
ACTIVE
ACTIVE
ACTIVE
ACTIVE
DW
20
20
20
20
25
2000
25
TBD
TBD
TBD
TBD
CU NIPDAU Level-1-220C-UNLIM
CU NIPDAU Level-1-220C-UNLIM
CU NIPDAU Level-1-220C-UNLIM
CU NIPDAU Level-1-220C-UNLIM
DW
DW
DW
2000
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan
-
The planned eco-friendly classification: Pb-Free (RoHS) or Green (RoHS
&
no Sb/Br)
-
please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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Addendum-Page 1
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