OPA861IDBVRG4 [TI]
Wide Bandwidth Operational Transconductance Amplifier 6-SOT-23 -40 to 85;
| 型号： | OPA861IDBVRG4 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | Wide Bandwidth Operational Transconductance Amplifier 6-SOT-23 -40 to 85 放大器 光电二极管 |
| 文件： | 总32页 (文件大小：1104K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
OPA861
www.ti.com
SBOS338G –AUGUST 2005–REVISED MAY 2013
Wide Bandwidth
Operational Transconductance
Amplifier (OTA)
Check for Samples: OPA861
The OTA or voltage-controlled current source can be
viewed as an ideal transistor. Like a transistor, it has
three terminals—a high impedance input (base), a
low-impedance input/output (emitter), and the current
output (collector). The OPA861, however, is self-
biased and bipolar. The output collector current is
zero for a zero base-emitter voltage. AC inputs
centered about zero produce an output current, which
is bipolar and centered about zero. The
transconductance of the OPA861 can be adjusted
with an external resistor, allowing bandwidth,
quiescent current, and gain trade-offs to be
optimized.
1
FEATURES
•
•
•
•
•
Wide Bandwidth (80MHz, Open-Loop, G = +5)
High Slew Rate (900V/µs)
High Transconductance (95mA/V)
External IQ-Control
Low Quiescent Current (5.4mA)
APPLICATIONS
•
•
•
•
•
•
•
Video/Broadcast Equipment
Communications Equipment
High-Speed Data Acquisition
Wideband LED Drivers
Control Loop Amplifiers
Wideband Active Filters
Line Drivers
Used as
a basic building block, the OPA861
simplifies the design of AGC amplifiers, LED driver
circuits for fiber optic transmission, integrators for fast
pulses, fast control loop amplifiers and control
amplifiers for capacitive sensors, and active filters.
The OPA861 is available in SO-8 and SOT23-6
surface-mount packages.
DESCRIPTION
The OPA861 is a versatile monolithic component
designed for wide-bandwidth systems, including high
performance video, RF and IF circuitry. The OPA861
is a wideband, bipolar operational transconductance
amplifier (OTA).
0
−
−
−
−
−
−
−
−
10
20
30
40
50
60
70
80
R
10MHz
Low−Pass Filter
C1
R
20kHz
Low−Pass Filter
VIN
VOUT
C2
1k
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Low−Pass Negative Impedance Converter (NIC) Filter
Frequency Response of 20kHz and 10MHz
Low−Pass NIC Filters
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2005–2013, Texas Instruments Incorporated
OPA861
SBOS338G –AUGUST 2005–REVISED MAY 2013
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION(1)
SPECIFIED
PACKAGE
DESIGNATOR
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT MEDIA,
QUANTITY
PRODUCT
PACKAGE
OPA861ID
OPA861IDR
Rails, 75
OPA861
SO-8
D
–45°C to +85°C
–45°C to +85°C
OPA861
N5R
Tape and Reel, 2500
Tape and Reel, 250
Tape and Reel, 3000
OPA861IDBVT
OPA861IDBVR
OPA861
SOT23-6
DBV
(1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
ABSOLUTE MAXIMUM RATINGS(1)
Power Supply
±6.5VDC
See Thermal Information
±1.2V
Internal Power Dissipation
Differential Input Voltage
Input Common-Mode Voltage Range
Storage Temperature Range: D
Lead Temperature (soldering, 10s)
Junction Temperature (TJ)
ESD Rating:
±VS
–65°C to +125°C
+260°C
+150°C
Human Body Model (HBM)(2)
1500V
1000V
Charge Device Model (CDM)
(1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress ratings only, and functional operations of the device at these and any other conditions
beyond those specified is not supported.
(2) Pin 2 for the SO-8 package > 500V HBM. Pin 4 for the SOT23-6 package > 500V HBM.
Figure 1. PIN CONFIGURATION
Top View
1
2
3
4
8
7
6
5
IQ Adjust
C
1
2
3
6
5
4
IQ Adjust
+VS
C
E
B
V+ = +5V
NC
−
VS
B
−
V
−
5V
=
NC
E
SOT23−6
SO−8
2
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Product Folder Links: OPA861
OPA861
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SBOS338G –AUGUST 2005–REVISED MAY 2013
ELECTRICAL CHARACTERISTICS: VS = ±5V
RL = 500Ω and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
MIN/MAX OVER TEMPERATURE
TYP
0°C to
70°C(3)
–40°C to
+85°C(3)
MIN/
MAX
TEST
PARAMETER
CONDITIONS
+25°C
+25°C(2)
UNITS
LEVEL(1)
OTA—Open-Loop (see Figure 33)
AC PERFORMANCE
G = +5, VO = 200mVPP
,
Bandwidth
80
77
75
74
MHz
min
B
RL = 500Ω
G = +5, VO = 1VPP
G = +5, VO = 5VPP
G = +5, VO = 5V Step
VO = 1V Step
80
80
MHz
MHz
V/µs
ns
typ
typ
min
typ
C
C
B
C
Slew Rate
900
4.4
860
850
840
Rise Time and Fall Time
Harmonic Distortion
G = +5, VO = 2VPP, 5MHz
RL = 500Ω
2nd-Harmonic
–68
–57
2.4
1.7
5.2
–55
–52
3.0
–54
–51
3.3
–53
–49
3.4
dB
max
max
max
max
max
B
B
B
B
B
3rd-Harmonic
RL = 500Ω
dB
Base Input Voltage Noise
Base Input Current Noise
Emitter Input Current Noise
OTA DC PERFORMANCE(4) (see Figure 33)
f > 100kHz
nV/√Hz
pA/√Hz
pA/√Hz
f > 100kHz
2.4
2.45
16.6
2.5
f > 100kHz
15.3
17.5
Minimum OTA Transconductance (gm
)
VO = ±10mV, RC = 50Ω, RE = 0Ω
VO = ±10mV, RC = 50Ω, RE = 0Ω
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, VC = 0V
95
95
±3
80
77
155
±15
±67
±6
75
mA/V
mA/V
mV
min
max
max
max
max
max
max
max
max
max
A
A
A
B
A
B
A
B
A
B
Maximum OTA Transconductance (gm
)
150
±12
160
B-Input Offset Voltage
±20
Average B-Input Offset Voltage Drift
B-Input Bias Current
±120
±6.6
±25
μV/°C
μA
±1
±30
±5
±5
Average B-Input Bias Current Drift
E-Input Bias Current
±20
±125
±500
±30
±250
nA/°C
μA
±100
±18
±140
±600
±38
Average E-Input Bias Current Drift
C-Output Bias Current
VB = 0V, VC = 0V
nA/°C
μA
VB = 0V, VC = 0V
Average C-Output Bias Current Drift
OTA INPUT (see Figure 33)
B-Input Voltage Range
VB = 0V, VC = 0V
±300
nA/°C
±4.2
455 || 2.1
10.5
±3.7
±3.6
±3.6
V
kΩ || pF
Ω
min
typ
B
C
B
B
B-Input Impedance
Min E-Input Resistance
12.5
6.7
13.0
6.5
13.3
6.3
max
min
Max E-Input Resistance
10.5
Ω
OTA OUTPUT
E-Output Voltage Compliance
E-Output Current, Sinking/Sourcing
C-Output Voltage Compliance
C-Output Current, Sinking/Sourcing
C-Output Impedance
IE = ±1mA
VE = 0
±4.2
±15
±3.7
±10
±4.0
±10
±3.6
±9
±3.6
±9
V
mA
min
min
min
min
typ
A
A
A
A
C
IC = ±1mA
VC = 0
±4.7
±15
±3.9
±9
±3.9
±9
V
mA
54 || 2
kΩ || pF
(1) Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
(2) Junction temperature = ambient for +25°C specifications.
(3) Junction temperature = ambient at low temperature limit; junction temperature = ambient + 7°C at high temperature limit for over
temperature specifications.
(4) Current is considered positive out of node.
Copyright © 2005–2013, Texas Instruments Incorporated
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SBOS338G –AUGUST 2005–REVISED MAY 2013
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ELECTRICAL CHARACTERISTICS: VS = ±5V (continued)
RL = 500Ω and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
MIN/MAX OVER TEMPERATURE
TYP
0°C to
70°C(3)
–40°C to
MIN/
MAX
TEST
PARAMETER
CONDITIONS
+25°C
+25°C(2)
+85°C(3)
UNITS
LEVEL(1)
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage
Minimum Operating Voltage
Maximum Quiescent Current
Minimum Quiescent Current
OTA Power-Supply Rejection Ratio (+PSRR)
THERMAL CHARACTERISTICS
Specification: ID, IDBV
±5
V
V
typ
max
min
max
min
max
C
A
B
A
A
A
±6.3
±2.0
5.9
±6.3
±2.0
7.0
±6.3
±2.0
7.4
V
RADJ = 250Ω
RADJ = 250Ω
ΔIC/ΔVS
5.4
5.4
±20
mA
mA
µA/V
4.9
4.3
3.4
±50
±60
±65
–40 to +85
°C
typ
C
Thermal Resistance θ JA
D
SO-8
Junction-to-Ambient
Junction-to-Ambient
+125
+150
°C/W
°C/W
typ
typ
C
C
DBV SOT23-6
4
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Product Folder Links: OPA861
OPA861
www.ti.com
SBOS338G –AUGUST 2005–REVISED MAY 2013
ELECTRICAL CHARACTERISTICS: VS = +5V
RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
MIN/MAX OVER TEMPERATURE
TYP
0°C to
70°C(3)
–40°C to
MIN/
MAX
TEST
PARAMETER
CONDITIONS
+25°C
+25°C(2)
+85°C(3)
UNITS
LEVEL(1)
OTA—Open-Loop (see Figure 33)
AC PERFORMANCE
G = +5, VO = 200mVPP
,
Bandwidth
73
72
72
70
MHz
min
B
RL = 500Ω
G = +5, VO = 1VPP
73
410
4.4
MHz
V/µs
ns
typ
min
typ
C
B
C
Slew Rate
G = +5, VO = 2.5V Step
VO = 1V Step
G = +5, VO = 2VPP, 5MHz
RL = 500Ω
395
390
390
Rise Time and Fall Time
Harmonic Distortion
2nd-Harmonic
–67
–57
2.4
1.7
5.2
–55
–50
3.0
–54
–49
3.3
–54
–48
3.4
dB
max
max
max
max
max
B
B
B
B
B
3rd-Harmonic
RL = 500Ω
dB
Base Input Voltage Noise
Base Input Current Noise
Emitter Input Current Noise
OTA DC PERFORMANCE(4) (see Figure 33)
f > 100kHz
nV/√Hz
pA/√Hz
pA/√Hz
f > 100kHz
2.4
2.45
16.6
2.5
f > 100kHz
15.3
17.5
Minimum OTA Transconductance (gm
)
VO = ±10mV, RC = 50Ω, RE = 0Ω
VO = ±10mV, RC = 50Ω, RE = 0Ω
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, VC = 0V
85
85
±3
70
67
145
±15
±67
±6
65
mA/V
mA/V
mV
min
max
max
max
max
max
max
max
typ
A
A
A
B
A
B
A
B
C
Maximum OTA Transconductance (gm
)
140
±12
150
B-Input Offset Voltage
±20
Average B-Input Offset Voltage Drift
B-Input Bias Current
±120
±6.6
±25
μV/°C
μA
±1
±5
Average B-Input Bias Current Drift
E-Input Bias Current
±20
±125
±500
nA/°C
μA
±30
±15
±100
±140
±600
Average E-Input Bias Current Drift
C-Output Bias Current
VB = 0V, VC = 0V
nA/°C
μA
VB = 0V, VC = 0V
OTA INPUT (see Figure 33)
Most Positive B-Input Voltage
Least Positive B-Input Voltage
B-Input Impedance
4.2
0.8
3.7
1.3
3.6
1.4
3.6
1.4
V
min
max
typ
B
B
C
B
B
V
kΩ || pF
Ω
455 || 2.1
11.8
Min E-Input Resistance
14.4
7.1
14.9
6.9
15.4
6.7
max
min
Max E-Input Resistance
11.8
Ω
OTA OUTPUT
Maximum E-Output Voltage Compliance
Minimum E-Output Voltage Compliance
E-Output Current, Sinking/Sourcing
Maximum C-Output Voltage Compliance
Minimum C-Output Voltage Compliance
C-Output Current, Sinking/Sourcing
C-Output Impedance
IE = ±1mA
IE = ±1mA
VE = 0
4.2
0.8
3.7
1.3
±7
3.6
1.4
3.6
1.4
V
V
min
max
min
min
max
min
typ
A
A
A
A
A
A
C
±8
±6.5
3.9
±6.5
3.9
mA
V
IC = ±1mA
IC = ±1mA
VC = 0
4.7
4.0
1.0
±7
0.3
1.1
1.1
V
±8
±6.5
±6.5
mA
kΩ || pF
54 || 2
(1) Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
(2) Junction temperature = ambient for +25°C specifications.
(3) Junction temperature = ambient at low temperature limit; junction temperature = ambient + 3°C at high temperature limit for over
temperature specifications.
(4) Current is considered positive out of node.
Copyright © 2005–2013, Texas Instruments Incorporated
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Product Folder Links: OPA861
OPA861
SBOS338G –AUGUST 2005–REVISED MAY 2013
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = +5V (continued)
RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
MIN/MAX OVER TEMPERATURE
TYP
0°C to
70°C(3)
–40°C to
MIN/
MAX
TEST
PARAMETER
CONDITIONS
+25°C
+25°C(2)
+85°C(3)
UNITS
LEVEL(1)
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage
Minimum Operating Voltage
Maximum Quiescent Current
Minimum Quiescent Current
OTA Power-Supply Rejection Ratio (+PSRR)
THERMAL CHARACTERISTICS
Specification: ID, IDBV
5
V
V
typ
max
min
max
min
max
C
A
B
A
A
A
12.6
4
12.6
4
12.6
4
V
RADJ = 250Ω
RADJ = 250Ω
ΔIC/ΔVS
4.7
4.7
±20
5.2
4.2
±50
6.0
3.4
±60
6.4
3.0
±65
mA
mA
µA/V
–40 to +85
°C
typ
C
Thermal Resistance θ JA
D
SO-8
Junction-to-Ambient
Junction-to-Ambient
+125
+150
°C/W
°C/W
typ
typ
C
C
DBV SOT23-6
6
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Product Folder Links: OPA861
OPA861
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SBOS338G –AUGUST 2005–REVISED MAY 2013
TYPICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted.
OTA TRANSCONDUCTANCE vs FREQUENCY
OTA TRANSCONDUCTANCE vs QUIESCENT CURRENT
1000
150
120
90
60
30
0
IO UT
VIN = 100mVPP
Ω
RL = 50
VIN
VIN = 10mVPP
Ω
50
gm = -0.8265.IQ2 + 24.197.IQ - 1.466
Ω
50
IQ = 5.4mA (102mA/V)
IQ = 6.5mA (117mA/V)
100
IOUT
VIN
50W
IQ = 1.9mA (51mA/V)
50W
IQ = 3.4mA (79mA/V)
10
1M
10M
100M
Frequency (Hz)
1G
1
2
3
4
5
6
7
8
0
Quiescent Current (mA)
Figure 2.
Figure 3.
OTA TRANSCONDUCTANCE vs INPUT VOLTAGE
OTA TRANSFER CHARACTERISTICS
160
140
120
100
80
8
6
4
2
0
IQ = 6.5mA
IQ = 5.4mA
IQ = 6.5mA
IQ = 5.4mA
IQ = 3.4mA
IQ = 3.4mA
IQ = 1.9mA
IOUT
−
−
−
−
60
2
4
6
8
IQ = 1.9mA
VIN
40
Ω
50
Ω
50
20
Small signal around input voltage.
0
−
−
−
−
−
−
−
−
−
−
−
40
30
20
10
0
10
20
30
40
70 60 50 40 30 20 10
0
10 20 30 40 50 60 70
Input Voltage (mV)
OTA Input Voltage (mV)
Figure 4.
Figure 5.
OTA SMALL-SIGNAL PULSE RESPONSE
OTA LARGE-SIGNAL PULSE RESPONSE
0.8
0.6
0.4
0.2
0
3
2
1
0
−
−
−
−
0.2
0.4
0.6
0.8
G = +5V/V
−
G = +5V/V
1
2
3
Ω
RL = 500
Ω
RL = 500
VIN = 1VPP
fIN = 20MHz
See Figure 48
VIN = 0.25VPP
fIN = 20MHz
See Figure 48
−
−
Time (10ns/div)
Time (10ns/div)
Figure 6.
Figure 7.
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TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted.
B-INPUT RESISTANCE vs QUIESCENT CURRENT
C-OUTPUT RESISTANCE vs QUIESCENT CURRENT
120
500
110
100
90
490
480
470
460
450
440
430
80
70
60
50
40
0
1
2
3
4
5
6
7
8
0
1
2
3
4
5
6
7
8
Quiescent Current (mA)
Quiescent Current (mA)
Figure 8.
Figure 9.
E-OUTPUT RESISTANCE vs QUIESCENT CURRENT
INPUT VOLTAGE AND CURRENT NOISE DENSITY
60
50
40
30
20
10
0
100
√
E−Input Current Noise (5.2pA/ Hz)
10
√
B−Input Voltage Noise (2.4nV/ Hz)
√
B−Input Current Noise (1.65pA/ Hz)
1
100
1k
10k 100k 1M 10M
0
1
2
3
4
5
6
7
8
Quiescent Current (mA)
Frequency (Hz)
Figure 10.
Figure 11.
1MHz OTA VOLTAGE AND CURRENT NOISE DENSITY
QUIESCENT CURRENT vs RADJ
vs QUIESCENT CURRENT ADJUST RESISTOR
16
8
7
6
5
4
3
2
1
0
√
E−Input Current Noise (pA/ Hz)
14
12
10
8
√
B−Input Voltage Noise (nV/ Hz)
6
√
B−Input Current Noise (pA/ Hz)
4
2
0
0
200 400 600 800 1000 1200 1400 1600 1800 2000
0.1
1
10
100
1k
10k
)
100k
Ω
Quiescent Current Adjust Resistor (
)
Ω
Quiescent Current Adjust Resistor (
Figure 12.
Figure 13.
8
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Product Folder Links: OPA861
OPA861
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SBOS338G –AUGUST 2005–REVISED MAY 2013
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted.
B-INPUT OFFSET VOLTAGE AND BIAS CURRENT
vs TEMPERATURE
QUIESCENT CURRENT vs TEMPERATURE
6
4
2
0
2
4
6
3
2
1
0
−
−
−
9
8
7
6
5
4
3
B−Input Offset Voltage
−
−
−
1
2
3
B−Input Bias Current
−
−
−
−
20
40
20
0
20
40
60
80
100
120
40
0
20
40
60
80
100
120
_
Ambient Temperature ( C)
_
Ambient Temperature ( C)
Figure 14.
Figure 15.
C-OUTPUT BIAS CURRENT vs TEMPERATURE
IQ/IADS Ratio vs RADJ
40
30
20
10
0
350
300
250
200
150
100
50
Five Representative Units
IQ
3
= -5E-18 x RADJ4 + 1E-12 x RADJ - 7E-08 x RADJ2 + 0.0046 x RADJ + 37.8
IADJ
−
−
−
−
10
20
30
40
IQ = Quiescent Current.
IADJ = Current flowing out of IQ adjust pin.
0
−
−
0.01
0.1
1
10
100
1k
10k
40
20
0
20
40
60
80
100
120
100k
_
Ambient Temperature ( C)
Quiescent Current Adjust Resistor (W )
Figure 16.
Figure 17.
QUIESCENT CURRENT vs ADJUST PIN BIAS CURRENT
250
200
150
100
50
0
0.01
0.1
1
10
100
1k
10k
100k
Quiescent Current Adjust Resistor (W)
Figure 18.
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TYPICAL CHARACTERISTICS: VS = +5V
At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted.
OTA TRANSCONDUCTANCE vs FREQUENCY
OTA TRANSCONDUCTANCE vs IQ
100
150
120
90
60
30
0
IOUT
IQ = 5.8mA
(93mA/V)
IQ = 4.7mA (80mA/V)
IQ = 3.1mA (60mA/V)
VIN
Ω
50
Ω
50
IQ = 1.65mA (37mA/V)
IOUT
VIN
Ω
50
Ω
50
Ω
RL = 50
VIN = 100mVPP
VIN = 10mVPP
10
1
10
100
1k
0
1
2
3
4
5
6
7
Frequency (Hz)
Quiescent Current (mA)
Figure 19.
Figure 20.
OTA TRANSCONDUCTANCE vs INPUT VOLTAGE
OTA TRANSFER CHARACTERISTICS
120
100
80
60
40
20
0
6
4
2
0
IQ = 5.8mA
IQ = 4.7mA
IQ = 5.8mA
IQ = 3.1mA
IQ = 3.1mA
IQ = 4.7mA
IQ = 1.65mA
IQ = 1.65mA
I
OUT
−
−
−
2
4
6
V
IN
Ω
50
Ω
50
Small−signal around input voltage.
−
−
−
−
−
−
−
−
20 10
30
20
10
0
10
20
30
50
40
30
0
10
20
30
40
50
Input Voltage (mV)
OTA Input Voltage (mV)
Figure 21.
Figure 22.
OTA SMALL-SIGNAL PULSE RESPONSE
OTA LARGE-SIGNAL PULSE RESPONSE
0.20
0.15
0.10
0.05
0
2.0
1.5
1.0
0.5
0
−
−
−
−
−
−
−
−
0.05
0.10
0.15
0.20
0.5
1.0
1.5
2.0
G = +5V/V
G = +5V/V
Ω
RL = 500
VIN = 0.7VPP
fIN = 20MHz
Ω
RL = 500
VIN = 0.07VPP
fIN = 20MHz
Time (10ns/div)
Time (10ns/div)
Figure 23.
Figure 24.
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TYPICAL CHARACTERISTICS: VS = +5V (continued)
At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted.
B-INPUT RESISTANCE vs QUIESCENT CURRENT
C-OUTPUT RESISTANCE vs QUIESCENT CURRENT
500
490
480
470
460
450
440
430
420
120
110
100
90
80
70
60
50
40
0
1
2
3
4
5
6
7
0
1
2
3
4
5
6
7
Quiescent Current (mA)
Quiescent Current (mA)
Figure 25.
Figure 26.
E-OUTPUT RESISTANCE vs QUIESCENT CURRENT
QUIESCENT CURRENT vs RADJ
60
7
6
5
4
3
2
1
0
50
40
30
20
10
0
0
1
2
3
4
5
6
7
0.1
1
10
100
1k
10k
)
100k
Quiescent Current (mA)
Ω
Quiescent Current Adjust Resistor (
Figure 27.
Figure 28.
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APPLICATION INFORMATION
The
OPA861
is
a
versatile
monolithic
TRANSCONDUCTANCE (OTA) SECTION—AN
OVERVIEW
transconductance amplifier designed for wide-
bandwidth systems, including high-performance
video, RF, and IF circuitry. The operation of the
OPA861 is discussed in the OTA (Operational
Transconductance Amplifier) section of this data
sheet. Over the years and depending on the writer,
the OTA section of an op amp has been referred to
as a Diamond Transistor, Voltage-Controlled Current
source, Transconductor, Macro Transistor, or positive
The symbol for the OTA section is similar to a
transistor (see Figure 29). Applications circuits for the
OTA look and operate much like transistor
circuits—the transistor is also a voltage-controlled
current source. Not only does this characteristic
simplify the understanding of application circuits, it
aids the circuit optimization process as well. Many of
the same intuitive techniques used with transistor
designs apply to OTA circuits. The three terminals of
the OTA are labeled B, E, and C. This labeling calls
attention to its similarity to a transistor, yet draws
distinction for clarity. While the OTA is similar to a
transistor, one essential difference is the sense of the
C-output current: it flows out the C terminal for
positive B-to-E input voltage and in the C terminal for
negative B-to-E input voltage. The OTA offers many
advantages over a discrete transistor. The OTA is
self-biased, simplifying the design process and
reducing component count. In addition, the OTA is far
more linear than a transistor. Transconductance of
the OTA is constant over a wide range of collector
second-generation
Corresponding symbols for these terms are shown in
Figure 29.
current
conveyor
(CCII+).
C
3
VIN1
B
1
IOUT
VIN2
2
E
Diamond
Transistor
Transconductor
(used here)
Voltage−Controlled
Current Source
C
currents—this feature implies
improvement of linearity.
a
fundamental
VIN1
VIN2
B
Z
IOUT
CCII+
E
BASIC CONNECTIONS
Current Conveyor II+
Macro Transistor
Figure 30 shows basic connections required for
operation. These connections are not shown in
subsequent circuit diagrams. Power-supply bypass
capacitors should be located as close as possible to
the device pins. Solid tantalum capacitors are
generally best.
Figure 29. Symbols and Terms
Regardless of its depiction, the OTA section has a
high-input impedance (B-input), a low-input/output
impedance (E-input), and a high-impedance current
source output (C-output).
RQ = 250W, roughly sets IQ = 5.4mA.
RC
1
8
+5V(1)
RE
0.1mF
+VS
RADJ
2
7
RS
250W
+
(25W to 200W)
2.2mF
VIN
3
4
6
Solid Tantalum
5
-VS
-5V(1)
0.1mF
2.2mF
+
Solid
Tantalum
NOTE: (1) VS = ±6.5V absolute maximum.
Figure 30. Basic Connections
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QUIESCENT CURRENT CONTROL PIN
With this control loop, quiescent current will be nearly
constant with temperature. Since this method differs
from the temperature-dependent behavior of the
internal current source, other temperature-dependent
behavior may differ from that shown in the Typical
Characteristics. The circuit of Figure 31 will control
the IQ of the OPA861 somewhat more accurately than
with a fixed external resistor, RQ. Otherwise, there is
no fundamental advantage to using this more
complex biasing circuitry. It does, however,
demonstrate the possibility of signal-controlled
quiescent current. This capability may suggest other
possibilities such as AGC, dynamic control of AC
behavior, or VCO.
The quiescent current of the transconductance
portion of the OPA861 is set with a resistor, RADJ
,
connected from pin to –VS. The maximum
1
quiescent current is 6mA. RADJ should be set
between 50Ω and 1kΩ for optimal performance of the
OTA section. This range corresponds to the 5mA
quiescent current for RADJ = 50Ω, and 1mA for RADJ
=
1kΩ. If the IQ adjust pin is connected to the negative
supply, the quiescent current will be set by the 250Ω
internal resistor.
Reducing or increasing the quiescent current for the
OTA section controls the bandwidth and AC behavior
as well as the transconductance. With RADJ = 250Ω,
this sets approximately 5.4mA total quiescent current
at 25°C. It may be appropriate in some applications to
trim this resistor to achieve the desired quiescent
current or AC performance.
BASIC APPLICATIONS CIRCUITS
Most applications circuits for the OTA section consist
of a few basic types, which are best understood by
analogy to a transistor. Used in voltage-mode, the
OTA section can operate in three basic operating
states—common emitter, common base, and
common collector. In the current-mode, the OTA can
be useful for analog computation such as current
amplifier, current differentiator, current integrator, and
current summer.
Applications circuits generally do not show the
resistor RQ, but it is required for proper operation.
With
a
fixed RADJ resistor, quiescent current
increases with temperature (see Figure 12 in the
Typical Characteristics section). This variation of
current with temperature holds the transconductance,
gm, of the OTA relatively constant with temperature
(another advantage over a transistor).
Common-E Amplifier or Forward Amplifier
Figure
32
compares
the
common-emitter
It is also possible to vary the quiescent current with a
control signal. The control loop in Figure 31 shows
1/2 of a REF200 current source used to develop
100mV on R1. The loop forces 125mV to appear on
R2. Total quiescent current of the OPA861 is
approximately 37 × I1, where I1 is the current made to
flow out of pin 1.
configuration for a BJT with the common-E amplifier
for the OTA section. There are several advantages in
using the OTA section in place of a BJT in this
configuration. Notably, the OTA does not require any
biasing, and the transconductance gain remains
constant over temperature. The output offset voltage
is close to 0, compared with several volts for the
common-emitter amplifier.
The gain is set in a similar manner as for the BJT
equivalent with Equation 1:
V+
OPA861
RL
1/2 REF200
G +
1
g
µ
100
A
m ) RE
IQ Adjust
I1
(1)
R1
1
Ω
1.25k
Just as transistor circuits often use emitter
degeneration, OTA circuits may also use
degeneration. This option can be used to reduce the
effects that offset voltage and offset current might
otherwise have on the DC operating point of the OTA.
The E-degeneration resistor may be bypassed with a
large capacitor to maintain high AC gain. Other
circumstances may suggest a smaller value capacitor
used to extend or optimize high-frequency
performance.
R2
425
Ω
TLV2262
Figure 31. Optional Control Loop for Setting
Quiescent Current
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The forward amplifier shown in Figure 33 and
Figure 34 corresponds to one of the basic circuits
used to characterize the OPA861. Extended
characterization of this topology appears in the
Typical Characteristics section of this datasheet.
V+
RS
RL
VO
VO
VI
Inverting Gain
VOS = Several Volts
8
R1
RC
C
Ω
160
Ω
500
RS
RE
3
B
OPA861
VI
E
2
−
V
G = 5V/V
IQ = 5.4mA
RE
(a) Transistor Common−Emitter Amplifier
Transconductance varies over temperature.
Ω
78
8
VO
Figure 33. Forward Amplifier Configuration and
Test Circuit
C
Ω
100
3
B
RL
OPA861
VI
E
2
RL1
Noninverting Gain
VOS = 0V
RE
VO
Network
8
Analyzer
3
(b) OTA Common−E Amplifier
Transconductance remains constant over temperature.
RIN
50W
OTA
R1
RL2
rE
100W
Figure 32. Common-Emitter vs Common-E
Amplifier
2
VI
RL = RL1 + RL2 || RIN
RE
The transconductance of the OTA with degeneration
can be calculated by Equation 2:
1
gm_deg
+
1
g
RL
1
m ) RE
rE
=
=
(2)
G =
gm
RE + rE
A positive voltage at the B-input, pin 3, causes a
positive current to flow out of the C-input, pin 8. This
1
At IQ = 5.4mA
rE
= 10.5W
gives
a noninverting gain where the circuit of
95mA/V
Figure 32a is inverting. Figure 32b shows an amplifier
connection of the OPA861, the equivalent of a
common-emitter transistor amplifier. Input and output
can be ground-referenced without any biasing. The
amplifier is non-inverting because of the sense of the
output current.
RL
G =
at IQ = 5.4mA
RE + 10.5W
Figure 34. Forward Amplifier Design Equations
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Common-C Amplifier
This low impedance can be converted to a high
impedance by inserting the buffer amplifier in series.
Figure 35b shows the OPA861 connected as an E-
follower—a voltage buffer. It is interesting to notice
that the larger the RE resistor, the closer to unity gain
the buffer will be. If the OPA861 is to be used as a
buffer, use RE ≥ 500Ω for best results. For the
OPA861 used as a buffer, the gain is given by
Equation 3:
Current-Mode Analog Computations
As mentioned earlier, the OPA861 can be used
advantageously for analog computation. Among the
application possibilities are functionality as a current
amplifier, current differentiator, current integrator,
current summer, and weighted current summer.
Table 1 lists these different uses with the associated
transfer functions.
1
G +
[ 1
1
R
1 ) gm
E
(3)
These functions can easily be combined to form
active filters. Some examples using these current-
mode functions are shown later in this document.
V+
G = 1
VOS = 0.7V
VI
V+
VO
RL
RE
VO
Noninverting Gain
VOS = Several Volts
−
V
(a) Transistor Common−Collector Amplifier
(Emitter Follower)
R1
1
G +
+ 1
VIN
1
R
1 ) gm
E
RE
1
8
ǒ Ǔ
RO +
ø RE
V-
gm
C
Ω
100
(a) Transistor Common-Base Amplifier
3
B
G = 1
VOS = 0V
OPA861
VI
RL
RL
RE
E
2
G =
= -
1
RE
RE +
gm
VO
8
VO
(b) OTA Common−C Amplifier
(Buffer)
C
100W
3
B
Inverting Gain
VOS = 0V
OPA861
E
2
Figure 35. Common-Collector vs Common-C
Amplifier
RL
RE
A low value resistor in series with the B-input is
recommended. This resistor helps isolate trace
parasitic from the inputs, reduces any tendency to
oscillate, and controls frequency response peaking.
Typical resistor values are from 25Ω to 200Ω.
VIN
(b) OTA Common-B Amplifier
Figure 36. Common-Base Transistor vs
Common-B OTA
Common-B Amplifier
Figure 36 shows the Common-B amplifier. This
configuration produces an inverting gain and a low
impedance input. Equation 4 shows the gain for this
configuration.
RL
RL
RE
G +
[ *
1
RE ) gm
(4)
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Table 1. Current-Mode Analog Computation Using the OTA Section
FUNCTIONAL ELEMENT
TRANSFER FUNCTION
IMPLEMENTATION WITH THE OTA SECTION
IOUT
IIN
R1
R2
R1
Current Amplifier
IOUT
+
IIN
R2
IOUT
IIN
1
IOUT
+
C
Current Integrator
R
ŕ
C R IINdt
IOUT
n
Current Summer
IOUT + 1 S Ij
j+1
I1
I2
In
IOUT
n
Rj
Weighted Current Summer
R1
Rn
IOUT + 1 S Ij
R
R
j+1
R
I1
In
OPA861 APPLICATIONS
Control-Loop Amplifier
DC-Restore Circuit
A new type of control loop amplifier for fast and
precise control circuits can be designed with the
OPA861. The circuit of Figure 37 illustrates a series
connection of two voltage control current sources that
have an integral (and at higher frequencies, a
proportional) behavior versus frequency. The control
loop amplifiers show an integrator behavior from DC
to the frequency represented by the RC time constant
of the network from the C-output to GND. Above this
frequency, they operate as an amp with constant
gain. The series connection increases the overall gain
to about 110dB and thus minimizes the control loop
deviation. The differential configuration at the inputs
enables one to apply the measured output signal and
the reference voltage to two identical high-impedance
inputs. The output buffer decouples the C-output of
the second OTA in order to insure the AC
performance and to drive subsequent output stages.
The OPA861 can be used advantageously with an
operational amplifier, here the OPA656, as a DC-
restore circuit. Figure 38 illustrates this design.
Depending on the collector current of the
transconductance amplifier (OTA) of the OPA861, a
switching function is realized with the diodes D1 and
D2.
When the C-output is sourcing current, the capacitor
C1 is being charged. When the C-output is sinking
current, D1 is turned off and D2 is turned on, letting
the voltage across C1 be discharged through R2.
The condition to charge C1 is set by the voltage
difference between VREF and VOUT. For the OTA C-
output to source current, VREF has to be greater than
VOUT. The rate of charge of C1 is set by both R1 and
C1. The discharge rate is given by R2 and C1.
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6
8
5
BUF602
VOUT
3
8
2
Ω
Ω
180
2
10pF
10pF
VREF
3
Ω
Ω
33
Ω
Ω
33
10
10
180
6
VIN
Figure 37. Control-Loop Amplifier Using Three OPA861s
C1
100pF
Ω
20
JFET−Input, Wideband
VOUT
VIN
OPA656
R2
D1
D2
Ω
100k
Ω
20
D1, D2 = 1N4148
Ω
RQ = 1k
R1
CCII
Ω
40.2
E
2
8
C
The OTA amplifier works as a current conveyor (CCII) in this circuit, with a current gain of 1.
R1 and C1 set the DC restoration time constant.
B
3
D1 adds a propagation delay to the DC restoration.
R2 and C1 set the decay time constant.
R2
Ω
100
VREF
Figure 38. DC Restorer Circuit
Negative Impedance Converter Filter: Low-Pass
The transfer function is shown in Equation 5:
VOUT
Filter
1
The OPA861 can be used as a negative impedance
converter to realize the low-pass filer shown in
Figure 39.
=
1 + sR(C1 + C2) + s2C1C2R2
VIN
(5)
with:
1
R
w0 +
Ǹ
C1C2 R
C1
R
C1C2
VIN
VOUT
Q =
C2
C1 + C2
Figure 39. Low-Pass Negative Impedance
Converter Filter
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Differential Line Driver/Receiver
The input impedance is shown in Equation 6:
1 ) sRC
1 ) 2sRC
1
2sC
The wide bandwidth and high slew rate of the
OPA861 current-mode amplifier make it an ideal line
driver. The circuit in Figure 42 makes use of two
OPA861s to realize a single-ended to differential
conversion. The high-impedance current source
output of the OPA861 allows it to drive low-
impedance or capacitive loads without series
resistances and avoids any attenuation that would
have otherwise occured in the resistive network.
ZIN
+
) R
(6)
Figure 40 shows the frequency responses for low-
pass, Butterworth filters set at 20kHz and 10MHz.
For the 20kHz filter, set
R
to 1kΩ and
1
2
C1 + C2 + 5.6mF
. For the 10MHz filter, the
parasitic capacitance at the output pin needs to be
taken into consideration. In the example of Figure 40,
the parasitic is 3pF, which gives us the settings of R
= 1.13kΩ, C1 = 10pF, and C2 = 17pF.
The OPA861 used as a differential receiver exhibits
excellent common-mode rejection ratio, as can be
seen in Figure 41.
0
0
−
−
−
−
−
−
−
−
−
10
20
30
40
50
60
70
80
90
−
−
−
−
−
−
−
−
10
20
30
40
50
60
70
80
−
100
0.001
1k
10k
100k
1M
10M
100M
1G
0.01
0.1
1
10
100
Frequency (Hz)
Frequency (MHz)
Figure 41. Differential Driver Common-Mode
Rejection Ratio for 2VPP Input Signals
Figure 40. Small-Signal Frequency Response for
a Low-Pass Negative Impedance Converter Filter
Ω
To 50 Load
Ω
Ω
50
50
VIN
Ω
50
Ω
10
Ω
100
Ω
Ω
10
50
Ω
50
Figure 42. Twisted-Pair Differential Driver and Receiver with the OPA861
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ACTIVE FILTERS USING THE OPA861 IN
CURRENT CONVEYOR STRUCTURE
of the operational amplifier becomes a negative
second type of Current Conveyor (CCII), as shown in
Figure 43. Both arrangements have identical transfer
functions and the same level of sensitivity to
deviations. The most recent implementation of active
filters in a Current-Conveyor structure produced a
second-order Bi-Quad filter. The value of the
resistance in the emitter of the Diamond Transistor
controls the filter characteristic. For more information,
refer to application note SBOS047, New Ultra High-
Speed Circuit Techniques with Analog ICs.
One further example of the versatility of the Diamond
Transistor and Buffer is the construction of high-
frequency (> 10MHz) active filters. Here, the Current
Conveyor structure, shown in Figure 43, is used with
the Diamond Transistor as a Current Conveyor.
IOUT
VOUT
E
B
C
+1
CCII−
C
C
VIN
R
R
R
R
IIN
C/2
C/2
Reciprocal Networks
4KQ2/R2C2
=
s2 + 2/RC[2Q(1 K) + 1]s + 4KQ2/R2C2
VOUT IOUT
=
+
T(s) =
VIN
IIN
−
VIN
VOUT
IOUT
IIN
N
N
−
Figure 43. Current Conveyor
VOUT
IOUT
=
VIN
IIN
Interreciprocal Networks
The method of converting RC circuit loops with
operational amplifiers in Current Conveyor structures
is based upon the adjoint network concept. A network
is reversible or reciprocal when the transfer function
does not change even when the input and output
have been exchanged. Most networks, of course, are
nonreciprocal. The networks of Figure 44, perform
interreciprocally when the input and output are
exchanged, while the original network, N, is
exchanged for a new network NA. In this case, the
transfer function remains the same, and NA is the
adjoing network. It is easy to construct an adjoint
network for any given circuit, and these networks are
the base for circuits in Current-Conveyor structure.
Individual elements can be interchanged according to
the list in Figure 45. Voltage sources at the input
become short circuits, and the current flowing there
becomes the output variable. In contrast, the voltage
output becomes the input, which is excitated by a
current source. The following equation describes the
+
VIN
VOUT
IOUT
IIN
N
NA
−
Figure 44. Networks
Element
VIN
Adjoint
IOUT
1
1
2
2
1
1
2
2
Signal
Sources
IIN
−
+
VOUT
R
R
1
1
2
2
1
1
2
2
Passive
Elements
C
C
1
1
3
4
3
4
+
Controlled
Sources
µV
µI
I
V
−
Figure 45. Individual Elements in the Current
Conveyor
R3
R2
interreciprocal features of the circuit: VOUT/VIN
=
VIN
VOUT
IOUT/IIN. Resistances and capacitances remain
unchanged. In the final step, the operational amplifier
with infinite input impedance and 0Ω output
impedance is transformed into a current amplifier with
0Ω input impedance and infinite output impedance. A
Diamond Transistor with the base at ground comes
quite close to an ideal current amplifier. The well-
known Sallen-Key low-pass filter with positive
feedback, is an example of conversion into Current-
Conveyor structure, see Figure 46. The positive gain
BUF602
C1
C2
R1
R1M
R2M
RB1
RB2
RB3
R1S
R2S
R3S
Figure 46. Universal Active Filter
Transfer Function
Filter Characteristics
Five filter types can be made with this structure:
The transfer function of the universal active filter of
Figure 46 is shown in Equation 7.
•
•
•
•
•
For a low-pass filter, set R2 = R3 = ∞,
For a high-pass filter, set R1 = R2 = ∞,
For a bandpass filter, set R1 = R3 = ∞,
For a band rejection filter, set R2 = ∞; R1 = R3,
For an all-pass filter, set R1 = R1S; R2 = R2S; and
R3 = R3S.
R
R
1M
1
s2C1C2R1M 2M ) sC1
)
R
3
R
R
VOUT
VIN
2
1
( )
F p +
+
R
R
s2C1C2R1M 2M ) sC1
)
1M
1
R
R
R
3S
2S
1S
(7)
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A few designs for a low-pass filter are shown in
Figure 47 and Table 2.
High-CMRR, Moderate Precision, Differential
I/O ADC Driver
The circuit shown in Figure 48 depicts an ADC driver
implemented with two OPA861s. Since the gain is set
here by the ratio of the internal 600Ω resistors and
RE, its accuracy will only be as good as the input
resistor of the ADS5272. The small-signal frequency
response for this circuit has 150MHz at –3dB
bandwidth for a gain of approximately 5.6dB, as
shown in Figure 49. The advantage of this circuit lies
in its high CMRR to 100kHz; see Figure 50. This
circuit also has more than 10 bits of linearity.
Table 2. Component Values for Filters Shown In
Figure 47
fO
R
RO
CO
2nF
1MHz
20MHz
50MHz
150
150
150
100
100
100
112.5pF
25pF
3
0
50MHz Filter
-3
-6
-9
ADS5272
OPA861
-12
-15
-18
-21
-24
-27
-30
-33
-36
-39
-42
1MHz Filter
Ω
600
VIN1
20MHz Filter
For All Filters:
RE
600
R2 = R3 = ¥
VCM
Ω
R1 = R1S = R2S = 1/2 R3S = R
R1M = R2M = R0
-45 C1 = C2 = C0
-48
OPA861
10k
100k
1M
10M
100M
1G
VIN2
Ω
600
Frequency (Hz)
Figure 47. Butterworth Low-Pass Filter with the
Universal Active Filter
Figure 48. High CMRR, Moderate Precision,
Differential I/O ADC Driver
The advantages of building active filters using a
Current Conveyor structure are:
•
The increase in output resistance of operational
amplifiers at high frequencies makes it difficult to
construct feedback filter structures (decrease in
stop-band attenuation).
6
5.6dB
3
•
All filter coefficients are represented by
resistances, making it possible to adjust the filter
frequency response without affecting the filter
coefficients.
0
−
−
−
3
6
9
•
•
The capacitors which determine the frequency are
located between the ground and the current
source outputs and are thus grounded on one
side. Therefore, all parasitic capacitances can be
viewed as part of these capacitors, making them
easier to comprehend.
The features which determine the frequency
characteristics are currents, which charge the
integration capacitors. This situation is similar to
the transfer characteristic of the Diamond
Transistor.
1M
10M
100M
1G
Frequency (Hz)
Figure 49. ADC Driver, Small-Signal Frequency
Response
20
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OPA861
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SBOS338G –AUGUST 2005–REVISED MAY 2013
NOISE PERFORMANCE
75
70
65
60
55
50
45
40
35
30
25
The OTA noise model consists of three elements: a
voltage noise on the B-input; a current noise on the
B-input; and a current noise on the E-input. Figure 51
shows the OTA noise analysis model with all the
noise terms included. In this model, all noise terms
are taken to be noise voltage or current density terms
in either nV/√Hz or pA/√Hz.
Input−Referred
en
VO
RL
20
1k
10k
100k
1M
10M
100M
1G
RS
RG
Frequency (Hz)
ibn
ibi
√
√
4kTRS
4kTRS
Figure 50. CMRR of the ADC Driver
DESIGN-IN TOOLS
Figure 51. OTA Noise Analysis Model
DEMONSTRATION BOARDS
The total output spot noise voltage can be computed
as the square root of the sum of all squared output
noise voltage contributors. Equation 8 shows the
general form for the output noise voltage using the
terms shown in Figure 51.
A printed circuit board (PCB) is available to assist in
the initial evaluation of circuit performance using the
OPA861. This module is available free, as an
unpopulated PCB delivered with descriptive
documentation. The summary information for the
board is shown below:
2
RL
[eN2 + (RSiBN)2 + 4kTRS + (RGiBI)2 + 4kTRG]
eO
=
1
gm
RG
+
LITERATURE
REQUEST
NUMBER
BOARD PART
NUMBER
PRODUCT
PACKAGE
(8)
OPA861ID
SO-8
DEM-OTA-SO-1A
SBOU035
THERMAL ANALYSIS
The board can be requested on the Texas
Instruments web site (www.ti.com).
Maximum desired junction temperature will set the
maximum allowed internal power dissipation as
described below. In no case should the maximum
junction temperature be allowed to exceed 150°C.
MACROMODELS AND APPLICATIONS
SUPPORT
Computer simulation of circuit performance using
SPICE is often useful when analyzing the
performance of analog circuits and systems. This
principle is particularly true for Video and RF amplifier
circuits where parasitic capacitance and inductance
can have a major effect on circuit performance. A
SPICE model for the OPA861 is available through the
Texas Instruments web page (www.ti.com). These
models do a good job of predicting small-signal AC
and transient performance under a wide variety of
operating conditions. They do not do as well in
predicting the harmonic distortion. These models do
not attempt to distinguish between the package types
in their small-signal AC performance.
Operating junction temperature (TJ) is given by
TA + PD
(PD) is the sum of quiescent power (PDQ) and
additional power dissipated in the output stage (PDL
×
θ JA. The total internal power dissipation
)
to deliver output current. Quiescent power is simply
the specified no-load supply current times the total
supply voltage across the part. PDL will depend on the
required output signal and load but would, for the
OPA861 be at a maximum when the maximum IO is
being driven into a voltage source that puts the
maximum voltage across the output stage. Maximum
IO is 15mA times a 9V maximum across the output.
Note that it is the power in the output stage and not
into the load that determines internal power
dissipation.
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As a worst-case example, compute the maximum TJ
using an OPA861IDBV in the circuit of Figure 32b
operating at the maximum specified ambient
temperature of +85°C and driving a –1V voltage
reference.
c) Careful selection and placement of external
components will preserve the high-frequency
performance of the OPA861. Resistors should be a
very low reactance type. Surface-mount resistors
work best and allow a tighter overall layout. Metal film
or carbon composition, axially-leaded resistors can
also provide good high-frequency performance.
Again, keep their leads and PC board traces as short
as possible. Never use wirewound type resistors in a
high-frequency application.
PD = 10V × 5.4mA + (15mA × 9V) = 185mW
Maximum TJ = +85°C + (0.19W × 150°C/W) = 114°C.
Although this is still well below the specified
maximum junction temperature, system reliability
considerations may require lower tested junction
temperatures. The highest possible internal
dissipation will occur if the load requires current to be
forced into the output for positive output voltages or
sourced from the output for negative output voltages.
This puts a high current through a large internal
voltage drop in the output transistors.
d) Connections to other wideband devices on the
board may be made with short, direct traces or
through onboard transmission lines. For short
connections, consider the trace and the input to the
next device as a lumped capacitive load. Relatively
wide traces (50mils to 100mils) should be used,
preferably with ground and power planes opened up
around them.
BOARD LAYOUT GUIDELINES
e) Socketing a high-speed part like the OPA861 is
not recommended. The additional lead length and
pin-to-pin capacitance introduced by the socket can
create an extremely troublesome parasitic network
that makes it almost impossible to achieve a smooth,
stable frequency response. Best results are obtained
by soldering the OPA861 onto the board.
Achieving optimum performance with
a
high-
frequency amplifier like the OPA861 requires careful
attention to board layout parasitics and external
component types. Recommendations that will
optimize performance include:
a) Minimize parasitic capacitance to any AC ground
for all of the signal I/O pins. Parasitic capacitance on
the inverting input pin can cause instability: on the
noninverting input, it can react with the source
impedance to cause unintentional bandlimiting. To
reduce unwanted capacitance, a window around the
signal I/O pins should be opened in all of the ground
and power planes around those pins. Otherwise,
ground and power planes should be unbroken
elsewhere on the board.
INPUT AND ESD PROTECTION
The OPA861 is built using a very high-speed
complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very
small geometry devices. These breakdowns are
reflected in the Absolute Maximum Ratings table. All
device pins are protected with internal ESD protection
diodes to the power supplies as shown in Figure 52.
b) Minimize the distance (< 0.25") from the power-
supply pins to high-frequency 0.1µF decoupling
capacitors. At the device pins, the ground and power-
plane layout should not be in close proximity to the
signal I/O pins. Avoid narrow power and ground
traces to minimize inductance between the pins and
the decoupling capacitors. The power-supply
connections should always be decoupled with these
capacitors. An optional supply decoupling capacitor
(0.1µF) across the two power supplies (for bipolar
operation) will improve 2nd-harmonic distortion
performance. Larger (2.2µF to 6.8µF) decoupling
capacitors, effective at lower frequency, should also
be used on the main supply pins. These may be
placed somewhat farther from the device and may be
shared among several devices in the same area of
the PC board.
+VCC
External
Pin
Internal
Circuitry
−
VCC
Figure 52. Internal ESD Protection
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30mA
continuous current. Where higher currents are
possible (for example, in systems with ±15V supply
parts driving into the OPA861), current-limiting series
resistors should be added into the two inputs. Keep
these resistor values as low as possible since high
values degrade both noise performance and
frequency response.
22
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OPA861
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SBOS338G –AUGUST 2005–REVISED MAY 2013
REVISION HISTORY
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision F (May 2011) to Revision G
Page
•
Changed transfer function equations in Negative Impedance Converter Filter: Low-Pass Filter section .......................... 17
Changes from Revision E (August 2008) to Revision F
Page
•
•
Updated Figure 30 .............................................................................................................................................................. 12
Updated Equation 8 ............................................................................................................................................................ 21
Changes from Revision D (August 2006) to Revision E
Page
•
Changed storage temperature range rating in Absolute Maximum Ratings table from –40°C to +125°C to –65°C to
+125°C .................................................................................................................................................................................. 2
Copyright © 2005–2013, Texas Instruments Incorporated
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Product Folder Links: OPA861
PACKAGE OPTION ADDENDUM
www.ti.com
2-May-2013
PACKAGING INFORMATION
Orderable Device
Status Package Type Package Pins Package
Eco Plan Lead/Ball Finish
MSL Peak Temp
Op Temp (°C)
Top-Side Markings
Samples
Drawing
Qty
(1)
(2)
(3)
(4)
OPA861ID
ACTIVE
SOIC
D
8
75
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
861
OPA861IDBVR
OPA861IDBVRG4
OPA861IDBVT
OBSOLETE
OBSOLETE
ACTIVE
SOT-23
SOT-23
SOT-23
DBV
DBV
DBV
6
6
6
TBD
TBD
Call TI
Call TI
Call TI
Call TI
-40 to 85
-40 to 85
-40 to 85
NSR
250
250
75
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
NSR
NSR
OPA861IDBVTG4
OPA861IDG4
OPA861IDR
ACTIVE
ACTIVE
SOT-23
SOIC
DBV
D
6
8
8
8
Green (RoHS
& no Sb/Br)
CU NIPDAU
CU NIPDAU
Call TI
Level-2-260C-1 YEAR
Level-2-260C-1 YEAR
Call TI
-40 to 85
-40 to 85
-40 to 85
-40 to 85
Green (RoHS
& no Sb/Br)
OPA
861
OBSOLETE
OBSOLETE
SOIC
D
TBD
OPA
861
OPA861IDRG4
SOIC
D
TBD
Call TI
Call TI
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a
continuation of the previous line and the two combined represent the entire Top-Side Marking for that device.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
2-May-2013
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
13-May-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
OPA861IDBVT
SOT-23
DBV
6
250
180.0
8.4
3.2
3.1
1.39
4.0
8.0
Q3
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
13-May-2013
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SOT-23 DBV
SPQ
Length (mm) Width (mm) Height (mm)
210.0 185.0 35.0
OPA861IDBVT
6
250
Pack Materials-Page 2
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