EVAL6569_技术文档

AN880

APPLICATION NOTE

®

THE L6569: A NEW HIGH VOLTAGE IC DRIVER FOR

ELECTRONIC LAMP BALLAST

by G. Calabrese and T. Castagnet

Figure 1: CFL series resonant half bridge inverter.

INTRODUCTION

Electronic lamp ballasts are now popular in both

consumer and industrial lighting. They offer power

saving, flicker free operation and reduced sizes.

Improvements to the light control and cost reduc-

tion of the ballast will broaden their market accep-

tance.

Today designers focus on reducing the cost of the

ballast, but also work to add features to the bal-

last like saving energy by dimming the light, or in-

creasing the life time with better preheat and pro-

tections. Such requirements have contributed to

the development of dedicated high voltage con-

trollers like the L6569, which are able to drive the

floating transistor of a symmetric half bridge in-

verter. This device is a simple, monolithic oscilla-

tor-half bridge driver that allows quick design of

the ballast.

Figure 2: Current and voltage of the STD3NA50

MOSFETs when driven in ZVS with

the L6569.

HIGH VOLTAGE IC DRIVERS IN BALLAST AP-

PLICATIONS

The voltage fed half bridge

I_D

V_DS

Voltage fed series resonant half bridge inverters

are currently used for Compact Fluorescent Lamp

ballasts (CFL), for Halogen Lamp transformers,

and for many European Tube Lamp (TL) ballasts.

This simple converter is preferred for new de-

signs, because it minimizes the off state voltage

of the power transistors to the peak line voltage,

and requires only one resonant choke. In addition

this choke protects the half bridge against short

circuits across lamp terminals. However overheat-

ing and overcurrent occur during open load op-

eration. The inverter robustness must be im-

proved, or some protections are required.

GND

LVG

GND

RF

GND

2 µs/dv ; 50 V/dv ; 0.1 A/dv

gle frequency with a saturable pulse transformer

(see fig. 1) to drive the transistors. This type of

design has a higher component count, a higher

tolerance on the switching frequency, and it can-

not adjust the lamp power.

The only way to design a cost effective, compact

and smart control of the lamp is to use a dedi-

cated I.C. that is able to drive the upper transistor

of an symmetric half bridge inverter. Such control-

lers require a high voltage capability for the float-

ing transistor driver [2]. MOSFETs are preferred

over Bipolar transistors as power switches be-

cause their gate driver requires a lower supply

current and a smaller silicon size [3].

The half bridge inverter operates in Zero Voltage

Switching (ZVS) resonant mode [1], to reduce the

transistor switching losses and the electromag-

netic interference generated by the output wiring

and the lamp.

Fully integrated ballast controllers

By varying the switching frequency, the half

bridge inverter is able to modulate the lamp

power. However most current designs use a sin-

1/14

February 2003

AN880 APPLICATION NOTE

and the circuit requires only 150 µA at power up.

THE L6569 AND ITS APPLICATIONS

The L6569

The L6569 integrates a high voltage Lateral

DMOS transistor in place of the usual external di-

ode [2] to charge the bootstrap capacitor for the

upper buffer. Figure 5 shows DMOS operating as

a synchronous rectifier.

The L6569 is able to directly control a symmetric

half bridge inverter of a fluorescent lamp ballast,

or a low voltage halogen lamp transformer.Two

270mA buffers drive the inverter MOSFETs in

complementary fashion with a 1.25µs built-in

dead time to prevent cross conduction. The buffer

for the upper Mosfet is driven through a 600V

level shifter realized in BCD off line technology.

The oscillator, similar to a CMOS 555 timer, oper-

ates from 25 to 150 kHz with a +/-5% maximum

tolerance. The internal 15V shunt regulator has a

9V Under Voltage Lock Out with an 1V hysteresis,

The applications

The primary application for the L6569 is the Com-

pact Fluorescent Lamp. With the oscillator, the

supply and the Mosfet drivers it is the core of the

application, and designers can customize the cir-

cuit to their requirements.

Figure 3: Block diagram of the L6569.

VS

BOOT

CHARGE

PUMP

UVLO

LEVEL

SHIFTER

HVG

HIGH

SIDE

DRIVER

RF

CF

OUT

LOGIC CONTROL

with DEAD TIME

LVG

LOW

SIDE

DRIVER

GND

Figure 4: Basic application diagram using the L6569 and two STD4NK50Z MOSFETs.

100nF

180KΩ

10µF

22Ω

STD4NK50Z

L6569

10KΩ

LAMP

10µF

AC LINE

22Ω

1nF

D02IN1385

2/14

AN880 APPLICATION NOTE

Figure 5: Bootstrap capacitor charge.

ON

15.6 V

600V

120

Ω

CHARGE PUMP CIRCUIT

ON

LOGIC

L6569

Figure 6: Basic diagram for 2x105 W lamp ballast in full bridge configuration.

HV

100nF

47

100nF

BOOT

V_S

RF

CF

V_S

RF

47

HVG

OUT

HVG

OUT

L6569

CF

EXTERNAL

OSCILLATOR

47

LVG

GND

STB9NK50Z

D02IN1386

Typical industrial TL ballasts requires complex

control with dimming or automation interface.

Here the L6569 is a driver between the power

and control blocks. To use it with an external os-

cillator, pin CF is used as an 0-12V logic input,

and the L6569 becomes a high voltage buffer.

Applications with power above 150W require a full

bridge inverter. Figure 6 shows how two L6569

drive such a MOSFET bridge. If no external con-

trol is required, the first L6569 master can control

the switching with its oscillator, and synchronizes

the other driver as (slave).

The L6569 start up

Two versions of the L6569 are available with dif-

ferent start up characteristics. The L6569 drives

the lower MOSFET ON at power-up until the sup-

ply voltage reaches the Under Voltage Lock Out.

The bootstrap capacitor is precharged to 4.6V

and both the lower and the upper MOSFETs will

switch immediately with the oscillator. This is in-

tended for inverters which use only one DC block-

ing capacitor connected to the power ground, as

shown on figure 4 for CFL ballast.

3/14

AN880 APPLICATION NOTE

The L6569A holds both MOSFETs OFF until the

Under Voltage Lock Out is reached. This is in-

tended for inverters using 2 decoupling capacitors

in half bridge as shown on figure 12. The inverter

is totally off, so that the voltage at the capacitors

center node is not unbalanced by the leakage

path during power on.

Figure 7: Current and voltage of the STP8NA50

MOSFET at turn off with the L6569.

T_GD= 245 ns ,Tc = 95 ns, E = 93 µJ

@ Tj = 50°C, RG = 22 Ω.

GD

T

Tc

D

I

CONSIDERATIONS ON THE L6569 ENVIRON-

MENT

To illustrate the benefits of the L6569 in the CFL

applications, a demonstration board was devel-

oped to supply Sylvania 18W DULUX lamp (ref:

CF18DT/E). The following chapters summarize

the application considerations applied in this de-

sign. The schematic, lay out and components list

are shown in appendix A.

V

GS

D

GND

V

50 ns/dv ; 1 A/dv ; 5 V/dv ; 50V/dv

The built-in dead time circuit acts when a MOS-

FET turns off, delaying the turn on of the opposite

transistor for 1.25 µs. The voltage V_OUTbetween

the 2 MOSFETs must switch within the minimum

dead time (0.85 µs), as shown on figure 8, to

avoid bridge cross conductions and transistors

overheat.

Symmetric half bridge operation

To supply a fluorescent lamp, the ballast has to

achieve 3 functions: pre heat, ignition, and normal

lamp operation. The serial resonance occurs be-

tween the choke and the capacitor in parallel with

the lamp. The choice of these components deter-

mines the lamp ignition voltage and the nominal

lamp current.

Figure 8: STD3NA50 MOSFET turn off when

driven by the L6569. T_C+ T_GD< T_D

Since the inverter using the L6569 and MOSFETs

can operate at a higher frequency than conven-

tional solutions, the size of the passive compo-

nents will be reduced. Such inverter can operate

up to 150 kHz in ZVS mode, and the switching

losses of the power transistors only limits the fre-

quency. In new design this frequency should be

set between 50 and 100 kHz. For instance with

an 18W lamp, a frequency increase from 33 to 50

kHz will lead to a 40% reduction of the choke

size.

T^D

V^DS

I^D

GND

T^C

LVG

GND

To operate in Zero Voltage Switching (ZVS), the

switching frequency is higher than the resonant

frequency. All operation phases of the ballast are

secure in this mode. When the bootstrap transis-

tor is conducting, no pulse current will flow from

pin BOOT to pin V_S, as it might happen in Zero

Current Switching. The bootstrap transistor re-

mains in its Safe Operating Area, and its dissipa-

tion is negligible.

T^GD

RF

GND

200 ns/dv ; 50 V/dv ; 0.1 A/dv

The MOSFET voltage selection

Since the ballast is connected to the ac mains, it

must handle any spurious voltage spikes. When

the front end RFI filter and the clamping device,

such as a varistor, absorbes totally the spike en-

ergy, MOSFETs can have the same 600V mini-

mum breakdown voltage BV_DSSas the L6569.

The MOSFET drive

The ZVS drive technique requires only a fast turn

off capability as shown on figure 2, and the tran-

sistor buffers are designed with a stronger sink

current. The two MOSFET buffers of the L6569

can sink a 400 mA peak current on capacitive

load. Typically these buffers can drive any MOS-

FETs in TO220 package.

Otherwise when the upper MOSFET is on, the re-

sidual default may be applied to the L6569. Al-

though the pin OUT breakdown voltage is higher

than 600V, it has a poor avalanche robustness.

Therefore the lower MOSFET protects the driver

by having a lower BV_DSS. A MOSFET with a mini-

mum BV_DSSup to 500V will achieve safely this

task.

Figure 7 shows an example with the STP8NA50

that has an 0.85 Ω resistance R_DS-ON

.

4/14

AN880 APPLICATION NOTE

Figure 9. L6569 driver protection against voltage spikes.

H.V.+

BV ^OUT> 600V

ON

15V

V_OUT

I

L6569

OFF

mA, a secondary winding on the resonant choke

is an easy supply alternative.

The auxiliary supply of the converter

The circuit consumption is defined by the MOS-

FETs gate charge, the I.C. consumption, the os-

cillator, and the shunt regulator. Several circuits

are possible.

In many applications a snubber is used to reduce

the dissipation in the MOSFETs. When this snub-

ber is used in conjunction with a start up resistor

(R_Sin Figure 10), a non dissipative supply is

achieved almost for free.

At start up the I.C. is consuming 150 µA, and there-

fore only a small supply resistor is required. During

operation the capacitor provides the supply current.

To avoid cross conduction, the capacitance is lim-

ited by the driver dead time T_D. Hence the capaci-

tive supply current I_Cis also limited.For a CFL bal-

last this circuit easily supplies the required

operating current. Using a CF18DT lamp ( I_L> 230

mA) the required capacitance is 470 pF on 230 Vac

line. At 50 kHz the average capacitive current is 6

mA, as described in appendix B.

The ballast shutdown

The L6569 allows several ways (see figg. 11, 12

and 13) to shutdown the ballast [4]: by acting on

the C_Finput oscillator pin to turn off the upper

MOSFET or by acting on the V_Ssupply pin with

the Under Voltage Lock Out.

Acting on C_F(Fig. 11) a limiting resistor R_Lhas to

be used, and it has to be: R_LC_F> 1µs.

When the shutdown is realized acting on Vs pin,

(see fig. 12) a limiting resistor Rs must be used to

slow down the discharge of the supply filter Cs.

The constant time of the discharge must be

greater than 10 periods of the switching fre-

quency:

10

C_sf_sw

R_S≥

Connecting the C_Fpin to ground GND stops the

oscillator, and the lower MOSFET will remain ON.

Therefore the bootstrap capacitor remains

When the required driver current is higher than 10

Figure 10: Non dissipative auxiliary supply using the transistor snubber.

1mA WHEN STARTING

6 mA WHEN 50 kHz SWITCHING

220kΩ

Rs

C

470 pF

310 V

bootstrap

circuit

Cs

L6569

5/14

AN880 APPLICATION NOTE

charged and the circuit can restart immediately.

This method is suitable when the inverter uses

only one DC blocking capacitor connected to the

power ground, as used on figure 11 for Compact

Fluorescent Lamp. Pulling the V_Svoltage below

the UVLO turns off the oscillator and gives the

same bridge configuration.

For the L6569A, discharging the V_Ssupply below

the UVLO turns off both MOSFETs. An SCR like

the X0202MA may be used for the reset function.

If the current flowing through the supply resistor is

higher than the SCR holding current (see figure

12), the SCR will remain on and the two MOS-

FETs off. Removing power or commutating the

SCR allows a new start up [4].

Otherwise a disable circuitry that turns off the two

MOSFETs (see figure13), can achieve the shut-

down function. Compared to the SCR solution,

the shutdown is immediate and the inverter can

restart on the disable order.

Figure 11: L6569 shutdown through the C_Foscillator pin.

L 6569

R_F

R_L

C_F

ON

Figure 12: Shutdown with a thyristor & a serial resistor to slow down the supply voltage decay.

L 6569A

R

Rs

OFF

C

R

shutdown

Figure 13: L6569 disable circuitry with both MOSFETs off.

H.V.

100nF

Vs

VS

BOOT

HVG

OUT

LVG

RF

5.6k

200

CF

GND

22

L6569

V_IN

W

4.7 k

HCF4011

BC327

DISABLE

6/14

AN880 APPLICATION NOTE

THE LAMP SEEN BY THE ELECTRONIC DE-

SIGNER

The preheat

Preheat techniques are used in CFL ballasts to

reduce the ignition lamp voltage. During this

phase the lamp is characterized by a high imped-

ance that forces the electrical conduction through

the preheat filaments. These filaments initially

have a low resistance that will increase by 5 times

during the preheat. The preheat typically lasts

from 400 ms to 1 s, and is achieved by controlling

either the current or the voltage of the filaments.

The lamp equivalent impedance

The compact fluorescent lamps are specified at

25 kHz (IEC 929). The MOSFETs and the L6569

allow to increase the switching frequency, but the

sensitivity of the lamp to the frequency needs to

be analyzed.

A few samples of the CF18DT/E lamp were

tested by varying the frequency and the current of

the lamp. The figure 14 shows the lamp imped-

ance versus its current as it varies from 0.1A to

0.23A with 5 frequencies from 25 to 150 kHz

(T_AMB= 25°C).

For a current control the filaments are in series

with the resonant network as shown on figure

16a. When the inverter frequency is constant, a

positive temperature coefficient thermistor (PTC)

in parallel with the lamp achieves the task by ad-

justing both the filament current and the preheat

duration. The board uses a 150Ω PTC with two

8.2 nF capacitors. The preheat lasts 0.8s and the

filament current is 0.45 Arms. The PTC is a cheap

device, but it is dissipative and works only once at

power-up.

Figure 14: Variation of the lamp impedance

versus its current for several

switching frequencies.

R lamp (Ohms)

1200

25 kHz

50 kHz

100 kHz

150 kHz

1000

800

600

400

Figure 16: Basic preheat current control diagram

(a); preheat filament energy curve (b)

200

0

0.05

0.1

0.15

0.2

0.25

0.3

I lamp (A)

From the tests the impedance appears insensitive

to the frequency for such lamps. The specified im-

pedance might be valid for higher frequency op-

eration. The relative lamp light output was meas-

ured as proposed in reference [5]. The light flux

increases slightly in that frequency range, but can

be considered constant.

I ^CTL

(A)

LAMP

Obviously the impedance is sensitive to the cur-

rent with a negative coefficient, and the ballast

operates with a non linear impedance [6]. When

current is half the nominal one, the impedance is

2.6 times higher, and the voltage is only 25%

higher (see figure 15).

E

If

E=R.I²

I ^CTL

Figure 15: Variation of the average impedance

(B)

and voltage of the lamp

R (Ohms)

U (V)

1200

200

t

Rlamp

Vlamp

900

600

300

150

100

50

The preheat can be achieved with a filament volt-

age control. The filaments are supplied by two

auxiliary windings of the resonant choke as

shown figure 17a. During the preheat the L6569

frequency is increased, and the choke operates

0

0.05

0.1

0.15

I (A)

0.2

0.25

7/14

AN880 APPLICATION NOTE

as a transformer supplying the voltage to the fila-

ments. Only few components are added around

the L6569 (see figure 18), and the control of the

preheat energy is less sensitive to the preheat du-

ration and the inverter frequency (see figure 17b).

The start up initialization

The initial conditions of the power switching start

up requires care; especially if the resonant and

switching frequencies are close to each other.

The resonant network is not loaded and the full

Figure 17: Basic preheat voltage control diagram (a); preheat filament energy curve (b)

(A)

V^CTL

LAMP

E

Vf

V^CTL

(B)

E=V²/R

t

Figure 18: Double frequency control for the L6569 with programmed frequency and duration.

1_Vs

2_RF

R^F

3_CF

R

C_F

L6569

C

C_{F_ST}

8/14

AN880 APPLICATION NOTE

line voltage V_DCis applied when the oscillator

starts. The ballast has to start directly with its

nominal conditions to remove any transient oscil-

lation. Hence the operation runs in ZVS mode

with no spurious lamp ignition. This situation does

not occur with the saturable transformer drive, be-

cause the saturation limits naturally the current by

increasing the frequency.

In the example the resonant capacitors are pre-

set to be compatible with the choke current rise

(see figure 19). The blocking capacitor is pre-

charged to approximately half V_DCby 2 biasing

resistors, and the lower Mosfet also discharges

the resonant capacitor to ground (see figure 20).

Therefore the blocking capacitor never goes

above 2/3 of the line voltage V_DC(250V rating),

the operation is safe in ZVS mode. The L6569 is

here preferred to the L6569A, because the lower

The lamp removal protection

Used in TL ballast, the lamp removal protection is

frequently also requested in the "plug-in" CFL bal-

last . Depending of the topology and the preheat

mode, the lamp removal behaves as:

- a noload resonant mode when the choke and

the capacitor are still connected to the in-

verter ; a required overcurrent protection in-

creases the frequency to reduce the current;

- an open circuit mode when the lamp filaments

are inserted in the resonant circuit.

When the circuit is open, the choke is not sup-

plied. The MOSFETs turn off slowly generating

bridge cross conduction, and undesirable dissipa-

tion losses (see figure 21). The detection stops

the switching to eliminate the cross conduction.

Figure 21: Drain current and voltage STP8NA50

Figure 19: Waveforms of the choke current and

the capacitor voltages in steady state

preheat.

MOSFET operating with noload.

I_D= 2 A peak

IDEAL INITIAL TIME

V_D

I ^I

V_GS

GND

I_D

GND

V^B

V^BI

5 s/dv ; 50 V/dv ; 0.5 A/dv

µ

100ns/dv ; 50 V/dv ; 5V/dv ; 1 A/dv

Mosfet is on at power-up.

Figure 20: Configuration of the resonant network during the initialization of the driver.

VS<UVLO

2.4 mH

I ^I

V^BI

ON

L6569

V^B

4nF

100 nF

Ω

2 x 180 k

9/14

AN880 APPLICATION NOTE

Figure 22: Open load detection example.

L6569

LAMP

3_CF

4_GND

10.R

18V

100.R

10.R

11.R

duced, and the ballast becomes cheaper.With its

supply and its oscillator the L6569 is versatile,

and its flexibility permits to design any improved

power control.

Several ways can achieve the protection task.

First it can be done by sensing the resonant cur-

rent through a MOSFET source resistor or a sec-

ondary winding on the choke. The switching is

stopped when a large current reduction is de-

tected by analog means.

BIBLIOGRAPHY

A logic circuit can also detect the presence of the

lamp filaments. One end of a filament is always

connected to a fixed voltage. If the other end of

the filament is connected through a high imped-

ance resistor to another voltage, the absence of

the filament can be easily detected by monitoring

the resistor voltage change as shown on figure

22.

[1]: AN 527 "Electronic fluorescent lamp ballast"

A.Vitanza, R.Scollo, SGS-THOMSON

[2]: Smart Power ICs, Chapter 8, High voltage in-

tegrated circuits for off-line power applications.

C.Diazzi, SGS-THOMSON

[3]: AN 512 "Characteristics of power semicon-

ductors" JM Peter, SGS-THOMSON

[4] : "The L6569 half bridge driver: the shutdown

function"

CONCLUSION

[5] : "Compatibility test of dimming electronic bal-

lasts used in daylighting and environment con-

trols", A.Buddenberg, Rensselaer Polytechnic In-

stitute

[6]: "PSpice High Frequency Dynamic Fluores-

cent Lamp Model", Bryce Hesterman, APEC’96,

p.641

The foregoing note shows how high voltage driv-

ers, like the L6569, simplify the design of the

lamp ballast. These devices includes all the cir-

cuitry to drive MOSFETs in half bridge inverter.

Since the optimized switching frequency in-

creases above 50 kHz with a low tolerance, the

size of the passive resonant components is re-

10/14

AN880 APPLICATION NOTE

APPENDIX A: CFL DEMONSTRATION BOARD

WITH THE L6569

The resonant ballast

The value of the choke (L₁) and the two capaci-

tors (C₇& C₈) in parallel with the lamp determine

the lamp ignition voltage and the nominal lamp

current. During the ignition the lamp impedance is

essentially infinite, and the filaments resistance is

only the serial load. To generate the ignition volt-

age, the switching frequency is set close to the

resonance frequency. In normal operation the

choke resonates with the capacitors C₇& C₈

(parallel loading), but also with the decoupling ca-

pacitor C₆(serial loading). The current mode pre-

heat uses a 150Ω Positive Temperature Coeffi-

cient thermistor. Inserted in the capacitive series

(C₇& C₈), the PTC produces a 0.45 Arms fila-

ment current during the initial 0.8 s (reference:

307C1253BHEAB from CERA-MITE).

A demonstration board was developed as an ex-

ample for Compact Fluorescent Lamp ballast. It is

optimised for a CF18DT/E/830 18W lamp from

Osram-Sylvania. Using the L6569 the circuit

achieves preheat, ignition and normal lamp op-

eration. The power transistors are two STD3NA50

500V-3Ω MOSFETs in I-PACK package.

Board description

The three sections of the board are an AC input

rectifier, the half bridge inverter, and the resonant

ballast. By changing the connection on the input

mains, the ballast can operate either on 120 Vac

mains with a voltage doubler rectification, or on

230 Vac mains with a full wave rectification. The

input resistor R₁limits the initial inrush current

charging the bulk capacitors. The L6569 operates

with a single 50 kHz switching frequency pro-

grammed by R₄and C₁. Two fast diodes D₂& D₃

synchronize the oscillator to keep the switching in

ZVS mode. The control circuit requires 4.5 mA to

supply the I.C., the MOS gate drives, and the os-

cillator. Its supply delivers at least 6.5 mA as de-

scribed in appendix B. The start up resistor also

balances the voltage across the two bulk capaci-

tors.

Basic ballast electrical characteristics

Input voltage: 120 or 230 Vac by input connec-

tions change

Switching frequency: 50kHz

Average dc line voltage range: Vdc from 260 to 355

Nominal supply current: 0.17A rms @ 310 Vdc

Nominal output power: 17W

Minimum ignition voltage: 700V peak @ 260 Vdc

Nominal preheat current: 0.45 Arms during 0.8s

@ 310 Vdc

Figure 23. CFL ballast diagram for a 18W

CFD18T/E lamp with 120/230 Vac inputs.

Figure 23.

Q1

STD4NK50

R9

180K

1/4W

C4100nF 50V

R8

120K

1/2W

V_S

BOOT

HVG

OUT

C2

47µF

250V

RF

R10 10K

1/4W

R4

27K

R2 22 1/4W

R3 22 1/4W

L1=2.4mH

4 x 1N4006

D7

L6569

1/4W

D4

CF

R5 100K

1/2W

LVG

Q2

STD4NK50

C7

8.2nF

630V

(*)

220V

C1

560pF

50V

GND

D6

D5

D8

1N4148

N

C5

47µF

250V

C3

4.7µF

25V

C9 470pF 630V

(*)

R7

180K

1/4W

C6

100nF

250V

C8

8.2nF

630V

(*)

R1 15 1W

RV1

PTC 150

350V

R6 47 1/4W

110V

D1

BYW100-100

ZPD 18V

D2

CFL LAMP

SYLVANIA DELUX T/E 18W

L1=2.4mH core TH LCC E2006-B4 Ref also VOGH 575 0409200 2.4mH

C7-C8=PS8n2J H3 630-2A TH

D96IN419C

D3 BYW100-100

(*) Polypropylene, Capacitors 630V rated V_L

11/14

AN880 APPLICATION NOTE

Figure 24: PCB Layout of the board.

Comp.

Side

Copper

Side

Figure 25: PCB component placement diagram.

The resonant choke

Customization of the board

The inductance of the choke is 2.4 mH with a

minimum saturation current of 0.65 A. In the prac-

tical example it has been done with:

Core: Thomson LCC E2006 material B4;

Air gap: 2 spacers of 0.4 mm each (total 0.8 mm);

AL = 75 nH;

Some flexibility is added to the board to extend its

evaluation. The MOSFETs have two foot prints to

mount either I-PAK or TO220 packages. And two

choke footprints are also avalaible for E1905A

and E2006A magnetic cores.

Bobbin: HC2006BA-06;

Number of turns: 175;

Measured saturation current: 1 A peak @ 25 °C;

12/14

AN880 APPLICATION NOTE

plies the lamp current during the lower MOS turn

off. To avoid any cross conduction its capacitance

is limited by the driver dead time T_D(see figure

26). Hence the capacitive supply current I_Cis also

limited.

APPENDIX B: Rating of the capacitive supply

with the L6569 driver

The supply is made with the snubber and a start

up resistor R_S.

T

D

I

L

A snubber circuit is used to minimize the MOS-

FETs dissipation. It also achieves a non dissipa-

tive supply as shown on figure 10.

C <

V_DC

I_CAV= C V_DCF_SW< I_LT_DF_SW

The MOSFETs gate charge, the driver consump-

tion, the oscillator, and the shunt regulator, define

the circuit consumption. We can estimate this

Where I_Lis the peak lamp current, and F_SWthe

switching frequency.

current is I_{S AV}

I_SAV> 2 I_G+ I_QS+ I_OSC+ I_REG

V_S

:

=

For a ballast such as a CFL one this circuit sup-

plies easily the required current. For instance with

a CF18DT lamp ( I_L> 230 mA) the capacitor is

1nF on 120Vac line, 470 pF on 230 Vac line. At

50 kHz the average capacitive current is 6 mA in

both cases.

= 2 Q_Gfsw + I_QS

+

+ I_REG

R_F

2

Where Q_Gthe MOSFET gate charge

I_QSthe driver supply current

V_Sthe supply voltage

R_Fthe oscillator resistor and V_Sthe driver

supply voltage

Figure 26: Cross conduction of the snubber

capacitor with the upper MOSFET:

capacitor current and voltage

waveforms.

I_REGthe shunt regulator current.

When V is lower than the UVLO threshold U_UVLO, the

S

driver is only consuming. Its current must be mini-

mal to reduce the dissipation of the resistor R_S.

The L6569 has a 150 µA start up current, and the

maximum resistance is 2MΩ for a 230Vac line ap-

plication.

T^D

V

^HVG+ V^OUT

We can also reduce the resistor value to get a

faster start up time T_S.

GND

IC

GND

RF

R_SC_SU_UVLO

T_S=

V_DC

200 ns/dv ; 50 V/dv ; 0.1 A/dv

Where C_Sis the supply capacitor, and V_DCthe

line voltage.

When the timer oscillates, the capacitor C sup-

13/14

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granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specification mentioned in this publication are

subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products

are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics.

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14/14


型号：	EVAL6569
厂家：	PANASONIC
描述：	THE L6569 A NEW HIGH VOLTAGE IC DRIVER FOR ELECTRONIC LAMP BALLAST THE L6569新的高压IC驱动灯的电子镇流器高压电子驱动
文件：	总14页 (文件大小：192K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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