LM2596TADJG [ONSEMI]
3.0 A, Step-Down Switching Regulator; 3.0 A，降压型开关稳压器
| 型号： | LM2596TADJG |
| 厂家： | 
ONSEMI |
| 描述： | 3.0 A, Step-Down Switching Regulator |
| 文件： | 总25页 (文件大小：449K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LM2596
3.0 A, Step-Down Switching
Regulator
The LM2596 regulator is monolithic integrated circuit ideally suited
for easy and convenient design of a step−down switching regulator
(buck converter). It is capable of driving a 3.0 A load with excellent
line and load regulation. This device is available in adjustable output
version and it is internally compensated to minimize the number of
external components to simplify the power supply design.
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Since LM2596 converter is a switch−mode power supply, its
efficiency is significantly higher in comparison with popular
three−terminal linear regulators, especially with higher input voltages.
The LM2596 operates at a switching frequency of 150 kHz thus
allowing smaller sized filter components than what would be needed
with lower frequency switching regulators. Available in a standard
5−lead TO−220 package with several different lead bend options, and
TO−220
TV SUFFIX
CASE 314B
1
5
Heatsink surface connected to Pin 3
2
D PAK surface mount package.
The other features include a guaranteed $4% tolerance on output
voltage within specified input voltages and output load conditions, and
$15% on the oscillator frequency. External shutdown is included,
featuring 80 mA (typical) standby current. Self protection features
include switch cycle−by−cycle current limit for the output switch, as
well as thermal shutdown for complete protection under fault
conditions.
TO−220
T SUFFIX
CASE 314D
1
5
Pin
1. V
in
2. Output
Features
• Adjustable Output Voltage Range 1.23 V − 37 V
• Guaranteed 3.0 A Output Load Current
• Wide Input Voltage Range up to 40 V
• 150 kHz Fixed Frequency Internal Oscillator
• TTL Shutdown Capability
• Low Power Standby Mode, typ 80 mA
• Thermal Shutdown and Current Limit Protection
• Internal Loop Compensation
• Moisture Sensitivity Level (MSL) Equals 1
• Pb−Free Packages are Available
3. Ground
4. Feedback
5. ON/OFF
2
D PAK
D2T SUFFIX
CASE 936A
1
5
Heatsink surface (shown as terminal 6 in
case outline drawing) is connected to Pin 3
Applications
ORDERING INFORMATION
See detailed ordering and shipping information in the package
dimensions section on page 23 of this data sheet.
• Simple High−Efficiency Step−Down (Buck) Regulator
• Efficient Pre−Regulator for Linear Regulators
• On−Card Switching Regulators
• Positive to Negative Converter (Buck−Boost)
• Negative Step−Up Converters
• Power Supply for Battery Chargers
DEVICE MARKING INFORMATION
See general marking information in the device marking
section on page 23 of this data sheet.
© Semiconductor Components Industries, LLC, 2008
1
Publication Order Number:
November, 2008 − Rev. 0
LM2596/D
LM2596
Typical Application (Adjustable Output Voltage Version)
R1
1.0k
Feedback
4
L1
33 mH
12 V
Unregulated
DC Input
+V
R2
3.1k
in
LM2596
C
FF
Output
1
5.0 V Regulated
Output 3.0 A Load
C
in
100 mF
D1
1N5822
2
GND 5 ON/OFF
C
out
220 mF
3
Block Diagram
+V
in
ON/OFF
Unregulated
DC Input
3.1 V Internal
Regulator
ON/OFF
1
5
C
in
Feedback
Current
Limit
R2
4
Fixed Gain
Error Amplifier
C
FF
Comparator
Driver
Regulated
Output
R1
Latch
Freq
Shift
L1
V
out
Output
30 kHz
3.0 Amp
Switch
2
GND
1.235 V
Band-Gap
Reference
C
out
D1
Thermal
Shutdown
150 kHz
Oscillator
3
Reset
Load
Figure 1. Typical Application and Internal Block Diagram
MAXIMUM RATINGS
Rating
Symbol
Value
45
Unit
Maximum Supply Voltage
ON/OFF Pin Input Voltage
V
in
V
V
V
−
−0.3 V ≤ V ≤ +V
−1.0
in
Output Voltage to Ground (Steady−State)
Power Dissipation
−
Case 314B and 314D (TO−220, 5−Lead)
Thermal Resistance, Junction−to−Ambient
Thermal Resistance, Junction−to−Case
P
Internally Limited
W
°C/W
°C/W
W
D
R
R
65
q
JA
JC
D
5.0
q
2
Case 936A (D PAK)
P
Internally Limited
Thermal Resistance, Junction−to−Ambient
Thermal Resistance, Junction−to−Case
Storage Temperature Range
R
70
5.0
°C/W
°C/W
°C
q
JA
JC
R
q
T
−65 to +150
2.0
stg
Minimum ESD Rating (Human Body Model: C = 100 pF, R = 1.5 kW)
Lead Temperature (Soldering, 10 seconds)
Maximum Junction Temperature
−
kV
−
260
°C
T
150
°C
J
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
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LM2596
PIN FUNCTION DESCRIPTION
Pin
Symbol
Description (Refer to Figure 1)
1
V
in
This pin is the positive input supply for the LM2596 step−down switching regulator. In order to minimize voltage transi-
ents and to supply the switching currents needed by the regulator, a suitable input bypass capacitor must be present
(C in Figure 1).
in
2
Output
This is the emitter of the internal switch. The saturation voltage V of this output switch is typically 1.5 V. It should be
kept in mind that the PCB area connected to this pin should be kept to a minimum in order to minimize coupling to
sensitive circuitry.
sat
3
4
GND
Circuit ground pin. See the information about the printed circuit board layout.
Feedback This pin is the direct input of the error amplifier and the resistor network R2, R1 is connected externally to allow pro-
gramming of the output voltage.
5
ON/OFF
It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the total input supply
current to approximately 80 mA. The threshold voltage is typically 1.6 V. Applying a voltage above this value (up to
+V ) shuts the regulator off. If the voltage applied to this pin is lower than 1.6 V or if this pin is left open, the regulator
in
will be in the “on” condition.
OPERATING RATINGS (Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee
specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics.)
Rating
Operating Junction Temperature Range
Supply Voltage
Symbol
Value
Unit
°C
T
J
−40 to +125
4.5 to 40
V
in
V
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LM2596
SYSTEM PARAMETERS
ELECTRICAL CHARACTERISTICS Specifications with standard type face are for T = 25°C, and those with boldface type apply
J
over full Operating Temperature Range −40°C to +125°C
Characteristics
LM2596 (Note 1, Test Circuit Figure 15)
Feedback Voltage (V = 12 V, I = 0.5 A, V = 5.0 V, )
Symbol
Min
Typ
Max
Unit
V
FB_nom
1.23
V
V
in
Load
out
Feedback Voltage (8.5 V ≤ V ≤ 40 V, 0.5 A ≤ I
≤ 3.0 A, V = 5.0 V)
V
FB
1.193
1.18
1.267
1.28
in
Load
out
Efficiency (V = 12 V, I
= 3.0 A, V = 5.0 V)
η
−
73
Typ
25
−
%
Unit
nA
in
Load
out
Characteristics
Symbol
Min
Max
Feedback Bias Current (V = 5.0 V)
I
b
100
200
out
Oscillator Frequency (Note 2)
f
135
120
150
1.5
165
kHz
V
osc
180
Saturation Voltage (I = 3.0 A, Notes 3 and 4)
V
1.8
2.0
out
sat
Max Duty Cycle “ON” (Note 4)
DC
95
%
A
Current Limit (Peak Current, Notes 2 and 3)
I
CL
4.2
3.5
5.6
6.9
7.5
Output Leakage Current (Notes 5 and 6)
Output = 0 V
Output = −1.0 V
I
L
mA
0.5
6.0
2.0
20
Quiescent Current (Note 5)
I
5.0
80
10
mA
Q
Standby Quiescent Current (ON/OFF Pin = 5.0 V (“OFF”))
(Note 6)
I
200
250
mA
stby
ON/OFF PIN LOGIC INPUT
Threshold Voltage
1.6
V
V
V
out
= 0 V (Regulator OFF)
V
IH
2.2
2.4
V
out
= Nominal Output Voltage (Regulator ON)
V
IL
1.0
0.8
V
ON/OFF Pin Input Current
ON/OFF Pin = 5.0 V (Regulator OFF)
ON/OFF Pin = 0 V (regulator ON)
I
−
−
15
30
mA
mA
IH
I
0.01
5.0
IL
1. External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance.
When the LM2596 is used as shown in the Figure 15 test circuit, system performance will be as shown in system parameters section.
2. The oscillator frequency reduces to approximately 30 kHz in the event of an output short or an overload which causes the regulated output
voltage to drop approximately 40% from the nominal output voltage. This self protection feature lowers the average dissipation of the IC by
lowering the minimum duty cycle from 5% down to approximately 2%.
3. No diode, inductor or capacitor connected to output (Pin 2) sourcing the current.
4. Feedback (Pin 4) removed from output and connected to 0 V.
5. Feedback (Pin 4) removed from output and connected to +12 V to force the output transistor “off”.
6. V = 40 V.
in
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LM2596
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 15)
1.0
0.8
0.6
1.4
1.2
V = 20 V
in
I
= 500 mA
Load
I
= 500 mA
Load
T = 25°C
J
1.0
0.8
0.6
0.4
Normalized at T = 25°C
J
0.4
0.2
3.3 V and 5.0 V
0
-0.2
-0.4
0.2
0
12 V and 15 V
-0.2
-0.4
-0.6
-0.6
-0.8
-1.0
-50
-25
0
25
50
75
100
125
0
5.0
10
15
20
25
30
35
40
T , JUNCTION TEMPERATURE (°C)
J
V , INPUT VOLTAGE (V)
in
Figure 2. Normalized Output Voltage
Figure 3. Line Regulation
6.0
5.5
2.0
1.5
1.0
0.5
0
V = 25 V
in
I
= 3.0 A
Load
5.0
I
= 500 mA
Load
4.5
4.0
L1 = 33 mH
= 0.1 W
R
ind
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
T , JUNCTION TEMPERATURE (°C)
J
T , JUNCTION TEMPERATURE (°C)
J
Figure 4. Dropout Voltage
Figure 5. Current Limit
20
18
200
V
= 5.0 V
V
= 5.0 V
out
180
160
140
120
100
80
ON/OFF
Measured at
Ground Pin
T = 25°C
J
16
14
V = 40 V
in
I
= 3.0 A
Load
12
10
V = 12 V
in
60
I
= 200 mA
Load
8.0
6.0
4.0
40
20
0
0
5.0
10
15
20
25
30
35
40
-50
-25
0
25
50
75
100
125
V , INPUT VOLTAGE (V)
in
T , JUNCTION TEMPERATURE (°C)
J
Figure 6. Quiescent Current
Figure 7. Standby Quiescent Current
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LM2596
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 15)
200
1.6
180
160
140
120
100
80
1.4
1.2
T = 25°C
J
-40°C
1.0
0.8 25°C
0.6
125°C
60
40
0.4
0.2
20
0
0
0
0
5
10
15
20
25
30
35
40
0.5
1.0
1.5
2.0
2.5
3.0
V , INPUT VOLTAGE (V)
in
SWITCH CURRENT (A)
Figure 8. Standby Quiescent Current
Figure 9. Switch Saturation Voltage
1.0
0.0
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
V
= 12 V Normalized
IN
at 25°C
−1.0
−2.0
−3.0
−4.0
−5.0
−6.0
−7.0
−8.0
−9.0
V
' 1.23 V
= 500 mA
out
I
Load
0
-25
0
25
50
75
100
125
−50
−25
0
25
50
75
100
125
-50
T , JUNCTION TEMPERATURE (°C)
J
T , JUNCTION TEMPERATURE (°C)
J
Figure 11. Minimum Supply Operating Voltage
Figure 10. Switching Frequency
100
80
60
40
20
0
-20
-40
-60
-80
-100
-50
-25
0
25
50
75
100
125
T , JUNCTION TEMPERATURE (°C)
J
Figure 12. Feedback Pin Current
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LM2596
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 15)
10 V
0
A
B
100 mV
Output
0
4.0 A
Voltage
Change
2.0 A
0
- 100 mV
3.0 A
4.0 A
2.0 A
0
C
D
Load
2.0 A
Current
1.0 A
0
2 ms/div
100 ms/div
Figure 13. Switching Waveforms
Figure 14. Load Transient Response
Vout = 5 V
A: Output Pin Voltage, 10 V/div
B: Switch Current, 2.0 A/div
C: Inductor Current, 2.0 A/div, AC−Coupled
D: Output Ripple Voltage, 50 mV/div, AC−Coupled
Horizontal Time Base: 5.0 ms/div
Adjustable Output Voltage Versions
Feedback
4
V
in
LM2596
L1
33 mH
V
out
5,000 V
1
Output
2
ON/OFF
C
FF
3
GND
5
8.5 V - 40 V
Unregulated
DC Input
R2
R1
C
in
100 mF
C
out
220 mF
D1
1N5822
Load
R2
Ǔ
R1
ǒ1.0 )ꢀ
V
+ V
out
refꢀ
V
out
ꢀ 1.0
ref
R2 + R1
ǒ
Ǔ
Where V = 1.23 V, R1
V
ref
between 1.0 k and 5.0 k
Figure 15. Typical Test Circuit
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LM2596
PCB LAYOUT GUIDELINES
As in any switching regulator, the layout of the printed
On the other hand, the PCB area connected to the Pin 2
(emitter of the internal switch) of the LM2596 should be
kept to a minimum in order to minimize coupling to sensitive
circuitry.
Another sensitive part of the circuit is the feedback. It is
important to keep the sensitive feedback wiring short. To
assure this, physically locate the programming resistors near
to the regulator, when using the adjustable version of the
LM2596 regulator.
circuit board is very important. Rapidly switching currents
associated with wiring inductance, stray capacitance and
parasitic inductance of the printed circuit board traces can
generate voltage transients which can generate
electromagnetic interferences (EMI) and affect the desired
operation. As indicated in the Figure 15, to minimize
inductance and ground loops, the length of the leads
indicated by heavy lines should be kept as short as possible.
For best results, single−point grounding (as indicated) or
ground plane construction should be used.
DESIGN PROCEDURE
Buck Converter Basics
This period ends when the power switch is once again
turned on. Regulation of the converter is accomplished by
varying the duty cycle of the power switch. It is possible to
describe the duty cycle as follows:
The LM2596 is a “Buck” or Step−Down Converter which
is the most elementary forward−mode converter. Its basic
schematic can be seen in Figure 16.
The operation of this regulator topology has two distinct
time periods. The first one occurs when the series switch is
on, the input voltage is connected to the input of the inductor.
The output of the inductor is the output voltage, and the
rectifier (or catch diode) is reverse biased. During this
period, since there is a constant voltage source connected
across the inductor, the inductor current begins to linearly
ramp upwards, as described by the following equation:
t
on
T
d +
, where T is the period of switching.
For the buck converter with ideal components, the duty
cycle can also be described as:
V
out
d +
V
in
Figure 17 shows the buck converter, idealized waveforms
of the catch diode voltage and the inductor current.
ǒV
Ǔt
on
IN * VOUT
IL(on)
+
V
on(SW)
L
During this “on” period, energy is stored within the core
material in the form of magnetic flux. If the inductor is
properly designed, there is sufficient energy stored to carry
the requirements of the load during the “off” period.
Power
Switch
Off
Power
Switch
On
Power
Switch
Off
Power
Switch
On
Power
Switch
L
V (FWD)
D
C
V
in
D
out
R
Load
Time
(AV)
I
pk
Figure 16. Basic Buck Converter
I
The next period is the “off” period of the power switch.
When the power switch turns off, the voltage across the
inductor reverses its polarity and is clamped at one diode
voltage drop below ground by the catch diode. The current
now flows through the catch diode thus maintaining the load
current loop. This removes the stored energy from the
inductor. The inductor current during this time is:
Load
I
min
Power
Switch
Power
Switch
Diode
Diode
Time
Figure 17. Buck Converter Idealized Waveforms
ǒV
Ǔt
off
OUT * VD
IL(off)
+
L
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LM2596
PROCEDURE (ADJUSTABLE OUTPUT VERSION: LM2596)
Procedure
Example
Given Parameters:
= Regulated Output Voltage
Given Parameters:
= 5.0 V
V
V
out
out
V
= Maximum DC Input Voltage
V
= 12 V
in(max)
in(max)
I
= Maximum Load Current
I
= 3.0 A
Load(max)
Load(max)
1. Programming Output Voltage
1. Programming Output Voltage (selecting R1 and R2)
To select the right programming resistor R1 and R2 value (see
Figure 1) use the following formula:
Select R1 and R2:
R2
R1
+ 1.23ǒ1.0 )
Ǔ
V
Select R1 = 1.0 kW
out
R2
R1
ref ǒ1.0 )
Ǔ
V
+ V
where V = 1.23 V
ref
out
V
5 V
out
+ ǒ * 1.0Ǔ
R2 + R1ǒ Ǔ
* 1.0
Resistor R1 can be between 1.0 k and 5.0 kW. (For best
temperature coefficient and stability with time, use 1% metal
V
1.23 V
ref
film resistors).
R2 = 3.0 kW, choose a 3.0k metal film resistor.
Vout
R2 + R1ǒ Ǔ
* 1.0
Vref
2. Input Capacitor Selection (C )
2. Input Capacitor Selection (C )
in
in
To prevent large voltage transients from appearing at the input
and for stable operation of the converter, an aluminium or
tantalum electrolytic bypass capacitor is needed between the
A 100 mF, 50 V aluminium electrolytic capacitor located near
the input and ground pin provides sufficient bypassing.
input pin +V and ground pin GND This capacitor should be
in
located close to the IC using short leads. This capacitor should
have a low ESR (Equivalent Series Resistance) value.
For additional information see input capacitor section in the
“Application Information” section of this data sheet.
3. Catch Diode Selection (D1)
3. Catch Diode Selection (D1)
A. Since the diode maximum peak current exceeds the
regulator maximum load current the catch diode current
rating must be at least 1.2 times greater than the maximum
load current. For a robust design, the diode should have a
current rating equal to the maximum current limit of the
LM2596 to be able to withstand a continuous output short.
B. The reverse voltage rating of the diode should be at least
1.25 times the maximum input voltage.
A. For this example, a 3.0 A current rating is adequate.
B. For robust design use a 30 V 1N5824 Schottky diode or
any suggested fast recovery diode in the Table 2.
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LM2596
PROCEDURE (ADJUSTABLE OUTPUT VERSION: LM2596) (CONTINUED)
Procedure
Example
4. Inductor Selection (L1)
4. Inductor Selection (L1)
A. Calculate E x T [V x ms] constant:
A. Use the following formula to calculate the inductor Volt x
microsecond [V x ms] constant:
5 ) 0.5
1000
ǒ
Ǔ
ǒ
Ǔ
V ms
E T + 12 * 5 * 1.5
V
) V
12 * 5 ) 0.5
150 kHz
OUT
* V
D
1000
ǒ Ǔ
V ms
E T + ǒV
Ǔ
SAT
* V
* V
IN
OUT
V
) V
150 kHz
5.5
IN
SAT
D
ǒ
Ǔ
ǒ
Ǔ
E T + 5.5
6.6 V ms
7.5
B. Match the calculated E x T value with the corresponding
number on the vertical axis of the Inductor Value Selection
Guide shown in Figure 18. This E x T constant is a
B. E x T = 27 [V x ms]
measure of the energy handling capability of an inductor and
is dependent upon the type of core, the core area, the
number of turns, and the duty cycle.
C. Next step is to identify the inductance region intersected by
the E x T value and the maximum load current value on the
horizontal axis shown in Figure 18.
C. I
= 3.0 A
Load(max)
Inductance Region = L40
D. Select an appropriate inductor from Table 3.
The inductor chosen must be rated for a switching
D. Proper inductor value = 33 mH
frequency of 150 kHz and for a current rating of 1.15 x I
The inductor current rating can also be determined by
calculating the inductor peak current:
.
Choose the inductor from Table 3.
Load
ǒVin outǓton
* V
I
)
p(max) + I
Load(max)
2L
where t is the “on” time of the power switch and
on
V
out
1.0
osc
t
+
x
on
V
f
in
5. Output Capacitor Selection (C
)
out
5. Output Capacitor Selection (C
)
out
A. Since the LM2596 is a forward−mode switching regulator
with voltage mode control, its open loop has 2−pole−1−zero
frequency characteristic. The loop stability is determined by
the output capacitor (capacitance, ESR) and inductance
values.
A. In this example is recommended Nichicon PM
capacitors: 470 mF/35 V or 220 mF/35 V
For stable operation use recommended values of the output
capacitors in Table 1.
Low ESR electrolytic capacitors between 220uFand 1500uF
provide best results.
B. The capacitors voltage rating should be at least 1.5 times
greater than the output voltage, and often much higher
voltage rating is needed to satisfy low ESR requirement
6. Feedforward Capacitor (C
)
FF
6. Feedforward Capacitor (C )
FF
It provides additional stability mainly for higher input voltages. For
Cff selection use Table 1. The compensation capacitor between
0.6 nF and 40 nF is wired in parallel with the output voltage setting
resistor R2, The capacitor type can be ceramic, plastic, etc..
In this example is recommended feedforward capacitor
15 nF or 5 nF.
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LM2596
LM2596 Series Buck Regulator Design Procedures (continued)
Table 1. RECOMMENDED VALUES OF THE OUTPUT CAPACITOR AND FEEDFORWARD CAPACITOR
(I
load
= 3 A)
Nichicon PM Capacitors
V
in
(V)
Capacity/Voltage Range/ESR (mF/V/mW)
40
1500/35/24
1200/35/26
1000/35/29
820/35/32
820/35/32
820/35/32
820/35/32
2
1000/35/29
820/35
1000/35/29
680/35/36
560/35/41
470/25/65
470/25/65
220/35/85
220/35/85
6
680/35/36
560/35/41
330/25/85
330/25/85
330/25/85
220/25/111
560/25/55
470/25/65
330/25/85
330/25/85
330/25/85
560/25/55
470/25/65
220/35/85
220/35/85
220/35/85
470/35/46
330/35/60
470/35/46
26
22
20
18
12
10
680/35/36
470/35/46
470/35/46
470/35/46
470/35/46
4
V
(V)
9
12
15
24
28
out
C
(nF]
40
15
5
2
1.5
1
0.6
0.6
FF
70
60
L42
L27
L35
L43
220uH
L36
50
40
L44
L27
L37
150uH
L29
L38
30
25
100uH
L30
L39
68uH
L31
L40
20
15
47uH
L21
L32
L40
33uH
L40
L22
22uH
10
9
L23
L34
8
L24
7
6
15uH
L25
1.5
L15
5
4
0.6
0.8
1.0
2.0
2.5
3.0
Maximum load current (A)
Figure 18. Inductor Value Selection Guides (For Continuous Mode Operation)
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11
LM2596
Table 2. DIODE SELECTION
3.0 A
Schottky
Fast Recovery
4.0 − 6.0 A
3.0 A
4.0 − 6.0 A
Through
Hole
Surface
Mount
Through Surface
Through
Hole
Surface
Mount
Through
Hole
Surface
Mount
Hole
Mount
V
R
20 V
1N5820
MBR320P
SR302
SK32
1N5823
SR502
SB520
30 V
1N5821
MBR330
SR303
SK33
30WQ03
1N5824
SR503
SB530
50WQ03
MUR320
31DF1
HER302
MURS320T3
MURD320
30WF10
MUR420
HER602
MURD620CT
50WF10
31DQ03
40 V
1N5822
MBR340
SR304
SK34
30WQ04
MBRS340T3
MBRD340
1N5825
SR504
SB540
MBRD640CT
50WQ04
(all diodes
rated
to at least
100 V)
(all diodes
rated
to at least
100 V)
(all diodes
rated
to at least
100 V)
(all diodes
rated
to at least
100 V)
31DQ04
50 V
60 V
MBR350
31DQ05
SR305
SK35
30WQ05
SB550
50WQ05
MBR360
DQ06
MBRS360T3
MBRD360
50SQ080
MBRD660CT
SR306
NOTE: Diodes listed in bold are available from ON Semiconductor.
http://onsemi.com
12
LM2596
Table 3. INDUCTOR MANUFACTURERS PART NUMBERS
Schott
Renco
Pulse Engineering
Coilcraft
Through
Hole
Surface
Mount
Through
Hole
Surface
Mount
Through
Hole
Surface
Mount
Inductance
Current
(A)
Surface Mount
(mH)
L15
L21
22
0.99
67148350
67148460
RL−1284−22−43
RL1500−2
PE−53815
PE−53815−S
DO3308−223
2
68
0.99
67144070
67144450
RL−5471−5
RL1500−6
PE−53821
PE−53821−S
DO3316−683
8
L22
L23
L24
L25
L26
L27
L28
L29
L30
L31
L32
L33
L34
L35
L36
L37
L38
L39
L40
L41
L42
L43
L44
47
33
1.17
1.40
1.70
2.10
0.80
1.00
1.20
1.47
1.78
2.20
2.50
3.10
3.40
1.70
2.10
2.50
3.10
3.50
3.50
3.50
2.70
3.40
3.40
67144080
67144090
67148370
67148380
67144100
67144110
67144120
67144130
67144140
67144150
67144160
67148390
67148400
67144170
67144180
67144190
67144200
67144210
67144220
67144230
67148410
67144240
67144250
67144460
67144470
67148480
67148490
67144480
67144490
67144500
67144510
67144520
67144530
67144540
67148500
67148790
−
RL−5471−6
RL−5471−7
RL−1283−22−43
RL−1283−15−43
RL−5471−1
RL−5471−2
RL−5471−3
RL−5471−4
RL−5471−5
RL−5471−6
RL−5471−7
RL−1283−22−43
RL−1283−15−43
RL−5473−1
RL−5473−4
RL−5472−1
RL−5472−2
RL−5472−3
RL−5472−4
RL−5472−5
RL−5473−4
RL−5473−2
RL−5473−3
−
−
−
−
−
−
−
−
−
−
−
−
−
−
−
−
−
−
−
−
−
−
−
PE−53822
PE−53823
PE−53824
PE−53825
PE−53826
PE−53827
PE−53828
PE−53829
PE−53830
PE−53831
PE−53932
PE−53933
PE−53934
PE−53935
PE−54036
PE−54037
PE−54038
PE−54039
PE−54040
PE−54041
PE−54042
PE−54043
PE−54044
PE−53822−S
PE−53823−S
PE−53825−S
PE−53824−S
PE−53826−S
PE−53827−S
PE−53828−S
PE−53829−S
PE−53830−S
PE−53831−S
PE−53932−S
PE−53933−S
PE−53934−S
PE−53935−S
PE−54036−S
PE−54037−S
PE−54038−S
PE−54039−S
PE−54040−S
PE−54041−S
PE−54042−S
DO3316−473
DO3316−333
DO3316−223
DO3316−153
DO5022P−334
DO5022P−224
DO5022P−154
DO5022P−104
DO5022P−683
DO5022P−473
DO5022P−333
DO5022P−223
DO5022P−153
−
22
15
330
220
150
100
68
47
33
22
15
220
150
100
68
−
−
−
−
−
DO5040H−683ML
DO5040H−473ML
DO5040H−333ML
DO5040H−223ML
−
47
−
33
67148290
67148300
−
22
150
100
68
−
-
−
DO5040H−683ML
http://onsemi.com
13
LM2596
APPLICATION INFORMATION
EXTERNAL COMPONENTS
regulator loop stability. The ESR of the output capacitor and
the peak−to−peak value of the inductor ripple current are the
main factors contributing to the output ripple voltage value.
Standard aluminium electrolytics could be adequate for
some applications but for quality design, low ESR types are
recommended.
An aluminium electrolytic capacitor’s ESR value is
related to many factors such as the capacitance value, the
voltage rating, the physical size and the type of construction.
In most cases, the higher voltage electrolytic capacitors have
lower ESR value. Often capacitors with much higher
voltage ratings may be needed to provide low ESR values
that, are required for low output ripple voltage.
Feedfoward Capacitor
(Adjustable Output Voltage Version)
This capacitor adds lead compensation to the feedback
loop and increases the phase margin for better loop stability.
For CFF selection, see the design procedure section.
Input Capacitor (Cin)
The Input Capacitor Should Have a Low ESR
For stable operation of the switch mode converter a low
ESR (Equivalent Series Resistance) aluminium or solid
tantalum bypass capacitor is needed between the input pin
and the ground pin, to prevent large voltage transients from
appearing at the input. It must be located near the regulator
and use short leads. With most electrolytic capacitors, the
capacitance value decreases and the ESR increases with
lower temperatures. For reliable operation in temperatures
below −25°C larger values of the input capacitor may be
needed. Also paralleling a ceramic or solid tantalum
capacitor will increase the regulator stability at cold
temperatures.
RMS Current Rating of C
in
The important parameter of the input capacitor is the RMS
current rating. Capacitors that are physically large and have
large surface area will typically have higher RMS current
ratings. For a given capacitor value, a higher voltage
electrolytic capacitor will be physically larger than a lower
voltage capacitor, and thus be able to dissipate more heat to
the surrounding air, and therefore will have a higher RMS
current rating. The consequence of operating an electrolytic
capacitor beyond the RMS current rating is a shortened
operating life. In order to assure maximum capacitor
operating lifetime, the capacitor’s RMS ripple current rating
should be:
The Output Capacitor Requires an ESR Value
That Has an Upper and Lower Limit
As mentioned above, a low ESR value is needed for low
output ripple voltage, typically 1% to 2% of the output
voltage. But if the selected capacitor’s ESR is extremely low
(below 0.05 W), there is a possibility of an unstable feedback
loop, resulting in oscillation at the output. This situation can
occur when a tantalum capacitor, that can have a very low
ESR, is used as the only output capacitor.
At Low Temperatures, Put in Parallel Aluminium
Electrolytic Capacitors with Tantalum Capacitors
Electrolytic capacitors are not recommended for
temperatures below −25°C. The ESR rises dramatically at
cold temperatures and typically rises 3 times at −25°C and
as much as 10 times at −40°C. Solid tantalum capacitors
have much better ESR spec at cold temperatures and are
recommended for temperatures below −25°C. They can be
also used in parallel with aluminium electrolytics. The value
of the tantalum capacitor should be about 10% or 20% of the
total capacitance. The output capacitor should have at least
50% higher RMS ripple current rating at 150 kHz than the
peak−to−peakinductor ripple current.
Irms > 1.2 x d x ILoad
where d is the duty cycle, for a buck regulator
V
t
on
T
out
d +
|V
+
V
in
|
t
on
T
out
|V | ) V
and d +
+
for a buck*boost regulator.
out
in
Output Capacitor (Cout
)
For low output ripple voltage and good stability, low ESR
output capacitors are recommended. An output capacitor
has two main functions: it filters the output and provides
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14
LM2596
Catch Diode
ripple voltage. On the other hand it does require larger
Locate the Catch Diode Close to the LM2596
The LM2596 is a step−down buck converter; it requires a
fast diode to provide a return path for the inductor current
when the switch turns off. This diode must be located close
to the LM2596 using short leads and short printed circuit
traces to avoid EMI problems.
inductor values to keep the inductor current flowing
continuously, especially at low output load currents and/or
high input voltages.
To simplify the inductor selection process, an inductor
selection guide for the LM2596 regulator was added to this
data sheet (Figure 18). This guide assumes that the regulator
is operating in the continuous mode, and selects an inductor
that will allow a peak−to−peak inductor ripple current to be
a certain percentage of the maximum design load current.
This percentage is allowed to change as different design load
currents are selected. For light loads (less than
approximately 300 mA) it may be desirable to operate the
regulator in the discontinuous mode, because the inductor
value and size can be kept relatively low. Consequently, the
percentage of inductor peak−to−peak current increases. This
discontinuous mode of operation is perfectly acceptable for
this type of switching converter. Any buck regulator will be
forced to enter discontinuous mode if the load current is light
enough.
Use a Schottky or a Soft Switching
Ultra−Fast Recovery Diode
Since the rectifier diodes are very significant sources of
losses within switching power supplies, choosing the
rectifier that best fits into the converter design is an
important process. Schottky diodes provide the best
performance because of their fast switching speed and low
forward voltage drop.
They provide the best efficiency especially in low output
voltage applications (5.0 V and lower). Another choice
could be Fast−Recovery, or Ultra−Fast Recovery diodes. It
has to be noted, that some types of these diodes with an
abrupt turnoff characteristic may cause instability or
EMI troubles.
A fast−recovery diode with soft recovery characteristics
can better fulfill some quality, low noise design requirements.
Table 2 provides a list of suitable diodes for the LM2596
regulator. Standard 50/60 Hz rectifier diodes, such as the
1N4001 series or 1N5400 series are NOT suitable.
2.0 A
Inductor
Current
Waveform
0 A
Inductor
2.0 A
Power
Switch
Current
The magnetic components are the cornerstone of all
switching power supply designs. The style of the core and
the winding technique used in the magnetic component’s
design has a great influence on the reliability of the overall
power supply.
Waveform
0 A
HORIZONTAL TIME BASE: 2.0 ms/DIV
Using an improper or poorly designed inductor can cause
high voltage spikes generated by the rate of transitions in
current within the switching power supply, and the
possibility of core saturation can arise during an abnormal
operational mode. Voltage spikes can cause the
semiconductors to enter avalanche breakdown and the part
can instantly fail if enough energy is applied. It can also
cause significant RFI (Radio Frequency Interference) and
EMI (Electro−Magnetic Interference) problems.
Figure 19. Continuous Mode Switching Current
Waveforms
Selecting the Right Inductor Style
Some important considerations when selecting a core type
are core material, cost, the output power of the power supply,
the physical volume the inductor must fit within, and the
amount of EMI (Electro−Magnetic Interference) shielding
that the core must provide. The inductor selection guide
covers different styles of inductors, such as pot core, E−core,
toroid and bobbin core, as well as different core materials
such as ferrites and powdered iron from different
manufacturers.
For high quality design regulators the toroid core seems to
be the best choice. Since the magnetic flux is contained
within the core, it generates less EMI, reducing noise
problems in sensitive circuits. The least expensive is the
bobbin core type, which consists of wire wound on a ferrite
rod core. This type of inductor generates more EMI due to
the fact that its core is open, and the magnetic flux is not
contained within the core.
Continuous and Discontinuous Mode of Operation
The LM2596 step−down converter can operate in both the
continuous and the discontinuous modes of operation. The
regulator works in the continuous mode when loads are
relatively heavy, the current flows through the inductor
continuously and never falls to zero. Under light load
conditions, the circuit will be forced to the discontinuous
mode when inductor current falls to zero for certain period
of time (see Figure 19 and Figure 20). Each mode has
distinctively different operating characteristics, which can
affect the regulator performance and requirements. In many
cases the preferred mode of operation is the continuous
mode. It offers greater output power, lower peak currents in
the switch, inductor and diode, and can have a lower output
When multiple switching regulators are located on the
same printed circuit board, open core magnetics can cause
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15
LM2596
interference between two or more of the regulator circuits,
inductor and/or the LM2596. Different inductor types have
different saturation characteristics, and this should be kept
in mind when selecting an inductor.
especially at high currents due to mutual coupling. A toroid,
pot core or E−core (closed magnetic structure) should be
used in such applications.
Do Not Operate an Inductor Beyond its
Maximum Rated Current
0.4 A
Inductor
Current
Exceeding an inductor’s maximum current rating may
cause the inductor to overheat because of the copper wire
losses, or the core may saturate. Core saturation occurs when
the flux density is too high and consequently the cross
sectional area of the core can no longer support additional
lines of magnetic flux.
This causes the permeability of the core to drop, the
inductance value decreases rapidly and the inductor begins
to look mainly resistive. It has only the DC resistance of the
winding. This can cause the switch current to rise very
rapidly and force the LM2596 internal switch into
cycle−by−cyclecurrent limit, thus reducing the DC output
load current. This can also result in overheating of the
Waveform
0 A
0.4 A
Power
Switch
Current
Waveform
0 A
HORIZONTAL TIME BASE: 2.0 ms/DIV
Figure 20. Discontinuous Mode Switching Current
Waveforms
GENERAL RECOMMENDATIONS
Output Voltage Ripple and Transients
Minimizing the Output Ripple
Source of the Output Ripple
In order to minimize the output ripple voltage it is possible
to enlarge the inductance value of the inductor L1 and/or to
use a larger value output capacitor. There is also another way
to smooth the output by means of an additional LC filter (20
mH, 100 mF), that can be added to the output (see Figure 30)
to further reduce the amount of output ripple and transients.
With such a filter it is possible to reduce the output ripple
voltage transients 10 times or more. Figure 21 shows the
difference between filtered and unfiltered output waveforms
of the regulator shown in Figure 30.
Since the LM2596 is a switch mode power supply
regulator, its output voltage, if left unfiltered, will contain a
sawtooth ripple voltage at the switching frequency. The
output ripple voltage value ranges from 0.5% to 3% of the
output voltage. It is caused mainly by the inductor sawtooth
ripple current multiplied by the ESR of the output capacitor.
Short Voltage Spikes and How to Reduce Them
The regulator output voltage may also contain short
voltage spikes at the peaks of the sawtooth waveform (see
Figure 21). These voltage spikes are present because of the
fast switching action of the output switch, and the parasitic
inductance of the output filter capacitor. There are some
other important factors such as wiring inductance, stray
capacitance, as well as the scope probe used to evaluate these
transients, all these contribute to the amplitude of these
spikes. To minimize these voltage spikes, low inductance
capacitors should be used, and their lead lengths must be
kept short. The importance of quality printed circuit board
layout design should also be highlighted.
The lower waveform is from the normal unfiltered output
of the converter, while the upper waveform shows the output
ripple voltage filtered by an additional LC filter.
Heatsinking and Thermal Considerations
The Through−Hole Package TO−220
The LM2596 is available in two packages, a 5−pin
2
TO−220(T, TV) and a 5−pin surface mount D PAK(D2T).
Although the TO−220(T) package needs a heatsink under
most conditions, there are some applications that require no
heatsink to keep the LM2596 junction temperature within
the allowed operating range. Higher ambient temperatures
require some heat sinking, either to the printed circuit (PC)
board or an external heatsink.
Voltage spikes
caused by
switching action
of the output
switch and the
parasitic
Filtered
Output
Voltage
The Surface Mount Package D2PAK and its
Heatsinking
The other type of package, the surface mount D PAK, is
inductance of the
output capacitor
2
designed to be soldered to the copper on the PC board. The
copper and the board are the heatsink for this package and
the other heat producing components, such as the catch
diode and inductor. The PC board copper area that the
Unfiltered
Output
Voltage
2
2
package is soldered to should be at least 0.4 in (or 260 mm )
HORIZONTAL TIME BASE: 5.0 ms/DIV
2
and ideally should have 2 or more square inches (1300 mm )
Figure 21. Output Ripple Voltage Waveforms
of 0.0028 inch copper. Additional increases of copper area
2
2
beyond approximately 6.0 in (4000 mm ) will not improve
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16
LM2596
heat dissipation significantly. If further thermal
improvements are needed, double sided or multilayer PC
boards with large copper areas should be considered. In
order to achieve the best thermal performance, it is highly
recommended to use wide copper traces as well as large
areas of copper in the printed circuit board layout. The only
exception to this is the OUTPUT (switch) pin, which should
not have large areas of copper (see page 8 ‘PCB Layout
Guideline’).
V
is the regulator output voltage,
is the load current.
The dynamic switching losses during turn−on and
O
I
Load
turn−off can be neglected if proper type catch diode is used.
Packages Not on a Heatsink (Free−Standing)
For a free−standing application when no heatsink is used,
the junction temperature can be determined by the following
expression:
TJ = (RqJA) (PD) + TA
Thermal Analysis and Design
The following procedure must be performed to determine
whether or not a heatsink will be required. First determine:
where (R )(P ) represents the junction temperature rise
caused by the dissipated power and T is the maximum
ambient temperature.
qJA
D
A
1. P
2. T
3. T
maximum regulator power dissipation in the
application.
maximum ambient temperature in the
application.
maximum allowed junction temperature
(125°C for the LM2596). For a conservative
design, the maximum junction temperature
should not exceed 110°C to assure safe
operation. For every additional +10°C
temperature rise that the junction must
withstand, the estimated operating lifetime
of the component is halved.
D(max)
Packages on a Heatsink
)
A(max
If the actual operating junction temperature is greater than
the selected safe operating junction temperature determined
in step 3, than a heatsink is required. The junction
temperature will be calculated as follows:
J(max)
TJ = PD (RqJA + RqCS + RqSA) + TA
where
R
qJC
R
qCS
R
qSA
is the thermal resistance junction−case,
is the thermal resistance case−heatsink,
is the thermal resistance heatsink−ambient.
If the actual operating temperature is greater than the
selected safe operating junction temperature, then a larger
heatsink is required.
4. R
5. R
package thermal resistance junction−case.
package thermal resistance junction−ambient.
qJC
qJA
(Refer to Maximum Ratings on page 2 of this data sheet or
and R values).
Some Aspects That can Influence Thermal Design
It should be noted that the package thermal resistance and
the junction temperature rise numbers are all approximate,
and there are many factors that will affect these numbers,
such as PC board size, shape, thickness, physical position,
location, board temperature, as well as whether the
surrounding air is moving or still.
R
qJC
qJA
The following formula is to calculate the approximate
total power dissipated by the LM2596:
PD = (Vin x IQ) + d x ILoad x Vsat
where d is the duty cycle and for buck converter
Other factors are trace width, total printed circuit copper
area, copper thickness, single− or double−sided, multilayer
board, the amount of solder on the board or even color of the
traces.
The size, quantity and spacing of other components on the
board can also influence its effectiveness to dissipate the heat.
V
V
t
on
T
O
d +
+
,
in
I
(quiescent current) and V can be found in the
Q
sat
LM2596 data sheet,
V
is minimum input voltage applied,
in
R4
Feedback
12 to 40 V
Unregulated
+V
in
L1
33 mH
DC Input
LM2596−ADJ
C
in
100 mF/50 V
ON/OFF
GND
R3
D1
1N5822
C
−12 V @ 0.7 A
Regulated
Output
out
220 mF
Figure 22. Inverting Buck−Boost Develops −12 V
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17
LM2596
ADDITIONAL APPLICATIONS
Using a delayed startup arrangement, the input capacitor
can charge up to a higher voltage before the switch−mode
regulator begins to operate.
Inverting Regulator
An inverting buck−boost regulator using the
LM2596−ADJ is shown in Figure 22. This circuit converts
a positive input voltage to a negative output voltage with a
common ground by bootstrapping the regulators ground to
the negative output voltage. By grounding the feedback pin,
the regulator senses the inverted output voltage and
regulates it.
In this example the LM2596−12 is used to generate a
−12 V output. The maximum input voltage in this case
cannot exceed +28 V because the maximum voltage
appearing across the regulator is the absolute sum of the
input and output voltages and this must be limited to a
maximum of 40 V.
This circuit configuration is able to deliver approximately
0.7 A to the output when the input voltage is 12 V or higher.
At lighter loads the minimum input voltage required drops
to approximately 4.7 V, because the buck−boost regulator
topology can produce an output voltage that, in its absolute
value, is either greater or less than the input voltage.
Since the switch currents in this buck−boost configuration
are higher than in the standard buck converter topology, the
available output current is lower.
This type of buck−boost inverting regulator can also
require a larger amount of startup input current, even for
light loads. This may overload an input power source with
a current limit less than 5.0 A.
The high input current needed for startup is now partially
supplied by the input capacitor C .
in
It has been already mentioned above, that in some
situations, the delayed startup or the undervoltage lockout
features could be very useful. A delayed startup circuit
applied to a buck−boost converter is shown in Figure 27.
Figure 29 in the “Undervoltage Lockout” section describes
an undervoltage lockout feature for the same converter
topology.
Design Recommendations:
The inverting regulator operates in a different manner
than the buck converter and so a different design procedure
has to be used to select the inductor L1 or the output
capacitor C
.
out
The output capacitor values must be larger than what is
normally required for buck converter designs. Low input
voltages or high output currents require a large value output
capacitor (in the range of thousands of mF).
The recommended range of inductor values for the
inverting converter design is between 68 mH and 220 mH. To
select an inductor with an appropriate current rating, the
inductor peak current has to be calculated.
The following formula is used to obtain the peak inductor
current:
I
(V ) |V |)
V
x t
on
Load in
O
in
2L
Such an amount of input startup current is needed for at
least 2.0 ms or more. The actual time depends on the output
voltage and size of the output capacitor.
Because of the relatively high startup currents required by
this inverting regulator topology, the use of a delayed startup
or an undervoltage lockout circuit is recommended.
I
[
)
peak
V
1
in
x
|V |
O
1.0
osc
where t
+
, and f
+ 52 kHz.
osc
on
V
) |V |
f
in
O
Under normal continuous inductor current operating
conditions, the worst case occurs when V is minimal.
in
R4
Feedback
12 to 40 V
Unregulated
+V
in
L1
33 mH
DC Input
LM2596−ADJ
C
C1
in
100 mF/50 V
0.1 mF
ON/OFF
GND
R3
D1
1N5822
C
−12 V @ 0.7 A
Regulated
Output
out
220 mF
R2
47k
Figure 23. Inverting Buck−Boost Develops −12 V
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18
LM2596
Shutdown
Input
+V
0
Off
+V
+V
in
in
LM2596−XX
On
1
R2
5.6 k
C
R1
100 mF 47 k
in
+V
in
+V
in
5
ON/OFF 3 GN
D
Shutdown
Input
1
5.0 V
0
LM2596−XX
C
Off
in
100 mF
R3
470
On
Q1
2N3906
R2
47 k
5
ON/OFF 3 GN
D
-V
out
R1
12 k
MOC8101
-V
out
NOTE: This picture does not show the complete circuit.
NOTE: This picture does not show the complete circuit.
Figure 24. Inverting Buck−Boost Regulator Shutdown
Figure 25. Inverting Buck−Boost Regulator Shutdown
Circuit Using an Optocoupler
Circuit Using a PNP Transistor
With the inverting configuration, the use of the ON/OFF
pin requires some level shifting techniques. This is caused
by the fact, that the ground pin of the converter IC is no
longer at ground. Now, the ON/OFF pin threshold voltage
(1.3 V approximately) has to be related to the negative
output voltage level. There are many different possible shut
down methods, two of them are shown in Figures 24 and 25.
Negative Boost Regulator
This example is a variation of the buck−boost topology
and it is called negative boost regulator. This regulator
experiences relatively high switch current, especially at low
input voltages. The internal switch current limiting results in
lower output load current capability.
The circuit in Figure 26 shows the negative boost
configuration. The input voltage in this application ranges
from −5.0 V to −12 V and provides a regulated −12 V output.
If the input voltage is greater than −12 V, the output will rise
above −12 V accordingly, but will not damage the regulator.
R4
C
out
470 mF
Feedback
+V
in
LM2596−ADJ
C
in
100 mF/
D1
1N5822
50 V
ON/OFF
GND
−12 V @ 0.7 A
Regulated
Output
R3
−12 V
Unregulated
DC Input
L1
33 mH
Figure 26. Negative Boost Regulator
Design Recommendations:
The same design rules as for the previous inverting
buck−boost converter can be applied. The output capacitor
values for the negative boost regulator is the same as for
inverting converter design.
Another important point is that these negative boost
converters cannot provide current limiting load protection in
the event of a short in the output so some other means, such
as a fuse, may be necessary to provide the load protection.
C
out
must be chosen larger than would be required for a what
standard buck converter. Low input voltages or high output
currents require a large value output capacitor (in the range
of thousands of mF). The recommended range of inductor
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19
LM2596
Delayed Startup
There are some applications, like the inverting regulator
already mentioned above, which require a higher amount of
startup current. In such cases, if the input power source is
limited, this delayed startup feature becomes very useful.
To provide a time delay between the time when the input
voltage is applied and the time when the output voltage
comes up, the circuit in Figure 27 can be used. As the input
voltage is applied, the capacitor C1 charges up, and the
voltage across the resistor R2 falls down. When the voltage
on the ON/OFF pin falls below the threshold value 1.3 V, the
regulator starts up. Resistor R1 is included to limit the
maximum voltage applied to the ON/OFF pin. It reduces the
power supply noise sensitivity, and also limits the capacitor
C1 discharge current, but its use is not mandatory.
+V
+V
in
in
LM2596−XX
1
C
in
100 mF
R2
10 k
R3
47 k
5
ON/OFF 3 GN
D
Z1
1N5242B
Q1
2N3904
R1
10 k
V
th
≈ 13 V
NOTE: This picture does not show the complete circuit.
When a high 50 Hz or 60 Hz (100 Hz or 120 Hz
respectively) ripple voltage exists, a long delay time can
cause some problems by coupling the ripple into the
ON/OFF pin, the regulator could be switched periodically
on and off with the line (or double) frequency.
Figure 28. Undervoltage Lockout Circuit for
Buck Converter
The following formula is used to obtain the peak inductor
current:
I
(V ) |V |)
Load in
V
x t
on
O
in
2L
+V
+V
I
[
)
in
in
peak
V
LM2596−XX
1
in
x
1
|V |
O
1.0
osc
where t
+
, and f
+ 52 kHz.
osc
on
C1
0.1 mF
V
) |V |
f
in
O
5
ON/OFF 3 GN
D
Under normal continuous inductor current operating
conditions, the worst case occurs when V is minimal.
C
in
100 mF
in
R1
47 k
R2
47 k
+V
in
+V
in
LM2596−XX
1
NOTE: This picture does not show the complete circuit.
C
in
100 mF
R2
15 k
R3
47 k
5
ON/OFF 3 GN
D
Figure 27. Delayed Startup Circuitry
Undervoltage Lockout
Z1
1N5242B
Some applications require the regulator to remain off until
the input voltage reaches a certain threshold level. Figure 28
shows an undervoltage lockout circuit applied to a buck
regulator. A version of this circuit for buck−boost converter
is shown in Figure 29. Resistor R3 pulls the ON/OFF pin
high and keeps the regulator off until the input voltage
reaches a predetermined threshold level with respect to the
ground Pin 3, which is determined by the following
expression:
V
th
≈ 13 V
Q1
2N3904
R1
15 k
V
out
NOTE: This picture does not show the complete circuit.
Figure 29. Undervoltage Lockout Circuit for
R2
Buck−Boost Converter
) ǒ1.0 ) Ǔ V
Z1
( )
Q1
V
[ V
th
BE
R1
Adjustable Output, Low−Ripple Power Supply
A 3.0 A output current capability power supply that
features an adjustable output voltage is shown in Figure 30.
This regulator delivers 3.0 A into 1.2 V to 35 V output.
The input voltage ranges from roughly 3.0 V to 40 V. In order
to achieve a 10 or more times reduction of output ripple, an
additional L−C filter is included in this circuit.
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20
LM2596
Feedback
40 V Max
Unregulated
DC Input
4
+V
in
L1
33 mH
L2
20 mH
LM2596−Adj
Output
Voltage
1
Output
2
ON/OFF
1.2 to 35 V @ 3.0 A
C
R2
50 k
in
100 mF
3
GN
D
5
C
out
220 mF
D1
1N5822
C1
100 mF
R1
1.21 k
Optional Output
Ripple Filter
Figure 30. 1.2 to 35 V Adjustable 3.0 A Power Supply with Low Output Ripple
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21
LM2596
THE LM2596 STEP−DOWN VOLTAGE REGULATOR WITH 5.0 V @ 3.0 A OUTPUT POWER CAPABILITY.
TYPICAL APPLICATION WITH THROUGH−HOLE PC BOARD LAYOUT
4
Feedback
Unregulated
DC Input
+V
in
L1
33 mH
LM2596−ADJ
Regulated
Output Filtered
1
+V = 10 V to 40 V
in
Output
2
ON/OFF
V
out2
= 5.0 V @ 3.0 A
R2
3.0 k
C
FF
3
GN
D
5
C1
100 mF
/50 V
C2
220 mF
/16 V
D1
1N5822
R1
1.0 k
ON/OFF
R2
R1
) ǒ1.0 )
Ǔ
V
+ V
out
ref
C1
C2
D1
L1
R1
R2
−
−
−
−
−
−
100 mF, 50 V, Aluminium Electrolytic
220 mF, 25 V, Aluminium Electrolytic
3.0 A, 40 V, Schottky Rectifier, 1N5822
33 mH, DO5040H, Coilcraft
1.0 kW, 0.25 W
V
ref
= 1.23 V
R1 is between 1.0 k and 5.0 k
3.0 kW, 0.25 W
Figure 31. Schematic Diagram of the 5.0 V @ 3.0 A Step−Down Converter Using the LM2596−ADJ
NOTE: Not to scale.
NOTE: Not to scale.
Figure 32. Printed Circuit Board Layout
Component Side
Figure 33. Printed Circuit Board Layout
Copper Side
References
• National Semiconductor LM2596 Data Sheet and Application Note
• National Semiconductor LM2595 Data Sheet and Application Note
• Marty Brown “Practical Switching Power Supply Design”, Academic Press, Inc., San Diego 1990
• Ray Ridley “High Frequency Magnetics Design”, Ridley Engineering, Inc. 1995
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22
LM2596
ORDERING INFORMATION
†
Device
Package
Shipping
LM2596TADJG
TO−220
(Pb−Free)
50 Units / Rail
LM2596TVADJG
LM2596DSADJG
LM2596DSADJR4G
TO−220 (F)
(Pb−Free)
50 Units / Rail
2
D PAK
50 Units / Rail
(Pb−Free)
2
D PAK
800 / Tape & Reel
(Pb−Free)
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specifications Brochure, BRD8011/D.
MARKING DIAGRAMS
2
TO−220
TV SUFFIX
CASE 314B
TO−220
T SUFFIX
CASE 314D
D PAK
DS SUFFIX
CASE 936A
LM
2596−ADJ
AWLYWWG
LM
2596T−ADJ
AWLYWWG
LM
2596T−ADJ
AWLYWWG
1
5
1
5
1
5
A
= Assembly Location
WL = Wafer Lot
= Year
WW = Work Week
= Pb−Free Package
Y
G
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23
LM2596
PACKAGE DIMENSIONS
TO−220
TV SUFFIX
CASE 314B−05
ISSUE L
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION D DOES NOT INCLUDE
INTERCONNECT BAR (DAMBAR) PROTRUSION.
DIMENSION D INCLUDING PROTRUSION SHALL
NOT EXCEED 0.043 (1.092) MAXIMUM.
C
B
−P−
OPTIONAL
CHAMFER
Q
F
E
A
U
INCHES
DIM MIN MAX
0.613 14.529 15.570
MILLIMETERS
L
S
MIN MAX
V
W
A
B
C
D
E
F
0.572
0.390
0.170
0.025
0.048
0.850
0.067 BSC
0.166 BSC
0.015
0.900
0.320
0.320 BSC
0.140
---
0.468
---
0.090
0.415
0.180
0.038
0.055
9.906 10.541
K
4.318
0.635
1.219
4.572
0.965
1.397
0.935 21.590 23.749
1.702 BSC
4.216 BSC
G
H
J
0.025
0.381
1.100 22.860 27.940
0.635
5X J
K
L
G
0.365
8.128
8.128 BSC
3.556
9.271
3.886
M
0.24 (0.610)
T
H
N
Q
S
U
V
W
5X D
0.153
0.620
N
--- 15.748
M
M
0.10 (0.254)
T P
0.505 11.888 12.827
0.735 --- 18.669
0.110 2.286 2.794
SEATING
PLANE
−T−
TO−220
T SUFFIX
CASE 314D−04
ISSUE F
NOTES:
SEATING
−T−
PLANE
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
B
C
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION D DOES NOT INCLUDE
INTERCONNECT BAR (DAMBAR) PROTRUSION.
DIMENSION D INCLUDING PROTRUSION SHALL
NOT EXCEED 10.92 (0.043) MAXIMUM.
−Q−
DETAIL A-A
B1
E
A
U
K
INCHES
DIM MIN MAX
0.613 14.529 15.570
MILLIMETERS
L
MIN MAX
A
0.572
0.390
1 2 3 4 5
B
0.415
0.415
0.180
0.038
0.055
9.906 10.541
9.525 10.541
B1 0.375
C
D
E
G
H
J
0.170
0.025
0.048
4.318
0.635
1.219
4.572
0.965
1.397
0.067 BSC
1.702 BSC
0.087
0.015
0.977
0.320
0.140
0.105
0.112
0.025
2.210
0.381
2.845
0.635
J
H
G
K
L
1.045 24.810 26.543
D 5 PL
0.365
0.153
0.117
8.128
3.556
2.667
9.271
3.886
2.972
Q
U
M
M
0.356 (0.014)
T Q
B
B1
DETAIL A−A
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24
LM2596
PACKAGE DIMENSIONS
D2PAK
D2T SUFFIX
CASE 936A−02
ISSUE C
NOTES:
−T−
TERMINAL 6
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
OPTIONAL
CHAMFER
A
E
U
2. CONTROLLING DIMENSION: INCH.
3. TAB CONTOUR OPTIONAL WITHIN DIMENSIONS A
AND K.
S
4. DIMENSIONS U AND V ESTABLISH A MINIMUM
MOUNTING SURFACE FOR TERMINAL 6.
5. DIMENSIONS A AND B DO NOT INCLUDE MOLD
FLASH OR GATE PROTRUSIONS. MOLD FLASH
AND GATE PROTRUSIONS NOT TO EXCEED 0.025
(0.635) MAXIMUM.
K
V
B
H
1
2
3
4 5
M
L
INCHES
MILLIMETERS
DIM
A
B
C
D
E
G
H
K
L
MIN
MAX
0.403
0.368
0.180
0.036
0.055
MIN
9.804
9.042
4.318
0.660
1.143
MAX
10.236
9.347
4.572
0.914
1.397
D
M
P
N
0.386
0.356
0.170
0.026
0.045
0.067 BSC
0.539
0.010 (0.254)
T
G
R
1.702 BSC
0.579 13.691
14.707
0.050 REF
1.270 REF
0.000
0.088
0.018
0.058
0.010
0.102
0.026
0.078
0.000
2.235
0.457
1.473
0.254
2.591
0.660
1.981
C
M
N
P
R
S
U
V
5_ REF
5_ REF
0.116 REF
0.200 MIN
0.250 MIN
2.946 REF
5.080 MIN
6.350 MIN
SOLDERING FOOTPRINT*
8.38
0.33
1.702
0.067
10.66
0.42
1.016
0.04
3.05
0.12
16.02
0.63
mm
inches
ǒ
Ǔ
SCALE 3:1
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should
Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates,
and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death
associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal
Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
LITERATURE FULFILLMENT:
N. American Technical Support: 800−282−9855 Toll Free
USA/Canada
Europe, Middle East and Africa Technical Support:
Phone: 421 33 790 2910
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Phone: 81−3−5773−3850
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Order Literature: http://www.onsemi.com/orderlit
Literature Distribution Center for ON Semiconductor
P.O. Box 5163, Denver, Colorado 80217 USA
Phone: 303−675−2175 or 800−344−3860 Toll Free USA/Canada
Fax: 303−675−2176 or 800−344−3867 Toll Free USA/Canada
Email: orderlit@onsemi.com
For additional information, please contact your local
Sales Representative
LM2596/D
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