MP2361DS-LF-Z [MPS]
Switching Regulator, Current-mode, 1400kHz Switching Freq-Max, PDSO8, ROHS COMPLIANT, MS-012AA, SOIC-8;
| 型号： | MP2361DS-LF-Z |
| 厂家： | 
MONOLITHIC POWER SYSTEMS |
| 描述： | Switching Regulator, Current-mode, 1400kHz Switching Freq-Max, PDSO8, ROHS COMPLIANT, MS-012AA, SOIC-8 开关 光电二极管 |
| 文件： | 总14页 (文件大小：296K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MP2361
2A, 23V, 1.4MHz
Step-Down Converter
The Future of Analog IC Technology
FEATURES
DESCRIPTION
•
•
•
2A Output Current
The MP2361 is a monolithic step-down switch
mode converter with a built-in internal power
MOSFET. It achieves 2A continuous output
current over a wide input supply range with
excellent load and line regulation.
0.18Ω Internal Power MOSFET Switch
Stable with Low ESR Output Ceramic
Capacitors
•
•
•
•
•
•
•
•
•
90% Efficiency
20μA Shutdown Mode
Fixed 1.4MHz Frequency
Thermal Shutdown
Cycle-by-Cycle Over Current Protection
Wide 4.75V to 23V Operating Input Range
Output Adjustable from 0.92V to 16V
Programmable Under Voltage Lockout
Available in QFN10, MSOP10, MSOP10E,
and SOIC8 Packages
Current mode operation provides fast transient
response and eases loop stabilization.
Fault condition protections include cycle-by-cycle
current limiting and thermal shutdown. In
shutdown mode the regulator draws 20μA of
supply
current.
Programmable
soft-start
minimizes the inrush supply current and the
output overshoot at initial startup.
The MP2361 requires a minimum number of
readily available standard external components.
APPLICATIONS
•
•
•
•
Distributed Power Systems
Battery Charger
DSL Modems
Pre-Regulator for Linear Regulators
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
TYPICAL APPLICATION
Efficiency vs
INPUT
Load Current
4.75V to 23V
C5
10nF
100
4
2
V
=5V
OUT
IN
BS
90
80
70
60
50
9
V
OUT
2.5V/2A
5
7
EN
SS
SW
D1
B220A
MP2361
10
FB
V
=2.5V
OUT
GND
COMP
V
=3.3V
OUT
6
8
C3
1.8nF
C4
10nF
C6
OPEN
0
0.5
1.0
1.5
2.0
LOAD CURRENT (A)
MP2361_TAC_S01
MP2361-EC01
MP2361 Rev. 1.4
9/22/2011
www.MonolithicPower.com
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© 2011 MPS. All Rights Reserved.
1
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
ORDERING INFORMATION
Part Number*
MP2361DQ*
Package
Top Marking
Free Air Temperature (TA)
–40°C to +85°C
QFN10 (3mm x 3mm)
MSOP10E
C8
MP2361DH**
MP2361DK***
MP2361DS****
2361D
–40°C to +85°C
MSOP10
2361D
–40°C to +85°C
SOIC8
MP2361DS
–40°C to +85°C
*For Tape & Reel, add suffix –Z (eg. MP2361DQ–Z);
For RoHS, compliant packaging, add suffix –LF (eg. MP2361DQ –LF–Z)
** For Tape & Reel, add suffix –Z (eg. MP2361DH–Z);
For RoHS, compliant packaging, add suffix –LF (eg. MP2361DH –LF–Z)
*** For Tape & Reel, add suffix –Z (eg. MP2361DK–Z);
For RoHS, compliant packaging, add suffix –LF (eg. MP2361DK –LF–Z)
**** For Tape & Reel, add suffix –Z (eg. MP2361DS–Z);
For RoHS, compliant packaging, add suffix –LF (eg. MP2361DS –LF–Z)
PACKAGE REFERENCE
TOP VIEW
TOP VIEW
TOP VIEW
TOP VIEW
NC
BS
NC
IN
1
2
3
4
5
10
9
SS
NC
1
2
3
4
5
10 SS
NC
BS
NC
IN
1
2
3
4
5
10
9
SS
EN
BS
NC
IN
9
8
7
6
EN
BS
VIN
1
2
3
4
8
7
6
5
SS
EN
8
COMP
FB
COMP
FB
8
COMP
FB
7
EN
7
SW
6
GND
VSW
GND
COMP
FB
SW
6
GND
SW
GND
EXPOSED PAD
ON BACKSIDE
Connect to GND
EXPOSED PAD
ON BACKSIDE
QFN10
(3mm x 3mm)
MSOP10E
MSOP10
SOIC8
MP2361 Rev. 1.4
9/22/2011
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2
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
ABSOLUTE MAXIMUM RATINGS (1)
Supply Voltage (VIN)..................................... 25V
Switch Node Voltage (VSW).......................... 26V
Bootstrap Voltage (VBS) .......................VSW + 6V
Feedback Voltage (VFB) .................–0.3V to +6V
Enable/UVLO Voltage (VEN)...........–0.3V to +6V
Comp Voltage (VCOMP) ...................–0.3V to +6V
Thermal Resistance (4)
θJA
θJC
QFN10 (3mmx3mm)...............50...... 12... °C/W
MSOP10................................150..... 65... °C/W
MSOP10E ..............................55...... 12... °C/W
SOIC8.....................................90...... 45... °C/W
Notes:
Continuous Power Dissipation
(TA = +25°C)(2)
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ(MAX), the junction-to-
ambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD(MAX)=(TJ(MAX)-
TA)/ θJA. Exceeding the maximum allowable power dissipation
will cause excessive die temperature, and the regulator will go
into thermal shutdown. Internal thermal shutdown circuitry
protects the device from permanent damage.
QFN10 (3mmx3mm) .................................. 2.5W
MSOP10 .................................................. 0.83W
MSOP10E.................................................. 2.3W
SOIC8...................................................... 1.38W
Junction Temperature.............................+150°C
Lead Temperature ..................................+260°C
Storage Temperature.............. –65°C to +150°C
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7 4-layer PCB.
Recommended Operating Conditions (3)
Supply Voltage (VIN).......................4.75V to 23V
Operating Junct. Temp (TJ).... –40°C to +125°C
MP2361 Rev. 1.4
9/22/2011
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3
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, TA = +25°C, unless otherwise noted.
Parameter
Symbol Condition
Min
Typ
0.920
0.18
10
0
3.5
Max
Units
V
Feedback Voltage
VFB
0.892
0.948
4.75V ≤ VIN ≤ 23V
Upper Switch On Resistance
Lower Switch On Resistance
Upper Switch Leakage
Current Limit (5)
RDS(ON)1
RDS(ON)2
Ω
Ω
μA
A
VEN = 0V; VSW = 0V
10
2.8
Current Sense Transconductance
Output Current to Comp Pin Voltage
GCS
1.95
A/V
Error Amplifier Voltage Gain
Error Amplifier Transconductance
Oscillator Frequency
AVEA
GEA
fS
400
930
1.4
V/V
μA/V
MHz
kHz
630
1230
ΔIC = ±10μA
Short Circuit Frequency
VFB = 0V
210
Soft-Start Pin Equivalent
Output Resistance
9
kΩ
Maximum Duty Cycle
Minimum On Time
DMAX VFB = 0.8V
tON
70
100
1.0
%
ns
V
EN Shutdown Threshold Voltage
Enable Pull Up Current
EN UVLO Threshold Rising
EN UVLO Threshold Hysteresis
Supply Current (Shutdown)
Supply Current (Quiescent)
Thermal Shutdown
VEN
IEN
ICC > 100μA
0.7
1.3
VEN = 0V
1.0
μA
V
VUVLO VEN Rising
2.37
2.50
210
20
2.62
mV
μA
mA
°C
IOFF
36
VEN ≤ 0.4V
ION
1.2
1.4
VEN ≥ 3V
160
Note:
5) Equivalent output current = 1.5A ≥ 50% Duty Cycle
2.0A ≤ 50% Duty Cycle
Assumes ripple current = 30% of load current.
Slope compensation changes current limit above 40% duty cycle.
MP2361 Rev. 1.4
9/22/2011
www.MonolithicPower.com
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4
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
PIN FUNCTIONS
QFN and
MSOP
Pin #
SOIC
Pin #
Name
Description
1
NC
BS
No Connect.
Bootstrap (C5). This capacitor is needed to drive the power switch’s gate above
the supply voltage. It is connected between SW and BS pins to form a floating
supply across the power switch driver. The voltage across C5 is about 5V and is
supplied by the internal +5V supply when the SW pin voltage is low.
2
1
3
4
5
NC
IN
No Connect.
Supply Voltage. The MP2361 operates from a +4.75V to +23V unregulated
input. C1 is needed to prevent large voltage spikes from appearing at the input.
2
3
SW
Switch. This connects the inductor to either IN through M1 or to GND through M2.
Ground. This pin is the voltage reference for the regulated output voltage. For
this reason care must be taken in its layout. This node should be placed outside
of the D1 to C1 ground path to prevent switching current spikes from inducing
voltage noise into the part.
6
4
GND
Feedback. An external resistor divider from the output to GND, tapped to the FB
pin sets the output voltage. To prevent current limit run away during a short
circuit fault condition the frequency foldback comparator lowers the oscillator
frequency when the FB voltage is below 400mV.
7
8
5
6
FB
Compensation. This node is the output of the transconductance error amplifier and the
input to the current comparator. Frequency compensation is done at this node by
connecting a series R-C to ground. See the compensation section for exact details.
COMP
Enable/UVLO. A voltage greater than 2.62V enables operation. Leave EN
unconnected for automatic startup. An Under Voltage Lockout (UVLO) function
can be implemented by the addition of a resistor divider from VIN to GND. For
complete low current shutdown it’s the EN pin voltage needs to be less than
700mV.
9
7
8
EN
SS
Soft-Start Pin. Connect SS to an external capacitor to program the soft-start. If
unused, leave it open.
10
MP2361 Rev. 1.4
9/22/2011
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5
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
OPERATION
The MP2361 is a current mode regulator. That
is, the COMP pin voltage is proportional to the
peak inductor current. At the beginning of a
cycle: the upper transistor M1 is off; the lower
transistor M2 is on (see Figure 1); the COMP
pin voltage is higher than the current sense
amplifier output; and the current comparator’s
output is low. The rising edge of the 1.4MHz
CLK signal sets the RS Flip-Flop. Its output
turns off M2 and turns on M1 thus connecting
the SW pin and inductor to the input supply.
The increasing inductor current is sensed and
amplified by the Current Sense Amplifier. Ramp
compensation is summed to Current Sense
Amplifier output and compared to the Error
Amplifier output by the Current Comparator.
When the Current Sense Amplifier plus Slope
Compensation signal exceeds the COMP pin
voltage, the RS Flip-Flop is reset and the
MP2361 reverts to its initial M1 off, M2 on state.
If the Current Sense Amplifier plus Slope
Compensation signal does not exceed the
COMP voltage, then the falling edge of the CLK
resets the Flip-Flop.
The output of the Error Amplifier integrates the
voltage difference between the feedback and
the 0.92V bandgap reference. The polarity is
such that the FB pin voltage lower than 0.92V
increases the COMP pin voltage. Since the
COMP pin voltage is proportional to the peak
inductor current an increase in its voltage
increases current delivered to the output. The
lower 10Ω switch ensures that the bootstrap
capacitor voltage is charged during light load
conditions. External Schottky Diode D1 carries
the inductor current when M1 is off.
4
IN
CURRENT
SENSE
AMPLIFIER
INTERNAL
5V
REGULATORS
+
--
5V
OSCILLATOR
SLOPE
COMP
210KHz/
1.4MHz
2
5
BS
CLK
+
--
+
S
R
Q
Q
SW
CURRENT
COMPARATOR
SHUTDOWN
COMPARATOR
--
0.7V
9
EN
LOCKOUT
COMPARATOR
+
--
+
2.29V/
2.50V
--
+
6
1.8V
GND
0.4V
0.92V
FB
--
FREQUENCY
FOLDBACK
COMPARATOR
ERROR
AMPLIFIER
7
10
8
SS
COMP
MP2361_BD01
Figure 1—Functional Block Diagram
MP2361 Rev. 1.4
9/22/2011
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MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
APPLICATION INFORMATION
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
COMPONENT SELECTION
Setting the Output Voltage
The output voltage is set using a resistive voltage
divider from the output voltage to FB pin. The
voltage divider divides the output voltage down to
the feedback voltage by the ratio:
⎛
⎞
⎟
⎟
⎠
VOUT
VOUT
VIN
⎜
ILP = ILOAD
+
× 1−
⎜
⎝
2× fS ×L
Where ILOAD is the load current.
R2
VFB = VOUT
R1+ R2
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is off. To
reduce losses due to the diode forward voltage
and recovery times, use a Schottky diode.
Thus the output voltage is:
R1+ R2
VOUT = 0.92 ×
R2
Where VOUT is the output voltage and VFB is the
feedback voltage.
Choose a diode whose maximum reverse
voltage rating is greater than the maximum
input voltage, and whose current rating is
greater than the maximum load current.
A typical value for R2 can be as high as 100kΩ,
but a typical value is 10kΩ. Using that value, R1
is determined by:
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required
to supply the AC current to the step-down
converter while maintaining the DC input
voltage. Use low ESR capacitors for the best
performance. Ceramic capacitors are preferred,
but tantalum or low-ESR electrolytic capacitors
may also suffice.
R1 = 10.87 × (VOUT − 0.92)
For example, for a 3.3V output voltage, R2 is
10kΩ, and R1 is 25.8kΩ.
Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value inductor
will result in less ripple current that will result in
lower output ripple voltage. However, the larger
value inductor will have a larger physical size,
higher series resistance, and/or lower saturation
current. A good rule for determining the
inductance to use is to allow the peak-to-peak
ripple current in the inductor to be approximately
30% of the maximum switch current limit. Also,
make sure that the peak inductor current is below
the maximum switch current limit. The inductance
value can be calculated by:
Since the input capacitor (C1) absorbs the input
switching current it requires an adequate ripple
current rating. The RMS current in the input
capacitor can be estimated by:
⎛
⎞
⎟
VOUT
VIN
VOUT
VIN
⎜
IC1 = ILOAD
×
× 1−
⎜
⎝
⎟
⎠
The worst-case condition occurs at VIN = 2VOUT
,
where:
ILOAD
IC1
=
2
⎛
⎞
⎟
⎟
⎠
VOUT
VOUT
⎜
L =
× 1−
For simplification, choose the input capacitor
whose RMS current rating greater than half of
the maximum load current.
⎜
⎝
fS × ΔIL
V
IN
Where fS is the switching frequency, ΔIL is the
peak-to-peak inductor ripple current and VIN is
the input voltage.
MP2361 Rev. 1.4
9/22/2011
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MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
The input capacitor can be electrolytic, tantalum
or ceramic. When using electrolytic or tantalum
capacitors, a small, high quality ceramic
capacitor, i.e. 0.1μF, should be placed as close
to the IC as possible. When using ceramic
capacitors, make sure that they have enough
capacitance to provide sufficient charge to
prevent excessive voltage ripple at input. The
input voltage ripple caused by capacitance can
be estimated by:
Compensation Components
The MP2361 employs current mode control for
easy compensation and fast transient response.
The system stability and transient response are
controlled through the COMP pin. COMP pin is
the output of the internal transconductance
error amplifier. A series capacitor-resistor
combination sets a pole-zero combination to
control the characteristics of the control system.
The DC gain of the voltage feedback loop is
given by:
⎛
⎜
⎝
⎞
⎟
⎟
⎠
ILOAD
VOUT
VIN
VOUT
⎜
ΔV
=
×
× 1−
IN
fS × C1
V
IN
VFB
AVDC = RLOAD × GCS × AVEA
×
VOUT
Output Capacitor
The output capacitor is required to maintain the
DC output voltage. Ceramic, tantalum, or low
ESR electrolytic capacitors are recommended.
Low ESR capacitors are preferred to keep the
output voltage ripple low. The output voltage
ripple can be estimated by:
Where RLOAD is the load resistor value, GCS is
the current sense transconductance and AVEA is
the error amplifier voltage gain.
The system has two poles of importance. One
is due to the compensation capacitor (C3) and
the output resistor of error amplifier, and the
other is due to the output capacitor and the load
resistor. These poles are located at:
⎛
⎜
⎝
⎞
⎟
⎟
⎛
⎞
⎟
⎟
⎠
VOUT
VOUT
VIN
1
⎜
⎜
ΔVOUT
=
× 1−
× RESR
+
⎜
⎝
fS × L
8 × fS × C2
⎠
GEA
Where L is the inductor value, RESR is the
equivalent series resistance (ESR) value of the
output capacitor and C2 is the output
capacitance value.
fP1
=
2π× C3× AVEA
1
fP2
GEA
=
2π × C2× RLOAD
In the case of ceramic capacitors, the
impedance at the switching frequency is
dominated by the capacitance. The output
voltage ripple is mainly caused by the
capacitance. For simplification, the output
voltage ripple can be estimated by:
Where
is
the
error
amplifier
transconductance.
The system has one zero of importance, due to the
compensation capacitor (C3) and the
compensation resistor (R3). This zero is located at:
⎛
⎜
⎝
⎞
VOUT
VOUT
VIN
1
⎜
⎟
⎟
⎠
ΔVOUT
=
× 1−
fZ1
=
2
8 × fS ×L × C2
2π × C3×R3
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero,
due to the ESR and capacitance of the output
capacitor, is located at:
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the
output ripple can be approximated to:
VOUT
VOUT
⎛
⎞
ΔVOUT
=
× ⎜1−
⎟ ×RESR
⎜
⎟
1
fS ×L
VIN
⎝
⎠
fESR
=
2π × C2×RESR
The characteristics of the output capacitor also
affect the stability of the regulation system. The
MP2361 can be optimized for a wide range of
capacitance and ESR values.
MP2361 Rev. 1.4
9/22/2011
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8
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
In this case, a third pole set by the
compensation capacitor (C6) and the
compensation resistor (R3) is used to
compensate the effect of the ESR zero on the
loop gain. This pole is located at:
3. Determine if the second compensation
capacitor (C6) is required. It is required if the
ESR zero of the output capacitor is located at
less than half of the switching frequency, or the
following relationship is valid:
fS
2
1
1
fP3
=
<
2π × C6 × R3
2π × C2× RESR
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
where the feedback loop has the unity gain is
important.
If this is the case, then add the second
compensation capacitor (C6) to set the pole fP3
at the location of the ESR zero. Determine the
C6 value by the equation:
C2 × RESR
C6 =
Lower crossover frequencies result in slower
line and load transient responses, while higher
crossover frequencies could cause system
unstable. A good rule of thumb is to set the
crossover frequency to below one-tenth of the
R3
External Bootstrap Diode
An external bootstrap diode may enhance the
efficiency of the regulator, the applicable
conditions of external BST diode are:
switching
frequency.
To
optimize
the
compensation components, the following
procedure can be used:
z
z
VOUT=5V or 3.3V; and
Duty cycle is high: D=
VOUT
VIN
>65%
1. Choose the compensation resistor (R3) to set
the desired crossover frequency. Determine the
R3 value by the following equation:
In these cases, an external BST diode is
recommended from the output of the voltage
regulator to BST pin, as shown in Figure2
2π × C2× fC VOUT
R3 =
×
External BST Diode
IN4148
GEA × GCS
VFB
BST
Where fC is the desired crossover frequency,
which is typically less than one tenth of the
switching frequency.
CBST
MP2361
5V or 3.3V
SW
+
COUT
L
2. Choose the compensation capacitor (C3) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, to below one forth
of the crossover frequency provides sufficient
phase margin. Determine the C3 value by the
following equation:
Figure 2—Add Optional External Bootstrap
Diode to Enhance Efficiency
The recommended external BST diode is
IN4148, and the BST cap is 0.1~1µF.
2
C3 >
π × R3 × fC
Where R3 is the compensation resistor value.
MP2361 Rev. 1.4
9/22/2011
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9
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
PCB Layout Guide
PCB layout is very important to achieve stable operation. Please follow these guidelines and take
Figure3 for references (QFN10,MSOP10, MSOP10E).
1) Keep the path of switching current short and minimize the loop area formed by Input cap, high-side
MOSFET and schottky diode.
2) Keep the connection of schottky diode between SW pin and input power ground as short and wide
as possible.
3) Ensure all feedback connections are short and direct. Place the feedback resistors and
compensation components as close to the chip as possible.
4) Route SW away from sensitive analog areas such as FB.
5) Connect IN, SW, and especially GND respectively to a large copper area to cool the chip to
improve thermal performance and long-term reliability. For single layer, do not solder exposed pad
of the IC.
SGND
SGND
R3
C8
R1
L1
C1
C2
D1
Vin
Vo
PGND
Bottom Layer
Top Layer
Figure3―PCB Layout
MP2361 Rev. 1.4
9/22/2011
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10
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
PACKAGE INFORMATION
QFN10 (3mm x 3mm)
2.90
3.10
0.30
0.50
1.45
1.75
PIN 1 ID
SEE DETAIL A
PIN 1 ID
MARKING
0.18
0.30
10
1
5
2.25
2.55
2.90
3.10
PIN 1 ID
INDEX AREA
0.50
BSC
6
TOP VIEW
BOTTOM VIEW
PIN 1 ID OPTION A
R0.20 TYP.
PIN 1 ID OPTION B
R0.20 TYP.
0.80
1.00
0.20 REF
0.00
0.05
SIDE VIEW
DETAIL A
NOTE:
2.90
1) ALL DIMENSIONS ARE IN MILLIMETERS.
0.70
1.70
2) EXPOSED PADDLE SIZE DOES NOT INCLUDE MOLD FLASH.
3) LEAD COPLANARITY SHALL BE 0.10 MILLIMETER MAX.
4) DRAWING CONFORMS TO JEDEC MO-229, VARIATION VEED-5.
5) DRAWING IS NOT TO SCALE.
0.25
0.50
2.50
RECOMMENDED LAND PATTERN
MP2361 Rev. 1.4
9/22/2011
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2011 MPS. All Rights Reserved.
11
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
MSOP10
0.114(2.90)
0.122(3.10)
6
10
0.187(4.75)
0.199(5.05)
0.114(2.90)
0.122(3.10)
PIN 1 ID
(NOTE 5)
5
1
0.007(0.18)
0.011(0.28)
0.0197(0.50)BSC
BOTTOM VIEW
TOP VIEW
GAUGE PLANE
0.010(0.25)
0.030(0.75)
0.037(0.95)
0.043(1.10)MAX
SEATING PLANE
0.004(0.10)
0.008(0.20)
0.002(0.05)
0.006(0.15)
0.016(0.40)
0.026(0.65)
0o-6o
FRONT VIEW
SIDE VIEW
NOTE:
1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN BRACKET IS
IN MILLIMETERS.
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH,
PROTRUSION OR GATE BURR.
0.181(4.60)
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH OR
PROTRUSION.
4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING)
SHALL BE 0.004" INCHES MAX.
5) PIN 1 IDENTIFICATION HAS THE HALF OR FULL CIRCLE OPTION.
6) DRAWING MEETS JEDEC MO-817, VARIATION BA.
7) DRAWING IS NOT TO SCALE.
0.040(1.00)
0.012(0.30)
0.0197(0.50)BSC
RECOMMENDED LAND PATTERN
MP2361 Rev. 1.4
9/22/2011
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2011 MPS. All Rights Reserved.
12
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
MSOP10E
MP2361 Rev. 1.4
9/22/2011
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
13
Preliminary Specifications Subject to Change
© 2011 MPS. All Rights Reserved.
MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER
SOIC8
0.189(4.80)
0.197(5.00)
0.050(1.27)
0.024(0.61)
8
5
0.063(1.60)
0.150(3.80)
0.157(4.00)
0.228(5.80)
0.244(6.20)
0.213(5.40)
PIN 1 ID
1
4
TOP VIEW
RECOMMENDED LAND PATTERN
0.053(1.35)
0.069(1.75)
SEATING PLANE
0.004(0.10)
0.010(0.25)
0.0075(0.19)
0.0098(0.25)
0.013(0.33)
0.020(0.51)
SEE DETAIL "A"
0.050(1.27)
BSC
SIDE VIEW
FRONT VIEW
0.010(0.25)
0.020(0.50)
x 45o
NOTE:
1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN
BRACKET IS IN MILLIMETERS.
GAUGE PLANE
0.010(0.25) BSC
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH,
PROTRUSIONS OR GATE BURRS.
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH
OR PROTRUSIONS.
4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING)
SHALL BE 0.004" INCHES MAX.
0.016(0.41)
0.050(1.27)
0o-8o
5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION AA.
6) DRAWING IS NOT TO SCALE.
DETAIL "A"
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.
Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS
products into any application. MPS will not assume any legal responsibility for any said applications.
MP2361 Rev. 1.4
9/22/2011
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2011 MPS. All Rights Reserved.
14
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