MP2108 [MPS]
2A, 6V, 720KHz Synchronous Buck Converter; 2A ， 6V ， 720kHz的同步降压转换器
| 型号： | MP2108 |
| 厂家： | 
MONOLITHIC POWER SYSTEMS |
| 描述： | 2A, 6V, 720KHz Synchronous Buck Converter |
| 文件： | 总10页 (文件大小：253K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
TM
MP2108
2A, 6V, 720KHz
Synchronous Buck Converter
TM
The Future of Analog IC Technology
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
DESCRIPTION
FEATURES
The MP2108 is a 2A, 720KHz synchronous
buck converter designed for low voltage
applications requiring high efficiency. It is
capable of providing output voltages as low as
0.9V, and integrates top and bottom switches to
minimize power loss and component count. The
720KHz switching frequency allows for small
filtering components, further reducing the
solution size.
•
•
•
•
•
•
•
•
•
•
•
2A Output Current
Synchronous Rectification
Internal 160mΩ and 190mΩ Power Switches
Input Range of 2.6V to 6V
Over 90% Efficiency
Zero Current Shutdown Mode
Under Voltage Lockout Protection
Soft-Start Operation
Thermal Shutdown
Internal Current Limit (Source & Sink)
Tiny 10-Pin MSOP and 3x3 QFN Packages
The MP2108 includes cycle-by-cycle current
limiting and under voltage lockout. Internal
power switches, combined with the tiny 10-pin
MSOP or 3mm x 3mm QFN packages, provide
a solution requiring a minimum of board space.
APPLICATIONS
•
•
•
•
SOHO Routers, PCMCIA Cards, Mini PCI
Handheld Computers, PDAs
Cell phones, Digital Still and Video Cameras
Small LCD Displays
EVALUATION BOARD REFERENCE
Board Number
Dimensions
EV2108DQ/DK-00A
2.5”X x 2.0”Y x 0.5”Z
“MPS” and “The Future of Analog IC Technology” are Trademarks of Monolithic
Power Systems, Inc.
TYPICAL APPLICATION
10nF
INPUT
2.6V to 6V
2
1
BST
VIN
10
6
3
7
OUTPUT
1.8V / 2A
RUN
SS
LX
OFF ON
MP2108
8
COMP
FB
VREF SGND PGND
10nF
3.3nF
9
5
4
10nF
MP2108_TAC_S01
MP2108 Rev. 0.93
2/28/2006
www.MonolithicPower.com
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© 2006 MPS. All Rights Reserved.
1
TM
MP2108 – 2A, 6V, 720KHz SYNCHRONOUS BUCK CONVERTER
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
PACKAGE REFERENCE
TOP VIEW
TOP VIEW
BST
VIN
1
2
3
4
5
10 RUN
BST
VIN
1
2
3
4
5
10
9
RUN
VREF
COMP
FB
9
8
7
6
VREF
COMP
FB
LX
LX
8
PGND
SGND
PGND
SGND
7
6
SS
SS
MP2108_PD01-MSOP10
EXPOSED PAD
ON BACKSIDE
MP2108_PD02-QFN10
Part Number**
Package
Temperature
–40°C to +85°C
Part Number*
Package
Temperature
QFN10
(3mm x 3mm)
MP2108DQ
MP2108DK
MSOP10
–40°C to +85°C
** For Tape & Reel, add suffix –Z (eg. MP2108DQ–Z)
For Tape & Reel, add suffix –Z (eg. MP2108DK–Z)
For Lead Free, add suffix –LF (eg. MP2108DK–LF–Z)
*
For Lead Free, add suffix –LF (eg. MP2108DQ–LF–Z)
ABSOLUTE MAXIMUM RATINGS (1)
Input Supply Voltage VIN ............................. 6.5V
LX Voltage VLX ..................... –0.3V to VIN + 0.3V
BST to LX Voltage .........................–0.3V to +6V
Voltage on All Other Pins...............–0.3V to +6V
Storage Temperature...............–55°C to +150°C
Recommended Operating Conditions (2)
Input Supply Voltage VIN ...................... 2.6V to 6
Output Voltage VOUT...........................0.9V to 5V
Operating Temperature..............–40°C to +85°C
Thermal Resistance (3)
MSOP10................................150..... 65... °C/W
QFN10 (3mm x 3mm).............50...... 12... °C/W
θJA
θJC
Notes:
1) Exceeding these ratings may damage the device.
2) The device is not guaranteed to function outside of its
operating conditions.
3) Measured on approximately 1” square of 1 oz copper.
ELECTRICAL CHARACTERISTICS
VIN = 5.0V, TA = +25°C, unless otherwise noted.
Parameter
Symbol Condition
Min
Typ
Max
Units
Input Voltage Range
Input Undervoltage Lockout
VIN
2.6
6
V
V
2.2
Input Undervoltage Lockout
Hysteresis
100
mV
Shutdown Supply Current
Operating Supply Current
VREF Voltage
RUN Input Low Voltage
RUN Input High Voltage
RUN Hysteresis
VRUN ≤ 0.3V
VRUN > 2V, VFB = 1.1V
VIN = 2.6V to 6V
0.5
1.2
2.4
1.0
1.8
µA
mA
V
V
V
VREF
VIL
VHL
0.4
1
1.5
100
mV
µA
RUN Input Bias Current
MP2108 Rev. 0.93
2/28/2006
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2
TM
MP2108 – 2A, 6V, 720KHz SYNCHRONOUS BUCK CONVERTER
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
ELECTRICAL CHARACTERISTICS (continued)
VIN = 5.0V, TA = +25°C, unless otherwise noted.
Parameter
Symbol Condition
Min
Typ
720
200
Max
Units
Oscillator
Switching Frequency
Maximum Duty Cycle
Minimum On Time
Error Amplifier
fSW
620
85
920
KHz
%
ns
DMAX VFB = 0.7V
TON
Voltage Gain
AVEA
GEA
400
300
±30
900
–100
V/V
µA/V
µA
mV
nA
Transconductance
COMP Maximum Output Current
FB Regulation Voltage
FB Input Bias Current
Soft-Start
VFB
IFB
880
920
VFB = 0.9V
Soft-Start Current
Output Switch On-Resistance
ISS
2
µA
VIN = 5V
VIN = 3V
VIN = 5V
VIN = 3V
190
280
160
230
3.5
mΩ
mΩ
mΩ
mΩ
A
Switch On Resistance
Synchronous Rectifier On Resistance
Switch Current Limit (Source)
2.5
Synchronous Rectifier Current Limit
(Sink)
350
160
mA
Thermal Shutdown
°C
PIN FUNCTIONS
Pin #
Name
Description
Power Switch Boost. BST powers the gate of the high-side N-Channel power MOSFET
switch. Connect a 10nF or greater capacitor between BST and LX.
1
BST
Internal Power Input. VIN supplies the power to the MP2108 through the internal LDO
regulator. Bypass VIN to PGND with a 10µF or greater capacitor. Connect VIN to the input
source voltage.
Output Switching Node. LX is the source of the high-side N-Channel switch and the drain
of the low-side N-Channel switch. Connect the output LC filter between LX and the output.
2
3
VIN
LX
Power Ground. PGND is the source of the N-Channel MOSFET synchronous rectifier.
Connect PGND to SGND as close to the MP2108 as possible.
SGND Signal Ground.
4
5
PGND
Soft-Start Input. Place a capacitor from SS to SGND to set the soft-start period. The
6
SS
MP2108 sources 2µA from SS to the soft-start capacitor at start up. As the voltage at SS
rises, the feedback threshold voltage increases to limit inrush current at startup.
Feedback Input. FB is the inverting input of the internal error amplifier. Connect a resistive
voltage divider from the output voltage to FB to set the output voltage.
Compensation Node. COMP is the output of the error amplifier. Connect a series RC
network to compensate the regulation control loop.
Internal 2.4V Regulator Bypass. Connect a 10nF capacitor between VREF and SGND to
bypass the internal regulator. Do not apply any load to VREF.
On/Off Control Input. Drive RUN high to turn on the MP2108, drive RUN low to turn the
MP2108 off. For automatic startup, connect RUN to VIN via a pullup resistor.
7
8
FB
COMP
VREF
RUN
9
10
MP2108 Rev. 0.93
2/28/2006
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TM
MP2108 – 2A, 6V, 720KHz SYNCHRONOUS BUCK CONVERTER
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
OPERATION
V
IN
2.6V to 6V
V
IN
2
C1
CURRENT
SENSE
AMPLIFIER
ENABLE
CKT & LDO
REGULATOR
GATE
DRIVE
REGULATOR
RUN
OFF ON
Vdr
+
--
10
BST
LX
V
REF
Vdr
2.4V
PWM
1
3
COMPARATOR
9
C6
C7
+
--
L1
V
OUT
CONTROL
LOGIC
Vdr
C2
720KHz
OSCILLATOR
RAMP
CURRENT
+
LIMIT
--
COMPARATOR
V
BP
R2
UVLO &
THERMAL
SHUTDOWN
+
--
PGND
FB
SS
4
7
6
C5
--
--
+
GM
ERROR
AMPLIFIER
V
FB
0.9V
CURRENT
LIMIT
THRESHOLD
R1
5
8
COMP
R3
C3
SGND
C4
MP2108_BD01
Figure 1—Functional Block Diagram
MP2108 Rev. 0.93
2/28/2006
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TM
MP2108 – 2A, 6V, 720KHz SYNCHRONOUS BUCK CONVERTER
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
The MP2108 measures the output voltage
through an external resistive voltage divider and
compares it to the internal 0.9V reference to
generate the error voltage at COMP. The
current-mode regulator uses the voltage at
COMP and compares it to the inductor current
to regulate the output voltage. The use of
current-mode regulation improves transient
response and control loop stability.
The average inductor current is controlled by
the voltage at COMP, which in turn, is
controlled by the output voltage. Thus the
output voltage controls the inductor current to
satisfy the load.
Since the high-side N-Channel MOSFET
requires voltage above VIN to drive its gate, a
bootstrap capacitor from LX to BST is required
to drive the high-side MOSFET gate. When LX
is driven low (through the low-side MOSFET),
the BST capacitor is internally charged. The
voltage at BST is applied to the high-side
MOSFET gate to turn it on. Voltage is
maintained until the high-side MOSFET is
turned off and the low-side MOSFET is turned
on, and the cycle repeats. Connect a 10nF or
greater capacitor from BST to SW to drive the
high-side MOSFET gate.
At the beginning of each cycle, the high-side
N-Channel MOSFET is turned on, forcing the
inductor current to rise. The current at the drain
of the high-side MOSFET is internally
measured and converted to a voltage by the
current sense amplifier.
That voltage is compared to the error voltage at
COMP. When the inductor current rises
sufficiently, the PWM comparator turns off the
high-side switch and turns on the low-side
switch, forcing the inductor current to decrease.
APPLICATION INFORMATION
C7
10nF
INPUT
2.6V to 6V
2
1
VIN
BST
10
3
7
OUTPUT
1.8V / 2A
RUN
SS
LX
FB
6
8
MP2108
COMP
VREF SGND PGND
C5
10nF
C3
3.3nF
9
5
4
C4
OPEN
C6
10nF
MP2108_TAC_F02
Figure 2—Typical Application Circuit
Internal Low-Dropout Regulator
Soft-Start
The internal power to the MP2108 is supplied
from the input voltage (VIN) through an internal
2.4V low-dropout linear regulator, whose output
is VREF. Bypass VREF to SGND with a 10nF
or greater capacitor for proper operation. The
internal regulator can not supply more current
than is required to operate the MP2108.
Therefore, do not apply any external load to
VREF.
The MP2108 includes a soft-start timer that
slowly ramps the output voltage at startup to
prevent excessive current at the input.
When power is applied to the MP2108, and
RUN is asserted, a 2µA internal current source
charges the external capacitor at SS. As the
capacitor charges, the voltage at SS rises. The
MP2108 internally limits the feedback threshold
voltage at FB to that of the voltage at SS. This
forces the output voltage to rise at the same
MP2108 Rev. 0.93
2/28/2006
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TM
MP2108 – 2A, 6V, 720KHz SYNCHRONOUS BUCK CONVERTER
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
rate as the voltage at SS, forcing a linear
output voltage ramp from 0V to the desired
regulation voltage during soft-start.
If using this technique, it is recommended that
the larger capacitor be a tantalum or electrolytic
type. All ceramic capacitors should be placed
close to the MP2108. For most applications, a
10µF ceramic capacitor will work.
The soft-start period is determined by the
equation:
Selecting the Output Capacitor
tSS = 0.45 × C5
The output capacitor (C2) is required to
maintain the DC output voltage. Low ESR
capacitors are preferred to keep the output
voltage ripple to a minimum. The characteristics
of the output capacitor also affect the stability of
the regulation control system. Ceramic,
tantalum, or low ESR electrolytic capacitors are
recommended.
Where C5 (in nF) is the soft-start capacitor from
SS to GND, and tSS (in ms) is the soft-start
period. Determine the capacitor required for a
given soft-start period by the equation:
C5 = 2.22× tSS
Use values for C5 between 10nF and 22nF to
set the soft-start period (between 4ms and
10ms).
The output voltage ripple is:
VRIPPLE
=
Setting the Output Voltage (see Figure 2)
Set the output voltage by selecting the resistive
voltage divider ratio. The voltage divider drops
the output voltage to the 0.9V feedback voltage.
Use 10kꢀ for the low-side resistor of the
voltage divider. Determine the high-side resistor
by the equation:
⎛
⎞
⎟
⎟
⎠
⎛
⎜
⎝
⎞
⎟
⎟
⎠
VOUT
VOUT
VIN
1
⎜
⎜
× 1−
× RESR
+
⎜
fSW × L
8 × fSW × C2
⎝
Where VRIPPLE is the output voltage ripple, fSW is
the switching frequency, VIN is the input voltage
and RESR is the equivalent series resistance of
the output capacitors.
V
⎛
⎞
OUT
Choose an output capacitor to satisfy the output
ripple requirements of the design. A 22µF
ceramic capacitor is suitable for most
applications.
⎜
⎜
⎟
R2 =
− 1 × R1
⎟
⎠
0.9V
⎝
Where R2 is the high-side resistor, VOUT is the
output voltage and R1 is the low-side resistor
and VOUT is the output voltage.
Selecting the Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor results in less ripple current that in turn
results in lower output ripple voltage. However,
the larger value inductor is likely to have a
larger physical size and higher series
resistance. Choose an inductor that does not
saturate under the worst-case load conditions.
A good rule for determining the inductance is to
allow peak-to-peak ripple current to be
approximately 30% to 40% of the maximum
load current. Make sure that the peak inductor
current (the load current plus half the peak-to-
peak inductor ripple current) is below 2.5A to
prevent loss of regulation due to the current
limit.
Selecting the Input Capacitor
The input current to the step-down converter is
discontinuous, so a capacitor is required to
supply the AC current to the step-down
converter while maintaining the DC input
voltage. A low ESR capacitor is required to
keep the noise at the IC to a minimum. Ceramic
capacitors are preferred, but tantalum or low
ESR electrolytic capacitors are also an option..
The capacitor can be electrolytic, tantalum or
ceramic. Because it absorbs the input switching
current it must have an adequate ripple current
rating. Use a capacitor with RMS current rating
greater than 1/2 of the DC load current.
For stable operation, place the input capacitor
as close to the IC as possible. A smaller high
quality 0.1µF ceramic capacitor may be placed
closer to the IC with the larger capacitor placed
further away.
MP2108 Rev. 0.93
2/28/2006
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TM
MP2108 – 2A, 6V, 720KHz SYNCHRONOUS BUCK CONVERTER
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
Calculate the required inductance value by the
equation:
If large value capacitors with relatively high
equivalent-series-resistance (ESR) are used,
the zero due to the capacitance and ESR of the
output capacitor can be compensated by a third
pole set by R3 and C4. The pole is:
VOUT
×
(
VIN − VOUT
)
L =
VIN × fSW × ∆I
Where ∆I is the peak-to-peak inductor ripple
current. It is recommended to choose ∆I to be
30%~40% of the maximum load current.
1
fP3
=
2π ×R3× C4
The system crossover frequency (the frequency
where the loop gain drops to 1dB or 0dB) is
important. Set the crossover frequency below
one tenth of the switching frequency to insure
stable operation. Lower crossover frequencies
result in slower response and worse transient
load recovery. Higher crossover frequencies
degrade the phase and/or gain margins and
can result in instability.
Compensation
The system stability is controlled through the
COMP pin. COMP is the output of the internal
transconductance error amplifier. A series
capacitor-resistor combination sets a pole-zero
combination to control the characteristics of the
control system.
The DC loop gain is:
Table 1—Compensation Values for Typical
Output Voltage/Capacitor Combinations
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
VFB
AVDC
=
× AVEA × GCS ×RLOAD
VOUT
VOUT
C2
R3
C3
C4
Where VFB is the feedback voltage, 0.9V, AVEA
is the transconductance error amplifier voltage
gain, 400 V/V and GCS is the current sense
transconductance, (roughly the output current
divided by the voltage at COMP), 4.5A/V.
1.8V 22µF Ceramic
2.5V 22µF Ceramic
3.3V 22µF Ceramic
6.8kꢀ
9.1kꢀ
12kꢀ
3.3nF None
2.2nF None
1.8nF None
47µF Tantalum
1.8V
13kꢀ
18kꢀ
24kꢀ
2nF
1nF
(300mꢀ)
RLOAD is the load resistance:
VOUT
47µF Tantalum
2.5V
1.2nF 750pF
1nF 560pF
RLOAD
=
(300mꢀ)
IOUT
47µF Tantalum
3.3V
(300mꢀ)
Where IOUT is the output load current.
The system has 2 poles of importance, one is
due to the compensation capacitor (C3), and
the other is due to the load resistance and the
output capacitor (C2), where:
Choosing the Compensation Components
The values of the compensation components
given in Table 1 yields a stable control loop for
the output voltage and capacitor given. To
optimize the compensation components for
conditions not listed in Table 1, use the
following procedure.
GEA
fP1
=
2π × AVEA × C3
P1 is the first pole, and GEA is the error amplifier
transconductance (300µA/V) and
1
fP2
=
2π×RLOAD × C2
The system has one zero of importance, due to
the compensation capacitor (C3) and the
compensation resistor (R3). The zero is:
1
fZ1
=
2π ×R3× C3
MP2108 Rev. 0.93
2/28/2006
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TM
MP2108 – 2A, 6V, 720KHz SYNCHRONOUS BUCK CONVERTER
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
Choose the compensation resistor to set the
desired crossover frequency. Determine the
value by the following equation:
If this is the case, then add the second
compensation capacitor.
Determine the value by the equation:
2π × C2× fC VOUT
C2×RESR(max)
R3 =
×
C4 =
GEA × GCS
VFB
R3
Where GEA is the EA transconductance
(300µA/V), and fC is the desired crossover
frequency (preferably 33KHz).
Where RESR(MAX) is the maximum ESR of the
output capacitor.
External Boost Diode
Choose the compensation capacitor to set the
zero below one fourth of the crossover
frequency. Determine the value by the following
equation:
It is recommended that an external boost diode
be added to help improve the regulator
efficiency. The diode can be a low cost diode
such as an IN4148 or BAT54.
2
5V
C3 >
π × R3 × fC
BOOST
DIODE
Determine if the second compensation
capacitor, C4 is required. It is required if the
ESR zero of the output capacitor happens at
less than half of the switching frequency. Or:
1
BST
10nF
MP2108
3
LX
MP2108_F03
π × C2× RESR × fSW > 1
Figure 3—External Boost Diode
where RESR is the equivalent series resistance
of the output capacitor.
This diode is also recommended for high duty
VOUT
cycle operation (when
>65%).
VIN
MP2108 Rev. 0.93
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TM
MP2108 – 2A, 6V, 720KHz SYNCHRONOUS BUCK CONVERTER
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
PACKAGE INFORMATION
MSOP10
MP2108 Rev. 0.93
2/28/2006
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TM
MP2108 – 2A, 6V, 720KHz SYNCHRONOUS BUCK CONVERTER
INITIAL RELEASE – SPECIFICATIONS SUBJECT TO CHANGE
QFN10 (3mm x 3mm)
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.
Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS
products into any application. MPS will not assume any legal responsibility for any said applications.
MP2108 Rev. 0.93
2/28/2006
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10
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