MAX16812ATI+_技术文档

19-0880; Rev 0; 7/07

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

General Description

Features

♦ Integrated 76V, 0.2Ω (typ) Power MOSFET

♦ 5.5V to 76V Wide Input Range

The MAX16812 is a peak-current-mode LED driver with

an integrated 0.2Ω power MOSFET designed to control

the current in a single string of high-brightness LEDs

(HBLEDs). The MAX16812 can be used in multiple con-

verter topologies such as buck, boost, or buck-boost.

The MAX16812 operates over a 5.5V to 76V wide sup-

ply voltage range.

♦ Adjustable LED Current with 5% Accuracy

♦ Floating Differential LED Current-Sense Amplifier

♦ Floating Dimming N-Channel MOSFET Driver

♦ PWM LED Dimming with:

PWM Control Signal

The MAX16812 features a low-frequency, wide-range

brightness adjustment (100:1), analog and PWM dim-

ming control input, as well as a resistor-programmable

EMI suppression circuitry to control the rise and fall

times of the internal switching MOSFET. A high-side

LED current-sense amplifier and a dimming MOSFET

driver are also included, simplifying the design and

reducing the total component count.

Analog Control Signal

Chopped V Input

IN

♦ Peak-Current-Mode Control

♦ 125kHz to 500kHz Adjustable Switching Frequency

♦ Adjustable UVLO and Soft-Start

♦ Output Overvoltage Protection

♦ 5µs LED Current Rise/Fall Times During Dimming

The MAX16812 uses peak-current-mode control,

adjustable slope compensation that allows for addition-

al design flexibility. The device has two current regula-

tion loops. The first loop controls the internal switching

MOSFET peak current, while the second current regula-

tion loop controls the LED current. Switching frequency

can be adjusted from 125kHz to 500kHz.

Minimize EMI

♦ Overtemperature and Short-Circuit Protection

Ordering Information

PIN-

PACKAGE

PKG

CODE

PART

TEMP RANGE

Additional features include adjustable UVLO, soft-start,

external enable/disable input, thermal shutdown, a

1.238V 1% accurate buffered reference, and an on-

chip oscillator. An internal 5.2V linear regulator supplies

up to 20mA to power external devices.

MAX16812ATI+ -40°C to +125°C 28 TQFN-EP* T2855-8

+Denotes a lead-free package.

*EP = Exposed pad.

Simplified Diagram

The MAX16812 is available in a thermally enhanced

5mm x 5mm, 28-pin TQFN-EP package and is specified

over the automotive -40°C to +125°C temperature range.

C

H_REG

DOUT

R

CS

VOUT

Applications

Automotive Lighting:

C

OUT

R

SRC

DRL, Fog Lights

LV

IN

SRC

GT

V

IN

Rear Combination Lights

Front and Rear Signal Lights

Interior Lighting

C

IN

EN

RT

Warning and Emergency Lighting

RT

DRV

SLP

MAX16812

Architectural and Industrial Lighting

R

TGRM

L_REG

TGRM

DIM

C

SLP

C

TGRM

COMP

R

C

COMP1

OV1

VOUT

OV2

R

COMP1

R

COMP2

Typical Application Circuit and Pin Configuration appear at

end of data sheet.

BUCK-BOOST CONFIGURATION

________________________________________________________________ Maxim Integrated Products

1

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,

or visit Maxim’s website at www.maxim-ic.com.

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

ABSOLUTE MAXIMUM RATINGS

(All voltages are referenced to AGND, unless otherwise noted.)

SGND ....................................................................-0.3V to +0.3V

IN, EN, LX, DIM ......................................................-0.3V to +80V

L_REG, GT, DRV ......................................................-0.3V to +6V

RT, REF, REFI, CS_OUT, FB, COMP, SRC,

DD to LV ....................................................................-1V to +80V

Maximum Current into Any Pin (except LX, SRC) ............ 20mA

Maximum Current into LX and SRC.......................................+2A

Continuous Power Dissipation (T = +70°C)

A

28-Pin TQFN 5mm x 5mm

SLP, TGRM, OV....................................................-0.3V to +6V

LV, HV, CS-, CS+, DGT, DD, H_REG ....................-0.3V to +80V

CS+, DGT, H_REG to LV........................................-0.3V to +12V

CS- to LV ...............................................................-0.3V to +0.3V

CS+ to CS- .............................................................-0.3V to +12V

(derate 34.65mW/°C* above +70°C) .........................2759mW

Operating Temperature Range .........................-40°C to +125°C

Junction Temperature......................................................+150°C

Storage Temperature Range.............................-65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

MAX6812

*As per JEDEC51 standard (multilayer board).

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional

operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to

absolute maximum rating conditions for extended periods may affect device reliability.

ELECTRICAL CHARACTERISTICS

(V = V = 12V, C

= 3.3µF, C

= 1µF, C

= 47nF, V

= 0V, R

= 0.2Ω, T = T = -40°C to +125°C, unless oth-

IN

EN

L_REG

H_REG

REF

TGRM

SRC

A

J

erwise noted. Typical values are at T = +25°C.)

A

PARAMETER

Input Voltage Range

SYMBOL

CONDITIONS

MIN

5.5

TYP

MAX

76.0

2.5

45

UNITS

V

IN

Q

Quiescent Supply

I

V

= 1V, V

= 0V

DIM

0.3

mA

µA

Ω

TGRM

Shutdown Supply Current

Internal MOSFET On-Resistance

Output Current Accuracy

Peak Switch Current Limit

Hiccup Switch Current

Switch Leakage Current

UNDERVOLTAGE LOCKOUT

IN Undervoltage Lockout

UVLO Hysteresis

I

≤ 300mV

EN

20

SHDN

R

I

= 1A, V > 10V, V = V = 5V

DRV

0.2

0.4

+5

DSON

LX

IN

GT

I

= 350mA, R = 1Ω

-5

%

LED

CS

I

2.6

3.1

6

3.6

A

LXLIM

A

I

V

= 0V, V = 76V, V = 0V

1

10

5.3

1.6

µA

LXLEAK

UVLO

EN

IN

LX

GT

rising

4.6

1.2

4.9

100

1.38

100

V

mV

V

EN Threshold Voltage

EN Hysteresis

V

EN_THUP

EN

mV

REFERENCE (REF) AND LOW-SIDE LINEAR REGULATOR (L_REG)

Startup Response Time

Reference Voltage

t

V

or V rising

50

µs

V

POR

IN

EN

V

I

= 10µA

1.190

25

1.238

1.288

60

REF

Reference Soft-Start Charging

Current

I

V

= 0V

40

µA

REF_SLEW

REF

L_REG Supply Voltage

L_REG Load Regulation

L_REG Dropout Voltage

V

= 7.5V, I

= 1mA

4.9

5.2

5.5

20

V

Ω

IN

L_REG

I

= 20mA

L_REG

= 25mA

400

mV

2

_______________________________________________________________________________________

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

ELECTRICAL CHARACTERISTICS (continued)

(V = V = 12V, C

= 3.3µF, C

= 1µF, C

= 47nF, V

= 0V, R

= 0.2Ω, T = T = -40°C to +125°C, unless oth-

IN

EN

L_REG

H_REG

REF

TGRM

SRC

A

J

erwise noted. Typical values are at T = +25°C.)

A

PARAMETER

SYMBOL

CONDITIONS

MIN

TYP

MAX

UNITS

PWM COMPARATOR

V

= 1V, V

= 0V, V

= 0.5V, V

OS

= 1V,

COMP

SRC

TGRM

COMP Input Leakage Current

SRC Input Leakage Current

I

-0.10

-5

+0.10

+5

µA

LKCOMP

= 0.5V

DIM

V

= 0V,

COMP

SRC

I

LKSRC

= 0.5V

DIM

Comparator Offset Voltage

Input Voltage Range

Propagation Delay

ERROR AMPLIFIER

FB Input Current

V

(V

- V ) = V

SRC

860

100

mV

V

OS(EA)

COMP

V

= V

+ 860mV

0

1.23

SRC

COMP

SRC

t

50mV overdrive

ns

PD

V

= 1V, V

= 1.2V

= 1V

-100

-23

0

+100

+23

nA

mV

V

FB

REFI

REFI Input Current

Error-Amplifier Offset Voltage

Input Common-Mode Range

Source Current

V

= V

= 1.2V

- 0.9V)

OS

COMP

= (V

1.5

COMP

I

(V

- V ) ≥ 0.5V

300

80

µA

V

COMP

REFI

FB

Sink Current

(V - V

) ≥ 0.5V

FB

REFI

COMP Clamp Voltage

DC Gain

V

= 1.2V, V = 0V

1.20

2.56

COMP

REF

FB

72

dB

MHz

Unity-Gain Bandwidth

0.8

ELECTRICAL CHARACTERISTICS

(V = V = 12V, C

= 3.3µF, C

= 1µF, C

= 47nF, V

= 0V, R = 0.2Ω, R = 1Ω, T = T = -40°C to +125°C,

SRC CS A J

IN

EN

L_REG

H_REG

REF

TGRM

unless otherwise noted. Typical values are at T = +25°C.)

A

PARAMETER

SYMBOL

CONDITIONS

MIN

TYP

MAX

UNITS

HIGH-SIDE UNDERVOLTAGE LOCKOUT AND LINEAR REGULATOR (H_REG) ((V - V ) = 21V)

HV

LV

H_REG Input-Voltage Threshold

H_REG Supply Voltage

V

is rising

= 0

3.60

4.75

3.887

5

4.20

5.40

80

V

H_REG

I

H_REG

H_REG Load Regulation

Dropout Voltage

I

= 0 to 3mA

= 5mA

Ω

H_REG

820

mV

HIGH-SIDE CURRENT-SENSE AMPLIFIERS (V - V ) = 21V

HV

LV

CS- Input Bias Current

CS+ Input Bias Current

Input Voltage Range

I

V

= V , (V

- V ) = -0.1V

500

+1

µA

V

CS-

LV

CS+

CS-

I

= V , (V

- V ) = 0.1V

-1

0

CS+

LV

CS-

= V

0.25

LV

Sinking

Sourcing

25

400

0

Minimum Output Current

I

µA

CS_OUT

Output Voltage Range

DC Voltage Gain

V

1.5

1.0

V

V/V

MHz

V

CS_OUT

4

Unity-Gain Bandwidth

Maximum REFI Input Voltage

0.8

V

REFI

_______________________________________________________________________________________

3

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

ELECTRICAL CHARACTERISTICS (continued)

(V = V = 12V, C

= 3.3µF, C

= 1µF, C

= 47nF, V

= 0V, R = 0.2Ω, R = 1Ω, T = T = -40°C to +125°C,

SRC CS A J

IN

EN

L_REG

H_REG

REF

TGRM

unless otherwise noted. Typical values are at T = +25°C.)

A

PARAMETER

SYMBOL

CONDITIONS

MIN

TYP

MAX

UNITS

HIGH-SIDE DIMMING LINEAR REGULATOR ((V - V ) = 21V)

HV

LV

V

= V , (V

- V ) = 0.3V,

CS+ CS-

LV

CS-

(V

- V ) = 1V, V

= 1V, V

TGRM

= 0V,

= 3V,

1.2

DD

LV

DIM

V

= 1V, V

= 1.0V, sinking

DGT

REFI

Minimum Output Current

Output Voltage Range

I

mA

V

DGT

MAX6812

V

= V , (V - V ) = 0.2V,

CS+ CS-

LV

CS-

(V

DD

- V ) = 1V, V

= 0V, V

DGT

LV

TGRM

V

= 1.0V, V

= 1V, sourcing

DIM

REFI

0.2

5.0

DC Gain

C

= 1nF to LV

60

dB

µA

DGT

DD Input Bias Current

I

(V

- V -) = 0.5V

-3

+3

DD

CS

V

(V

= 0V, V

= 1V, V

= 1.2V,

REFI

TGRM

DIM

DD Input Low Threshold

0.25

0.50

0.75

V

- V

LV)

> 1.5V, V

falling

DD

DGT

DIMMING ((V - V ) = 21V)

HV

LV

DIM Input Bias Current

I

V

= 1.1V

DIM

-1

+1

µA

V

DIM

TGRM Input High Threshold

1.18

1.23

1

1.27

TGRM Reset High-to-TGRM Low

Pulse Width

µs

Ω

TGRM Reset Switch R

= 1.3V

TGRM

20

DS(ON)

Dimming Rise and Fall LED

Current Times

5

µs

OVERVOLTAGE PROTECTION (OV)

OV Input High Threshold

V

rising

1.180

-1

1.230

14

1.292

+1

V

OV

OV Input Threshold Hysteresis

OV Input Bias Current

mV

µA

I

= 1.1V

OV

INTERNAL OSCILLATOR CLOCK

RT = 2MΩ to AGND

RT = 50kΩ to AGND

470

105

525

125

570

155

Internal Clock Frequency

f

kHz

µA

OSC

SLOPE COMPENSATION INPUT (SLP)

SLP Input Current

I

V

= 0V

SLP

150

SLP

LOW-SIDE GATE DRIVE (DRV)

DRV Output Low Impedance

DRV Output High Impedance

INTERNAL POWER MOSFET

GT Input Leakage Current

R

DRV sinking 20mA

DRV sourcing 20mA

3

30

45

Ω

DRV_LO

R

10

DRV_HI

V

= 0 to 5V

= 50V

-1

+1

µA

V

GT

LX

Internal MOSFET Gate-to-Source

Threshold Voltage

V

2.5

8

TH

Internal MOSFET Gate Charge

Q

nC

g

4

_______________________________________________________________________________________

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

Typical Operating Characteristics

(V = V = 12V, C

= 3.3µF, C

= 1µF, V

= 0V, T = +25°C, unless otherwise noted.)

TGRM A

IN

EN

L_REG

H_REG

SWITCH CURRENT LIMIT

vs. TEMPERATURE

R

vs. I

R vs. V

DS(ON) GT

DS(ON)

LX

0.45

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0

3.300

3.250

3.200

3.150

3.100

3.050

3.000

2.950

2.900

T

= +25°C

A

0.40

0.35

0.30

0.25

0.20

0.15

0.10

0.05

0

T

= +125°C

A

T

= +25°C

A

T

= -40°C

A

1.0

1.5

2.0

(A)

2.5

3.0

2.2 2.8 3.4 4.0 4.6 5.2 5.8 6.4 7.0

(V)

-40 -25 -10

5

20 35 50 65 80 95 110 125

I

V

GT

TEMPERATURE (°C)

LX

SHUTDOWN CURRENT

vs. TEMPERATURE

IN UVLO THRESHOLD

vs. TEMPERATURE

V

vs. TEMPERATURE

REF

1.25

1.24

1.23

1.22

1.21

30

25

20

15

10

5

5.20

5.15

5.10

5.05

5.00

V

RISING

IN

I

= 10µA

REF

0

-40 -25 -10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

-40 -25 -10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

-40 -25 -10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

IN UVLO THRESHOLD

vs. TEMPERATURE

EN UVLO THRESHOLD

vs. TEMPERATURE

5.10

1.50

1.45

1.40

1.35

1.30

1.25

1.20

1.15

1.10

1.05

1.00

V

FALLING

V

RISING

IN

EN

5.09

5.08

5.07

5.06

5.05

5.04

5.03

5.02

5.01

5.00

-40 -25 -10

5

20 35 50 65 80 95 110 125

-40 -25 -10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

_______________________________________________________________________________________

5

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Typical Operating Characteristics (continued)

(V = V = 12V, C

= 3.3µF, C

= 1µF, V

= 0V, T = +25°C, unless otherwise noted.)

IN

EN

L_REG

H_REG

TGRM

A

EN UVLO THRESHOLD

vs. TEMPERATURE

OSCILLATOR FREQUENCY

vs. TEMPERATURE

V

vs. I

L_REG

600

500

400

300

200

100

0

1.50

5.5

5.4

5.3

5.2

5.1

5.0

4.9

4.8

4.7

4.6

V

FALLING

EN

R = 2MΩ

T

1.45

1.40

1.35

1.30

1.25

1.20

1.15

1.10

1.05

1.00

T

= +125°C

A

MAX6812

T

= +25°C

A

R = 180kΩ

T

A

= -40°C

R = 50kΩ

T

V

= 7.5V

IN

4.5

0

-40 -25 -10

5

20 35 50 65 80 95 110 125

-40 -25 -10

5

20 35 50 65 80 95 110 125

2

4

6

8

10 12 14 16 18 20

(mA)

TEMPERATURE (°C)

I

L_REG

V

THRESHOLD

H_REG

OSCILLATOR FREQUENCY vs. R

T

vs. TEMPERATURE

600

500

400

300

200

100

0

4.2

4.1

4.0

3.9

3.8

3.7

3.6

3.5

3.4

0.01

0.1

1

10

-40 -25 -10

5

20 35 50 65 80 95 110 125

R (MΩ)

T

TEMPERATURE (°C)

V

vs. TEMPERATURE

V

vs. I

H_REG

5.00

4.95

4.90

4.85

4.80

4.75

4.70

4.65

4.60

4.55

4.50

5.2

5.1

5.0

4.9

4.8

4.7

4.6

4.5

4.4

4.3

4.2

(V - V ) = 6V

HV

LV

(V - V ) = 21V

HV

LOAD

LV

V

= 12V

IN

I

= 3mA

V

IS MEASURED

H_REG

WITH RESPECT TO V

LV

0

0.5

1.0

1.5

(mA)

2.0

2.5

3.0

-40 -25 -10

5

20 35 50 65 80 95 110 125

I

H_REG

TEMPERATURE (°C)

6

_______________________________________________________________________________________

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

Pin Description

1

NAME

FB

FUNCTION

Low-Side Error Amplifier’s Inverting Input

2

COMP

Low-Side Error Amplifier’s Output. Connect a compensation network from COMP to FB for stable operation.

Reference Input. V

LED current.

provides the reference voltage for the high-side current-sense amplifier to set the

REFI

3

REFI

4

5

6

REF

+1.23V Reference Output. Connect an appropriate soft-start capacitor from REF to AGND.

is proportional to the current through R

CS_OUT High-Side Current-Sense Amplifier Output. V

.

CS

CS_OUT

AGND

Analog Ground

Enable Input/Undervoltage Lockout. Connect EN to IN through a resistive voltage-divider to program the

UVLO threshold. Connect EN directly to IN to set up the device for 5V internal threshold. Apply a logic-

level input to EN to enable/disable the device.

7

EN

8

IN

Positive Power-Supply Input. Bypass with a 1µF ceramic capacitor to AGND.

5V Low-Side Regulator Output. Bypass with a 3.3µF ceramic capacitor to AGND.

Signal Ground

9

L_REG

SGND

DD

10

11

12

MOSFET’s Drain Voltage-Sense Input. Connect DD to the drain of the external dimming MOSFET.

External Dimming MOSFET’s Gate Drive

DGT

High-Side Current-Sense Amplifier’s Positive Input. Connect R between CS+ and CS-. CS+ is

CS

referenced to LV.

13

14

15

16

17

18

CS+

CS-

High-Side Current-Sense Amplifier’s Negative Input. Connect R between CS- and CS+. CS- is

CS

referenced to LV.

High-Side Reference Voltage Input. A DC voltage at LV sets the lowest reference point for the high-side

current-sense and dimming MOSFET control circuitry.

LV

High-Side Regulator Output. H_REG provides a regulated supply for high-side circuitry. Bypass with a 1µF

ceramic capacitor to LV.

H_REG

HV

High-Side Positive Supply Voltage Input. HV provides power for dimming and LED current-sense circuitry.

HV is referenced to LV.

Internal MOSFET Gate Driver Output. Connect to a resistor between DRV and GT to set the rise and fall

times at LX.

DRV

19

GT

LX

Internal MOSFET GATE. Connect a resistor between GT and DRV to set the rise and fall times at LX.

20, 21

22, 23

Internal MOSFET Drain

SRC

Internal Power MOSFET Source

Slope Compensation Setting. Connect an appropriate external capacitor from SLP to AGND to generate a

ramp signal for stable operation.

24

SLP

25

26

TGRM

DIM

Dimming Comparator’s Reference/Ramp Generator

Dimming Control Input

Resistor-Programmable Internal Oscillator Setting. Connect a resistor from RT to AGND to set the internal

oscillator frequency.

27

28

—

RT

OV

EP

Overvoltage Protection Input. Connect OV to HI through a resistive voltage-divider to AGND to set the

overvoltage limit for the load. When the voltage at OV exceeds the 1.238V (typ) threshold, the gate drive

(DRV) for the switching MOSFET is disabled. Once V goes below 1.238V by 14mV, the switching

OV

MOSFET turns on again.

Exposed Pad. Connect EP to a large-area ground plane for effective power dissipation. Do not use as the

IC ground connection.

_______________________________________________________________________________________

7

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

DD

DS

0.5V

CMP

HV

DRMP

ADIM

DGT

CS+

LDOH

POR

3.88V

H_REG

DIM

RAMP

REF

1.2X

1.1X

1X

MAX6812

CS-

LX

CMP

IHI

CSA

LX

SRC

LV

IN

t

= 200ns

D

SRC

GT

2.5V

V

= 1.2V

= 0.3V

REFI

PREG

BG

V

RAMP

V

REF

V

DD

UVLO/

POR

LDOL

S

Q

L_REG

EN

G1

DRV

LATCH

1.2V

0.6V

R

SGND

HICCUP

REF

1X

EN

LOGIC

CONTROL

RT

OSC

I

LIM

DIM

SIGNAL

DIM

V

BE

CMP

PWM

1.238V

CMP

X0.2

SLP

TGRM

OV

MAX16812

COMP

FB

ERROR

AMPLIFIER

AND

DIMMING

S/H

2µs PULSE

LOW TO DISCHARGE

X1

CS_OUT

REFI

OVP

1.238V

SGND

AGND

Figure 1. Functional Diagram

_______________________________________________________________________________________

8

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

Current-Mode Control

The MAX16812 offers a current-mode control operation

Detailed Description

The MAX16812 is a current-mode PWM LED driver

feature with leading-edge blanking that blanks the

sensed current signal applied to the input of the PWM

current-mode comparator. In addition, a current-limit

comparator monitors the same signal at all times and

provides cycle-by-cycle current limit. An additional hic-

cup comparator limits the absolute peak current to two

times the cycle-by-cycle current limit. The leading-edge

blanking of the current-sense signal prevents noise at

the PWM comparator input from prematurely terminat-

ing the on-cycle. The switch current-sense signal con-

tains a leading-edge spike that results from the

MOSFET gate-charge current, and the capacitive and

diode reverse-recovery current of the power circuit. The

MAX16812’s capacitor-adjustable slope-compensation

feature allows for easy stabilization of the inner switch-

ing MOSFET current-mode loop. Upon triggering the

hiccup current limit, the soft-start capacitor on REF is

discharged and the gate drive to DRV is disabled.

Once the inductor current falls below the hiccup cur-

rent limit, the soft-start capacitor is released and it

begins to charge after 10µs.

with an integrated 0.2Ω power MOSFET for use in dri-

ving HBLEDs. By using two current regulation loops,

5% LED current accuracy is achieved. One current reg-

ulation loop controls the internal MOSFET peak current

through a sense resistor (R

) from SRC to ground,

SRC

while the other current regulation loop controls the

average LED current in a single LED string through

another sense resistor (R ) in series with the LEDs.

CS

The MAX16812 includes a cycle-by-cycle current limit

that turns off the gate drive to the internal MOSFET dur-

ing an overcurrent condition. The MAX16812 features a

programmable oscillator that simplifies and optimizes

the design of magnetics. The MAX16812 is well suited

for inputs from 5.5V to 76V. An external resistor in

series with the internal MOSFET gate can control the

rise and fall times on the drain of the internal switching

MOSFET, therefore minimizing EMI problems.

The MAX16812 high-frequency, current-mode PWM

HBLED driver integrates all the necessary building

blocks for driving a series LED string in an adjustable

constant current mode with PWM dimming. Current-

mode control with leading-edge blanking simplifies

control-loop design, and an external adjustable slope-

compensation control stabilizes the inner current-mode

loop when operating at duty cycles above 50%.

Slope Compensation

The MAX16812 uses an internal ramp generator for

slope compensation. The internal ramp signal resets at

the beginning of each cycle and slews at the rate pro-

grammed by the external capacitor connected at SLP

An input undervoltage lockout (UVLO) programs the

input supply startup voltage. An external voltage-

divider on EN programs the supply startup voltage. If

EN is directly connected to the input, the UVLO is set at

5V. A single external resistor from RT to AGND pro-

grams the switching frequency from 125kHz to 500kHz.

and an internal I

current source of 150µA. An inter-

SLP

nal attenuator attenuates the actual slope compensa-

tion signal by a factor of 0.2. Adjust the MAX16812

slew-rate capacitor by using the following equation:

I

SLP

SR

C

= 0.2 ×

SLOPE

Wide contrast (100:1) PWM dimming can be achieved

with the MAX16812. A DC input on DIM controls the

dimming duty cycle. The dimming frequency is set by

the sawtooth ramp frequency on TGRM (see the PWM

Dimming section). In addition, PWM dimming can be

achieved by applying a PWM signal to DIM with TGRM

set to a DC voltage less than 1.238V. A floating high-

voltage driver drives an external n-channel MOSFET in

series with the LED string. REFI allows analog dimming

of the LED current, further increasing the effective dim-

ming range over PWM alone. The MAX16812 has a 5µs

preprogrammed LED current rise and fall time.

where I

is the charging current in mA and C

is

SLOPE

SLP

the slope compensation capacitance on the SLP in µF,

and SR is the designed slope in mV/µs.

When using the MAX16812 for internal switching MOS-

FET duty cycles greater than 50%, the following condi-

tions must be met to avoid current-loop subharmonic

oscillations.

0.5 × R

× V

IND_OFF

SRC

SR ≥

mV/µs

is in volts, and L is in

IND_OFF

L

A nonlatching overvoltage protection limits the voltage

on the internal switching MOSFET under open-circuit

conditions in the LED string. The internal thermal shut-

down circuit protects the device if the junction tempera-

ture should exceed +165°C.

where R

is in mΩ, V

SRC

µH. L is the inductor connected to the LX pin of the

internal switching MOSFET and V is the voltage

IND_OFF

across the inductor during the off-time of the internal

MOSFET.

_______________________________________________________________________________________

9

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

minimizing output-voltage overshoot. While the part is in

UVLO, C is discharged (Figure 3). Upon coming out

Undervoltage Lockout

The MAX16812 features an adjustable UVLO through

the enable input (EN). Connect EN directly to IN to use

the 5V default UVLO. Connect EN to IN through a resis-

tive divider to ground to set the UVLO threshold. The

REF

of UVLO, an internal current source starts charging C

REF

during the soft-start cycle. Use the following equation to

calculate total soft-start time:

MAX16812 is enabled when V

(typ) threshold.

exceeds the 1.38V

EN

1.238

t

= C

×

REF

ST

I

REF

Calculate the EN UVLO resistor-divider values as fol-

lows (see Figure 2):

where I

is 40µA, C

is in µF, and t is in sec-

REF ST

REF

MAX6812

⎛

⎞

V

onds. Operation begins when REF ramps above 0.6V.

Once the soft-start is complete, REF is regulated to

1.238V, the internal voltage reference.

EN

- V

EN

R

= R

x

UV2

UV1

⎜

⎟

V

⎝

⎠

UVLO

where R

is in the 20kΩ range, V is the 1.38V (typ)

EN

UV1

Low-Side Internal

Switching MOSFET Driver Supply (L_REG)

L_REG is the regulated (5.2V) internal supply voltage

capable of delivering 20mA. L_REG provides power to

the gate drive of the internal switching power MOSFET.

EN threshold voltage, and V

is the desired input-

UVLO

voltage UVLO threshold in volts. Due to the 100mV hys-

teresis of the UVLO threshold, capacitor C is

EN

required to prevent chattering at the UVLO threshold

due to line impedance drops at power-up and during

dimming. If the undervoltage setting is very close to the

required minimum operating voltage, there can be

V

is referenced to AGND. Connect a 3.3µF

L_REG

ceramic capacitor from L_REG to AGND.

jumps in the voltage at IN while dimming. C

should

EN

High-Side Regulator (H_REG)

be large enough to limit the ripple on EN to less than

100mV (EN hysteresis) under these conditions so that it

does not turn on and off due to the ripple on IN.

H_REG is a low-dropout linear regulator referenced to

LV. H_REG provides the gate drive for the external

n-channel dimming MOSFET and also powers up the

MAX16812’s LED current-sense circuitry. Bypass

H_REG to LV with a 1µF ceramic capacitor.

Soft-Start

The soft-start feature of the MAX16812 allows the LED

string current to ramp up in a controlled manner, thus

V

IN

V

IN

R

UV2

MAX16812

EN

REF

C

REF

C

EN

UV1

AGND

Figure 2. UVLO Threshold Setting

Figure 3. Soft-Start Setting

10 ______________________________________________________________________________________

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

is low, COMP is disconnected from the output of the

error amplifier and CS_OUT is simultaneously discon-

nected from the buffered LED current-sense output sig-

nal (Figure 5). When the internal dimming signal is high,

the output of the op amp is connected to COMP and

CS_OUT is connected to the buffered LED current-

sense signal at the same time (Figure 4). This enables

the compensation capacitor to hold the charge when

the DIM signal has turned off the internal switching

MOSFET gate drive. To maintain the charge on the

High-Side Current-Sense Output (CS_OUT)

A high-side transconductance amplifier converts the

voltage across the LED current-sense resistor (R

)

CS

into an internal current output. This current flows

through an internal resistor connected to AGND. The

voltage gain for the LED current-sense signal is 4. The

amplified signal is then buffered and connected

through an internal switch to CS_OUT.

Internal Error Amplifier

The MAX16812 includes a built-in voltage-error amplifi-

er, which can be used to close the feedback loop. The

internal LED current-sense output signal is buffered

internally and then connected to CS_OUT through an

internal switch. CS_OUT is connected to the inverting

input (FB) pin of the error amplifier through a resistor.

See Figures 4 and 5. The reference voltage for the out-

put current is connected to REFI, the noninverting input

of the error amplifier. When the internal dimming signal

compensation capacitors C

and C

, the

COMP2

COMP1

capacitors should be of the low-leakage ceramic type.

When the internal dimming signal is enabled, the voltage

on the compensation capacitor forces the converter into

steady state almost instantaneously. The voltage on

COMP is subtracted from the internal slope compensa-

tion signal and is then connected to one of the inputs of

the PWM comparator. The PWM comparator input is of

the CMOS type with very low bias currents.

C

COMP2

STATE A

C

COMP1

R

COMP2

R

OUT

COMP1

X1

COMP

EA

REFI

Figure 4. Internal Error Amplifier Connection (Dimming Signal High)

C

COMP2

STATE B

C

COMP1

R

COMP2

R

OUT

COMP1

X1

COMP

EA

REFI

Figure 5. Internal Error Amplifier Connections (Dimming Signal Low)

______________________________________________________________________________________ 11

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

voltage produced by this current (through the current-

Analog Dimming

The MAX16812 offers analog dimming of the LED cur-

rent by allowing the application of an external voltage

at REFI. The output current is proportional to the volt-

age at REFI. Use a potentiometer from REF or directly

apply an external voltage source at REFI.

sense resistor) exceeds the current-limit (ILIM) com-

parator threshold, the MOSFET driver (DRV) quickly

terminates the current on-cycle. The 200ns leading-

edge blanking circuit suppresses the leading-edge

spike on the current-sense waveform from appearing at

the current-limit comparator. There is also a hiccup

comparator (HICCUP) that limits the peak current in the

internal switch set at twice the peak limit setting.

PWM Comparator

The PWM comparator uses the instantaneous switch

current, the error-amplifier output, and the slope com-

pensation to determine when the gate drive DRV to the

internal n-channel switching MOSFET turns off. In nor-

mal operation, gate drive DRV to the n-channel MOS-

FET turns off when:

Internal n-Channel

Switching MOSFET Driver (DRV)

L_REG provides power for the DRV output. Connect a

resistor from DRV to gate GT of the internal switching

MOSFET to control the switching MOSFET rise and fall

times, if necessary.

MAX6812

I

x R

≥ V

- V

OFFSET SCOMP

SW

SRC

COMP

where I

is the current through the internal n-channel

is the switch current-sense

SW

switching MOSFET, R

External Dimming

MOSFET Gate Drive (DGT)

DGT is the gate drive to the external dimming MOSFET

referenced to LV. H_REG provides the power to the

gate drive.

SRC

resistor, V

amplifier, V

is the output voltage of the internal

COMP

OFFSET

is the internal DC offset, which is a

V

BE

drop, and V

is the ramp function that starts

SCOMP

at zero and slews at the programmed slew rate (SR).

Internal Switching MOSFET Current Limit

Overvoltage Protection

The overvoltage protection (OVP) comparator com-

pares the voltage at OV with a 1.238V (typ) internal ref-

erence. When the voltage at OV exceeds the internal

reference, the OVP comparator terminates PWM switch-

ing and no further energy is transferred to the load.

Connect OV to HV through a resistive voltage-divider to

ground to set the overvoltage threshold at the output.

The current-sense resistor (R

), connected between

SRC

the source of the internal MOSFET and ground, sets the

current limit. The SRC input has a voltage trip level

(V

) of 600mV for the cycle-by-cycle current limit. Use

SRC

the following equation to calculate the value of R

:

SRC

V

SRC

R

=

SRC

I

LXLIM

Setting the Overvoltage Threshold

Connect OV to HV or to the high-side of the LEDs

through a resistive voltage-divider to set the overvolt-

age threshold at the output (Figure 6).

where I

is the peak current that flows through the

LXLIM

switching MOSFET at full load and low line. When the

V

LED+

V

LED+

HV

OV

MAX16812

R

OV1

OV2

OV1

OV

R

OV2

AGND

Figure 6. OVP Setting

12 ______________________________________________________________________________________

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

The overvoltage protection (OVP) comparator com-

pares the voltage at OV with a 1.238V (typ) internal ref-

erence. Use the following equation to calculate resistor

values:

REF

DIM

L_REG

V

− V

OV

⎛

⎞

OV_LIM

R

= R

x

OV2

R

TGRM

DIM1

DIM2

OV1

⎜

⎟

MAX16812

V

⎝

⎠

OV

TGRM

where V

is the 1.238V OV threshold. Choose R

OV1

OV

and R

to be reasonably high-value resistors to pre-

OV2

C

TGRM

vent the discharge of filter capacitors. This prevents

degraded performance during dimming.

R

AGND

Internal Oscillator Switching Frequency

The oscillator switching frequency is programmed by a

resistor connected from RT to AGND. To program the

oscillator frequency above 125kHz, choose the appro-

priate resistor RT from the curves shown in the

Figure 7. PWM Dimming from REF

Oscillator Frequency vs. R graph in the Typical

T

Operating Characteristics section.

PWM dimming can also be achieved by connecting

TGRM to a DC voltage less than V and applying the

PWM Dimming

REF

PWM signal at DIM. The moment the internal dimming

signal goes low, gate drive DRV to the internal switching

MOSFET is turned off. The error amplifier goes to state B

(see the Internal Error Amplifier section and Figures 4

and 5). The peak current in the inductor prior to dis-

PWM dimming can be achieved by driving DIM with an

analog voltage less than V

. See Figure 7. An exter-

REF

nal resistor on TGRM from L_REG in conjunction with

the ramp capacitor, C , from TGRM to AGND cre-

TGRM

ates a sawtooth ramp that is compared with the DC

voltage on DIM. The output of the comparator is a pul-

abling DRV is I . Gate drive DGT to the external dim-

LX

ming MOSFET is held high. Then after a switchover

sating dimming signal. The frequency f

sawtooth signal on TGRM is given by:

of the

RAMP

period, gate voltage V

on the external dimming

DGT

MOSFET is linearly controlled to reduce the LED current

to 0. The fall time of the LED current is controlled by an

internal timing circuit to 5µs for the MAX16812. During

this period, the gate (DRV) to the internal switching

MOSFET is enabled. After the fall time, the gate drive to

the external dimming MOSFET is turned off and the gate

drive to the internal switching MOSFET is still held high

after the switchover period. The peak current in the

3.67

f

≅

RAMP

C

× R

TGRM

Use the following formula to calculate the voltage V

necessary for a given output duty cycle, D:

,

DIM

V

= D x 1.238V

DIM

inductor is controlled at I . Then after a time period of

LX

where V

is the DC voltage applied to DIM in volts.

DIM

20µs, the gate drive is disabled. The scope shots in

Figures 8–11 show the dimming waveforms.

The DC voltage for DIM can also be created by con-

necting DIM to REF through a resistive voltage-divider.

Using the required dimming input voltage, V

, calcu-

DIM

late the resistor values for the divider string using the

following equation:

R

DIM2

= [V

/ (V

- V

)] x R

DIM DIM1

DIM

REF

where V

is the voltage on REF.

REF

______________________________________________________________________________________ 13

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX16812 fig08

MAX16812 fig10

V

OUT

10V/div

V

OUT

100mA/div

0A, 0V

100mA/div

0A, 0V

I

LED

I

LED

DRV

2V/div

0V

2V/div

0V

MAX6812

V

DRV

10µs/div

Figure 8. LED Current, Output Voltage, and DRV Waveforms

when DIM Signal Goes Low

Figure 10. LED Current, Output Voltage, and DRV Waveforms

when DIM Signal Goes High

MAX16812 fig09

MAX16812 fig11

I

LED

I

LED

100mA/div

V

DIM

V

DIM

5V/div

0A, 0V

V

DRV

V

DRV

2V/div

0V

10µs/div

Figure 11. LED Current, DIM Signal, and DRV Waveforms when

DIM Signal Goes High

Figure 9. LED Current, DIM Signal, and DRV Waveforms when

DIM Signal Goes Low

When the DIM signal goes high, the LED current is

gradually increased to the programmed value. The rise

time of the LED current is controlled to 5µs for the

MAX16812 by controlling the voltage on DGT. After the

rise time, an internal sensing circuit monitors the volt-

age across the drain to the source of the external dim-

ming MOSFET. The LED current is now controlled at the

programmed value by a linear current regulating cir-

cuit. Once the voltage across the drain to source of the

dimming MOSFET drops below 0.5V, the reference for

the linear current regulating circuit is increased to 1.1

times the programmed value. The gate drive (DRV) to

the internal switching MOSFET is enabled and the error

amplifier is returned to state A (see the Internal Error

Amplifier section and Figures 4 and 5).

Fault Protection

The MAX16812 features built-in overvoltage protection

and thermal shutdown. Connect a resistive voltage-

divider between HV, OV, and AGND to program the over-

voltage protection. In the case of a short circuit across

the LED string, the temperature of the external dimming

MOSFET could exceed the maximum allowable junction

temperature. This is due to excess power dissipation in

the MOSFET. Use the fault protection circuit shown in

Figure 12 to protect the external dimming MOSFET.

Internal thermal shutdown in the MAX16812 safely turns

off the IC when the junction temperature exceeds

+165^°C.

14 ______________________________________________________________________________________

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

V

IN

100kΩ

GND

TO EN PIN OF

MAX16812

TOVER

5.1V

ZENER

MAX6501

TO L_REG PIN

OF MAX16812

V

CC

4.7µF

Figure 12. Dimming MOSFET Protection

where V

is the maximum input voltage, f

is the

Inductor Selection

The minimum required inductance is a function of the

operating frequency, the input-to-output voltage differ-

INMAX

SW

switching frequency, and V

is the output voltage.

OUT

Boost Configuration: In the boost converter, the aver-

age inductor current varies with the input voltage and

the maximum average current occurs at the lowest

input voltage. For the boost converter, the average

inductor current is equal to the input current. In this

case, the inductance, L, is calculated as:

ential and the peak-to-peak inductor current (∆I ).

L

Higher ∆I allows for a lower inductor value while a

L

lower ∆I requires a higher inductor value. A lower

L

inductor value minimizes size and cost, improves large-

signal transient response, but reduces efficiency due to

higher peak currents and higher peak-to-peak output

ripple voltage for the same output capacitor. On the

other hand, higher inductance increases efficiency by

V

x V

− V

(

)

INMIN

OUT INMIN

L =

V

x f

x ∆I

OUT SW L

reducing the ripple current, ∆I . However, resistive

L

losses due to the extra turns can exceed the benefit

gained from lower ripple current levels, especially when

the inductance is increased without allowing for larger

inductor dimensions. A good compromise is to choose

where V

output voltage, and f

Figure 14.

is the minimum input voltage, V

is the

INMIN

OUT

is the switching frequency. See

SW

Buck-Boost Configuration: In a buck-boost converter

(see the Typical Application Circuit), the average

inductor current is equal to the sum of the input current

and the LED current. In this case, the inductance, L, is:

∆I equal to 30% of the full load current. The inductor

L

saturating current specification is also important to

avoid runaway current during output overload and con-

tinuous short-circuit conditions.

Buck Configuration: In a buck configuration (Figure

13), the average inductor current does not vary with the

input. The worst-case peak current occurs at the high-

est input voltage. In this case, the inductance, L, for

continuous conduction mode is given by:

V

x V

INMIN

OUT

L =

V

+ V

x f

x ∆I

(

OUT

INMIN

)

SW L

where V

is the minimum input voltage, V

is the

INMIN

output voltage, and f

OUT

is the switching frequency.

SW

V

x V

(

− V

)

OUT

INMAX OUT

L =

V

x f

x ∆I

INMAX

SW L

______________________________________________________________________________________ 15

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

C

OUT

V

IN

D

R

OUT

CS

C

H_REG

IN

IN HV

H_REG

LX

LV DD DGT CS-

CS+

SRC

R

EN

SRC

MAX6812

R

RT

GT

RT

C

L_REG

MAX16812

G

L_REG

DRV

SLP

R

TGRM

C

SLP

TGRM

DIM

C

TGRM

COMP

FB

OV SGND AGND

REF

REFI CS_OUT

V

OUT

R

COMP1

C

REF

C

R

COMP1

OV1

OV2

R

REF1

R

COMP2

R

REF2

C

COMP2

Figure 13. Buck Configuration

C

H_REG

R

D

OUT

CS

V

OUT

R

V

IN

C

OUT

SRC

CS-

CS+

DGT

DD

H_REG HV

LX

SRC

LV

V

IN

GT

C

R

G

EN

RT

IN1

R

RT

DRV

SLP

C

L_REG

MAX16812

L_REG

C

R

SLP

TGRM

DIM

C

TGRM

COMP

FB

OV SGND AGND

REF

REFI CS_OUT

V

OUT

R

COMP1

C

REF

C

R

COMP1

OV1

R

REF1

R

OV2

COMP2

R

REF2

C

COMP2

Figure 14. Boost Configuration

16 ______________________________________________________________________________________

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

L1

L2

C

S

C

H_REG

D

OUT

VOUT

R

CS

V

IN

C

OUT

R

SRC

V

IN

LV

IN

SRC

GT

C

IN1

EN

RT

R

G

C

L_REG

DRV

SLP

MAX16812

L_REG

C

SLP

R

TGRM

DIM

COMP

C

TGRM

VOUT

C

COMP1

R

OV1

COMP2

R

REF1

R

COMP1

OV2

R

REF2

C

COMP2

Figure 15. SEPIC Configuration

In a buck-boost configuration, the output capacitance,

OUT

Output Capacitor

C

is:

The function of the output capacitor is to reduce the

output ripple to acceptable levels. The ESR, ESL, and

the bulk capacitance of the output capacitor contribute

to the output ripple. In most of the applications, the out-

put ESR and ESL effects can be dramatically reduced

by using low-ESR ceramic capacitors. To reduce the

ESL effects, connect multiple ceramic capacitors in

parallel to achieve the required capacitance.

2 × V

× I

OUT

C

≥

OUT

∆V × (V

+ V

) × f

R

OUT

INMIN SW

where V

the output current.

is the voltage across the load and I

is

OUT

Input Capacitor

In a buck configuration, the output capacitance, C

is calculated using the following equation:

,

OUT

An input capacitor connected between IN and ground

must be used when configuring the MAX16812 as a

buck converter. Use a low-ESR input capacitor that can

handle the maximum input RMS ripple current.

Calculate the maximum RMS ripple using the following

equation:

(V

− V

) × V

INMAX

OUT OUT

C

≥

OUT

2

∆V × 2 × L × V

× f

SW

R

INMAX

where ∆V is the maximum allowable output ripple.

R

I

×

V

× (V

- V

)

In a boost configuration, the output capacitance, C

is calculated as:

,

OUT

INMIN

OUT

I

=

IN(RMS)

V

INMIN

(V

− V

) × 2 × I

× f

OUT

INMIN

OUT

When using the MAX16812 in a boost or buck-boost

configuration, the input capacitor’s RMS current is low

and the input capacitance can be small. However, an

additional electrolytic capacitor may be required to pre-

vent oscillations due to line impedances.

C

≥

OUT

∆V × V

R

OUT

SW

where C

is the output capacitor.

OUT

______________________________________________________________________________________ 17

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

• Keep the high-current paths short, especially at the

ground terminals. This practice is essential for stable,

jitter-free operation. Keep switching loops short.

Layout Recommendations

Typically, there are two sources of noise emission in a

switching power supply: high di/dt loops and high dv/dt

surfaces. For example, traces that carry the drain cur-

rent often form high di/dt loops. Similarly, the drain of

the internal MOSFET connected to the LX pin presents

a dv/dt source. Keep all PCB traces carrying switching

currents as short as possible to minimize current loops.

Use ground planes for best results.

• Connect AGND and SGND to a ground plane.

Ensure a low-impedance connection between all

ground points.

• Keep the power traces and load connections short.

This practice is essential for high efficiency. Use

thick copper PCBs to enhance full-load efficiency.

Careful PCB layout is critical to achieve low switching

losses and clean, stable operation. Use a multilayer

board whenever possible for better noise immunity and

power dissipation. Follow these guidelines for good

PCB layout:

MAX6812

• Ensure that the feedback connection to FB is short

and direct.

• Route high-speed switching nodes away from the

sensitive analog areas.

• To prevent discharge of the compensation capaci-

• Use a large copper plane under the MAX16812

package. Ensure that all heat-dissipating compo-

nents have adequate cooling. Connect the exposed

pad of the device to the ground plane.

tors, C

and C

, during the off-time of

COMP2

COMP1

the dimming cycle, ensure that the PCB area close

to these components has extremely low leakage.

• Isolate the power components and high-current

paths from sensitive analog circuitry.

18 ______________________________________________________________________________________

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

Typical Application Circuit

BUCK-BOOST CONFIGURATION

C

H_REG

R

CS

D

OUT

V

OUT

C

OUT

R

SRC

R

CS-

CS+

DGT

DD

H_REG HV

LX

LV

IN

SRC

GT

V

IN

C

IN1

EN

RT

G

RT

DRV

SLP

C

L_REG

MAX16812

L_REG

C

SLP

R

TGRM

DIM

C

TGRM

COMP

FB

OV SGND AGND

REF

REFI CS_OUT

V

OUT

R

COMP1

C

REF

C

R

COMP1

OV1

R

REF1

R

OV2

COMP2

R

REF2

C

COMP2

Pin Configuration

Chip Information

PROCESS: BiCMOS

TOP VIEW

TRANSISTOR COUNT: 8699

21 20 19 18 17 16 15

14

13

SRC 22

SRC 23

CS-

CS+

12 DGT

24

25

26

27

28

SLP

TGRM

DIM

RT

DD

11

10

9

MAX16812

SGND

L_REG

IN

*EP

+

8

OV

1

2

3

4

5

6

7

*EP = EXPOSED PAD

TQFN

______________________________________________________________________________________ 19

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

Package Information

(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,

go to www.maxim-ic.com/packages.)

MAX6812

PACKAGE OUTLINE,

16, 20, 28, 32, 40L THIN QFN, 5x5x0.8mm

1

K

21-0140

2

20 ______________________________________________________________________________________

Integrated High-Voltage LED Driver

with Analog and PWM Dimming Control

MAX6812

Package Information (continued)

(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,

go to www.maxim-ic.com/packages.)

PACKAGE OUTLINE,

16, 20, 28, 32, 40L THIN QFN, 5x5x0.8mm

2

K

21-0140

2

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are

implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 21

is a registered trademark of Maxim Integrated Products, Inc.

Heaney


型号：	MAX16812ATI+
厂家：	MAXIM INTEGRATED PRODUCTS
描述：	Integrated High-Voltage LED Driver with Analog and PWM Dimming Control 集成高压LED驱动器，模拟和PWM调光控制驱动器高压
文件：	总21页 (文件大小：353K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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