LT1578I [Linear]
1.5A, 200kHz Step-Down Switching Regulator; 1.5A ， 200kHz的降压型开关稳压器
| 型号： | LT1578I |
| 厂家： | 
Linear |
| 描述： | 1.5A, 200kHz Step-Down Switching Regulator |
| 文件： | 总28页 (文件大小：289K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT1578/LT1578-2.5
1.5A, 200kHz Step-Down
Switching Regulator
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FEATURES
DESCRIPTIO
TheLT®1578isa200kHzmonolithicbuckmodeswitching
regulator. A 1.5A switch is included on the die along with
allthenecessaryoscillator, controlandlogiccircuitry. The
topology is current mode for fast transient response and
goodloopstability.TheLT1578isamodifiedversionofthe
LT1507 that has been optimized for noise sensitive appli-
cations. It will operate over a 4V to 15V input range.
■
1.5A Switch Current
High Efficiency—Low Loss 0.2Ω Switch
■
■
■
Constant 200kHz Switching Frequency
4V to 15V Input VoltageRange
Minimum Output: 1.21V
■
■
■
■
■
■
■
■
Low Supply Current: 1.9mA
Low Shutdown Current: 20µA
Easily Synchronizable Up to 400kHz
Cycle-by-Cycle Current Limit
Reduced EMI Generation
Low Thermal Resistance SO-8 Package
Uses Small Low Value Inductors
In addition, the reference voltage has been lowered to al-
low the device to produce output voltages down to 1.2V.
Quiescent current has been reduced by a factor of two.
Switchonresistancehasbeenreducedby30%.Switchtran-
sition times have been slowed to reduce EMI generation.
The oscillator frequency has been reduced to 200kHz to
maintain high efficiency over a wide output current range.
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APPLICATIO S
The pinout has been changed to improve PC layout by al-
lowingthehighcurrent,highfrequencyswitchingcircuitry
to be easily isolated from low current, noise sensitive con-
trol circuitry. The new SO-8 package includes a fused
groundleadthatsignificantlyreducesthethermalresistance
of the device to extend the ambient operating temperature
range. Standard surface mount external parts can be used
including the inductor and capacitors.
■
Portable Computers
■
Battery-Powered Systems
■
Battery Chargers
Distributed Power Systems
■
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATION
Efficiency vs Load Current
90
85
3.3V Buck Converter
INPUT
5V TO 15V
C2
0.33µF
+
C3*
D2
1N914
10µF TO
50µF
80
75
70
65
60
55
50
L1**
15µH
V
BOOST
IN
OUTPUT**
3.3V, 1.25A
V
SW
FB
LT1578
SHDN
GND
OPEN = ON
V
C
R1
8.66k
C1
* RIPPLE CURRENT RATING ≥ I /2
OUT
+
R2
4.99k
C
D1
V
V
= 3.3V
100µF, 10V
SOLID
C
OUT
= 5V
** INCREASE L1 TO 30µH FOR LOAD
CURRENTS ABOVE 0.6A AND TO
60µH ABOVE 1A
100pF
1N5818
IN
L = 25µH
TANTALUM
SEE APPLICATIONS INFORMATION
1578 TA01
0
0.50 0.75 1.00
1.25 1.50
0.25
LOAD CURRENT (A)
1578 TA02
1
LT1578/LT1578-2.5
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
ORDER PART
TOP VIEW
Input Voltage .......................................................... 16V
BOOST Pin Above Input Voltage ............................. 10V
SHDN Pin Voltage..................................................... 7V
SENSE Pin Voltage .................................................... 4V
FB Pin Voltage (Adjustable Part)............................ 3.5V
FB Pin Current (Adjustable Part)............................ 1mA
SYNC Pin Voltage ..................................................... 7V
Operating Junction Temperature Range
LT1578C............................................... 0°C to 125° C
LT1578I ........................................... –40°C to 125°C
Storage Temperature Range ................ –65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
NUMBER
1
2
3
4
8
7
6
5
V
SYNC
SW
LT1578CS8
LT1578IS8
LT1578CS8-2.5
LT1578IS8-2.5
SHDN
V
IN
FB/SENSE
BOOST
GND
V
C
S8 PACKAGE
8-LEAD PLASTIC SO
S8 PART MARKING
θJA = 80°C/ W WITH FUSED GROUND PIN
CONNECTED TO GROUND PLANE OR
LARGE LANDS
1578
1578I
157825
578I25
Consult factory for Military grade parts.
The ● denotes specifications which apply over the full operating tempera-
ture range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.
ELECTRICAL CHARACTERISTICS
PARAMETER
CONDITIONS
All Conditions
All Conditions
MIN
TYP
MAX
UNITS
Feedback Voltage
1.195 1.21
1.18
2.46
2.44
1.225
1.24
2.54
2.56
V
V
V
V
●
●
Sense Voltage (Fixed 2.5)
2.5
Sense Pin Resistance
Reference Voltage Line Regulation
Feedback Input Bias Current
5.7
9.5
0.01
0.5
13.7
0.03
2
kΩ
%/V
µA
4.3V ≤ V ≤ 15V
●
●
IN
Error Amplifier Voltage Gain (Notes 2, 10)
Error Amplifier Transconductance (Note 10) ∆I (V ) = ±10µA
200
800
400
400
1050
1300
1700
µMho
µMho
C
●
V Pin to Switch Current Transconductance
Error Amplifier Source Current
Error Amplifier Sink Current
1.5
110
130
0.8
2.1
2
A/V
µA
µA
V
V
A
C
V
V
= 1.1V
= 1.4V
●
●
40
50
190
200
FB
FB
V Pin Switching Threshold
Duty Cycle = 0
C
V Pin High Clamp
C
Switch Current Limit
V Open, V = 1.1V, DC ≤ 50%
C
●
1.5
3.5
FB
Slope Compensation (Note 8)
Switch On Resistance (Note 7)
DC = 80%
0.3
0.2
A
Ω
Ω
I
= 1.5A
0.35
0.45
SW
●
●
Maximum Switch Duty Cycle
V
FB
= 1.1V
90
86
94
94
%
%
Minimum Switch Duty Cycle (Note 9)
Switch Frequency
8
200
%
kHz
kHz
%/V
V
V Set to Give 50% Duty Cycle
180
160
220
240
0.15
1.0
4.3
3.0
C
●
●
●
●
●
Switch Frequency Line Regulation
Frequency Shifting Threshold on FB Pin
Minimum Input Voltage (Note 3)
Minimum Boost Voltage (Note 4)
4.3V ≤ V ≤ 15V
0
IN
∆f = 10kHz
0.4
0.74
4.0
2.3
V
V
I
≤ 1.5A
SW
2
LT1578/LT1578-2.5
The ● denotes specifications which apply over the full operating tempera-
ture range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.
ELECTRICAL CHARACTERISTICS
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Boost Current (Note 5)
I
I
= 0.5A
= 1.5A
●
●
9
27
18
50
mA
mA
SW
SW
V
Supply Current (Note 6)
●
1.9
20
2.7
50
75
mA
µA
µA
IN
Shutdown Supply Current
V
= 0V, V ≤ 15V, V = 0V, V Open
SHDN
IN
SW
C
●
●
Lockout Threshold
V Open
C
2.34
2.42
2.50
V
Shutdown Thresholds
V Open Device Shutting Down
Device Starting Up
●
●
0.13
0.25
0.37
0.45
0.60
0.7
V
V
C
Synchronization Threshold
Synchronizing Range
SYNC Pin Input Resistance
1.5
2.2
400
V
kHz
kΩ
250
40
Note 1: Absolute Maximum Ratings are those values beyond which the life
Note 7: Switch on resistance is calculated by dividing V to V voltage
IN SW
by the forced current (1.5A). See Typical Performance Characteristics for
the graph of switch voltage at other currents.
Note 8: Slope compensation is the current subtracted from the switch
current limit at 80% duty cycle. See Maximum Output Load Current in the
Applications Information section for further details.
Note 9: Minimum on-time is 400ns typical. For a 200kHz operating
frequency this means the minimum duty cycle is 8%. In frequency
foldback mode, the effective duty cycle will be less than 8%.
of a device may be impaired.
Note 2: Gain is measured with a V swing equal to 200mV above the
C
switching threshold level to 200mV below the upper clamp level.
Note 3: Minimum input voltage is not measured directly, but is guaranteed
by other tests. It is defined as the voltage where internal bias lines are still
regulated so that the reference voltage and oscillator frequency remain
constant. Actual minimum input voltage to maintain a regulated output will
depend on output voltage and load current. See Applications Information.
Note 4: This is the minimum voltage across the boost capacitor needed to
Note 10: Transconductance and voltage gain refer to the internal amplifier
exclusive of the voltage divider. To calculate gain and transconductance
referred to the sense pin on the fixed voltage parts, divide values shown by
the ratio 2.5/1.21.
guarantee full saturation of the internal power switch.
Note 5: Boost current is the current flowing into the boost pin with the pin
held 5V above input voltage. It flows only during switch on time.
Note 6: Input supply current is the bias current drawn by the input pin
with switching disabled.
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TYPICAL PERFORMANCE CHARACTERISTICS
Switch Voltage Drop
Switch Peak Current Limit
Feedback Pin Voltage
0.5
0.4
0.3
0.2
0.1
0
2.5
2.0
1.5
1.0
0.5
0
1.23
1.22
1.21
1.20
1.19
TYPICAL
125°C
25°C
MINIMUM
–20°C
–25
0
25
50
75
125
0
0.50 0.75 1.00
SWITCH CURRENT (A)
1.25 1.50
0
20
80
–50
100
0.25
40
60
DUTY CYCLE (%)
100
JUNCTION TEMPERATURE (°C)
1576 G01
1576 G02
1576 G03
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LT1578/LT1578-2.5
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TYPICAL PERFORMANCE CHARACTERISTICS
Shutdown Pin Bias Current
(VSHDN = Lockout Threshold)
Shutdown Pin Bias Current
(VSHDN = Shutdown Threshold)
Shutdown Thresholds
4
3
2
1
0
180
160
140
120
100
80
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
AT 2.44V STANDBY THRESHOLD
(CURRENT FLOWS OUT OF PIN)
START-UP
SHUTDOWN
60
CURRENT REQUIRED TO FORCE
SHUTDOWN (FLOWS OUT OF PIN).
AFTER SHUTDOWN, CURRENT
DROPS TO A FEW µA
40
20
0
–25
0
25
50
75
125
–50
100
–25
0
25
50
75
125
–25
0
25
50
75
125
–50
100
–50
100
JUNCTION TEMPERATURE (°C)
JUNCTION TEMPERATURE (°C)
JUNCTION TEMPERATURE (°C)
1576 G06
1576 G04
1576 G05
Standby Thresholds
Shutdown Supply Current
Shutdown Supply Current
25
20
15
10
5
2.46
2.45
2.44
2.43
2.42
2.41
2.40
30
25
20
15
10
5
V
IN
= 10V
V
= 0V
SHDN
ON
STANDBY
0
0
0
5
10
15
50
100 125
–50 –25
0
25
75
0
0.1
0.2
0.3
0.4
INPUT VOLTAGE (V)
JUNCTION TEMPERATURE (°C)
SHUTDOWN VOLTAGE (V)
1576 G08
1576 G07
1576 G010
Error Amplifier Transconductance
Frequency Foldback
Error Amplifier Transconductance
2000
1500
1000
500
200
150
100
50
1600
1400
1200
1000
800
600
400
200
0
250
200
150
100
50
SWITCHING FREQUENCY
PHASE
GAIN
V
C
C
OUT
2.4pF
R
OUT
570k
V
1 × 10–3
(
)
FB
0
ERROR AMPLIFIER EQUIVALENT CIRCUIT
= 50Ω
0
R
LOAD
100
FEEDBACK PIN CURRENT
1.0
1.5
FEEDBACK VOLTAGE (V)
–500
–50
0
10
1k
10k
100k
1M
–25
0
25
50
75
125
–50
100
0
0.5
2.0
FREQUENCY (Hz)
JUNCTION TEMPERATURE (°C)
1576 G09
1576 G11
1576 G12
4
LT1578/LT1578-2.5
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TYPICAL PERFORMANCE CHARACTERISTICS
Minimum Input Voltage to Start
with 3.3V Output
Switching Frequency
Switch Current Limit Foldback
3.0
2.5
2.0
1.5
1.0
0.5
0
4.50
240
220
200
180
160
4.25
4.00
3.75
3.50
0
0.4
0.6
0.8
1.0
1.2
0.2
1
10
100
1000
–25
0
25
50
75
125
–50
100
FEEDBACK PIN VOLTAGE (V)
LOAD CURRENT (mA)
JUNCTION TEMPERATURE (°C)
1578 G19
1576 G14
1576 G13
Maximum Output Current
at VOUT = 3.3V
Maximum Output Current
at VOUT = 2.5V
Maximum Output Current
at VOUT = 5V
1.6
1.6
1.4
1.2
1.0
1.6
1.4
1.2
1.0
L = 60µH
L = 60µH
L = 60µH
1.4
1.2
1.0
L = 30µH
L = 15µH
L = 30µH
L = 15µH
L = 30µH
L = 15µH
0.8
0.6
0.8
0.6
0.8
0.6
0.4
0.2
0
0.4
0.2
0
0.4
0.2
0
6
8
10
12
9
12
4
6
8
10
INPUT VOLTAGE (V)
12
14
4
14
6
15
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
1578 G17
1578 G15
1578 G16
VC Pin Shutdown Threshold
BOOST Pin Current
30
25
20
15
10
5
1.0
0.8
0.6
0.4
0.2
0
0
0
0.50 0.75 1.00
1.25 1.50
0.25
–25
0
25
50
75
125
–50
100
SWITCH CURRENT (A)
JUNCTION TEMPERATURE (°C)
1576 G20
1576 G21
Kool Mµ is a registered trademark of Magnetics, Inc.
Metglas is a registered trademark of AlliedSignal, Inc.
5
LT1578/LT1578-2.5
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PIN FUNCTIONS
VSW (Pin 1): The switch pin is the emitter of the on-chip
power NPN switch. This pin is driven up to the input pin
voltage during switch on time. Inductor current drives the
switch pin negative during switch off time. Negative volt-
age is clamped with the external catch diode. Maximum
negative switch voltage allowed is –0.8V.
pin should be attached to a large copper area to improve
thermal resistance.
VC (Pin 5): The VC pin is the output of the error amplifier
and the input to the peak switch current comparator. It is
normally used for frequency compensation, but can do
double duty as a current clamp or control loop override.
This pin sits at about 1V for very light loads and 2V at
maximum load. It can be driven to ground to shut off the
regulator, but if driven high, current must be limited to
4mA.
VIN (Pin 2): This is the collector of the on-chip power NPN
switch. This pin powers the internal circuitry and internal
regulator.AtNPNswitchonandoff,highdI/dtedgesoccur
throughthispin. Keeptheexternalbypassandcatchdiode
close to this pin. Trace inductance in this path will create
a voltage spike at switch off, adding to the VCE voltage
across the internal NPN.
FB/SENSE (Pin 6): The feedback pin is used to set output
voltage using an external voltage divider that generates
1.21V at the pin with the desired output voltage. The fixed
voltage (2.5V) parts have the divider included on the chip
and the FB pin is used as a sense pin, connected directly
to the 2.5V output. Three additional functions are per-
formed by the FB pin. When the pin voltage drops below
0.7V, the switch current limit and the switching frequency
arereducedand theexternalsyncfunctionisdisabled.See
FeedbackPinFunctionsectioninApplicationsInformation
for details.
BOOST (Pin 3): The BOOST pin is used to provide a drive
voltage, higher than the input voltage, to the internal
bipolarNPNpowerswitch. Withoutthisaddedvoltage, the
typical switch voltage loss would be about 1.5V. The
additional boost voltage allows the switch to saturate with
its voltage drop approximating that of a 0.2Ω FET struc-
ture. Efficiencyimprovesfrom75%forconventionalbipo-
lar designs to >88% for the LT1578.
GND(Pin4):TheGNDpinconnectionneedsconsideration
for two reasons. First, it acts as the reference for the
regulated output, so load regulation will suffer if the
“ground” end of the load is not at the same voltage as the
GND pin of the IC. This condition will occur when load
current or other currents flow through metal paths be-
tween the GND pin and the load ground point. Keep the
ground path short between the GND pin and the load and
use a ground plane when possible. The second consider-
ation is EMI caused by GND pin current spikes. Internal
capacitance between the VSW pin and the GND pin creates
very narrow (<10ns) current spikes in the GND pin. If the
GND pin is connected to system ground with a long metal
trace, this trace may radiate EMI. Keep the path between
theinputbypassandtheGNDpinshort.TheGNDpinofthe
SO-8 package is directly attached to the internal tab. This
SHDN (Pin 7): The shutdown pin is used to turn off the
regulator and to reduce input drain current to a few
microamperes. Actually, this pin has two separate thresh-
olds, one at 2.42V to disable switching, and a second at
0.4V to force complete micropower shutdown. The 2.42V
threshold functions as an accurate undervoltage lockout
(UVLO). This can be used to prevent the regulator from
operating until the input voltage has reached a predeter-
mined level.
SYNC (Pin 8): The SYNC pin is used to synchronize the
internal oscillator to an external signal. It is directly logic
compatible and can be driven with any signal between
10% and 90% duty cycle. The synchronizing range is
equal to initial operating frequency, up to 400kHz. When
not used, this pin should be grounded. See Synchronizing
section in Applications Information for details.
6
LT1578/LT1578-2.5
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BLOCK DIAGRAM
The LT1578 is a constant frequency, current mode buck
converter. This means that there is an internal clock and
twofeedbackloopsthatcontrolthedutycycleofthepower
switch. In addition to the normal error amplifier, there is a
current sense amplifier that monitors switch current on a
cycle-by-cycle basis. A switch cycle starts with an oscilla-
tor pulse which sets the RS flip-flop to turn the switch on.
When switch current reaches a level set by the inverting
input of the comparator, the flip-flop is reset and the
switch turns off. Output voltage control is obtained by
using the output of the error amplifier to set the switch
current trip point. This technique means that the error
amplifier commands current to be delivered to the output
rather than voltage. A voltage fed system will have low
phase shift up to the resonant frequency of the inductor
and output capacitor, then an abrupt 180° shift will occur.
The current fed system will have 90° phase shift at a much
lower frequency, but will not have the additional 90° shift
until well beyond the LC resonant frequency. This makes
itmucheasiertofrequencycompensatethefeedbackloop
and also gives much quicker transient response.
High switch efficiency is attained by using the BOOST pin
to provide a voltage to the switch driver which is higher
thantheinputvoltage,allowingtheswitchtosaturate.This
boosted voltage is generated with an external capacitor
and diode. Two comparators are connected to the shut-
down pin. One has a 2.42V threshold for undervoltage
lockout and the second has a 0.4V threshold for complete
shutdown.
0.025Ω
INPUT
+
–
CURRENT SENSE
2.9V BIAS
REGULATOR
INTERNAL
CC
AMPLIFIER DC
V
VOLTAGE GAIN = 35
SLOPE COMP
BOOST
Σ
0.8V
200kHz
OSCILLATOR
S
R
SYNC
Q1
POWER
SWITCH
R
DRIVER
CIRCUITRY
CURRENT
COMPARATOR
S
FLIP-FLOP
+
–
SHUTDOWN
COMPARATOR
–
+
V
SW
0.4V
FREQUENCY
SHDN
SHIFT CIRCUIT
3.5µA
FOLDBACK
CURRENT
LIMIT
Q2
+
–
CLAMP
–
FB
LOCKOUT
COMPARATOR
+
ERROR
V
C
AMPLIFIER
2.42V
1.21V
g
= 1000µMho
m
GND
1578 BD
Figure 1. Block Diagram
7
LT1578/LT1578-2.5
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APPLICATIONS INFORMATION
FEEDBACK PIN FUNCTIONS
More Than Just Voltage Feedback
The feedback (FB) pin on the LT1578 is used to set output
voltage and provide several overload protection features.
The first part of this section deals with selecting resistors
to set output voltage and the remaining part talks about
foldback frequency and current limiting created by the FB
pin. Please read both parts before committing to a final
design. The fixed 2.5V LT1578-2.5 has internal divider
resistors and the FB pin, renamed SENSE, is connected
directly to the 2.5V output.
The feedback pin is used for more than just output voltage
sensing. It also reduces switching frequency and current
limit when output voltage is very low (see the Frequency
Foldback graph in Typical Performance Characteristics).
ThisisdonetocontrolpowerdissipationinboththeICand
the external diode and inductor during short-circuit con-
ditions. A shorted output requires the switching regulator
to operate at very low duty cycles, and the average current
throughthediodeandinductorisequaltotheshort-circuit
current limit of the switch (typically 2A for the LT1578,
folding back to less than 0.77A). Minimum switch on time
limitations would prevent the switcher from attaining a
sufficiently low duty cycle if switching frequency were
maintained at 200kHz, so frequency is reduced by about
5:1 when the feedback pin voltage drops below 0.7V (see
FrequencyFoldbackgraph). Thisdoesnotaffectoperation
with normal load conditions; one simply sees a gear shift
in switching frequency during start-up as the output
voltage rises.
The suggested value for the output divider resistor (see
Figure 2) from FB to ground (R2) is 5k or less, and a
formula for R1 is shown below. The output voltage error
caused by ignoring the input bias current on the FB pin is
less than 0.25% with R2 = 5k. Please read the following
if divider resistors are increased above the suggested
values.
R2 V
−1.21
(
)
OUT
R1=
1.21
LT1578
V
SW
TO FREQUENCY
SHIFTING
OUTPUT
5V
1.4V
Q1
ERROR
AMPLIFIER
R1
1.21V
+
–
R3
1k
R4
1k
FB
+
R5
5k
Q2
R2
5k
TO SYNC CIRCUIT
V
C
GND
1578 F02
Figure 2. Frequency and Current Limit Foldback
8
LT1578/LT1578-2.5
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APPLICATIONS INFORMATION
graphically in Typical Performance Characteristics and as
shown in the formula below:
In addition to lower switching frequency, the LT1578 also
operates at lower switch current limit when the feedback
pin voltage drops below 0.7V. Q2 in Figure 2 performs this
function by clamping the VC pin to a voltage less than its
normal 2.1V upper clamp level. This foldback current limit
greatly reduces power dissipation in the IC, diode and
inductorduringshort-circuitconditions.Externalsynchro-
nization is also disabled to prevent interference with
foldback operation. Again, it is nearly transparent to the
userundernormalloadconditions.Theonlyloadsthatmay
be affected are current sources, such as lamps and mo-
tors, that maintain high load current with output voltage
less than 50% of final value. In these rare situations the
feedbackpincanbeclampedabove0.7Vtodefeatfoldback
current limit. Caution: clamping the feedback pin means
thatfrequencyshiftingwillalsobedefeated,soacombina-
tion of high input voltage and dead shorted output may
cause the LT1578 to lose control of current limit.
IP = 1.5A for DC ≤ 50%
IP = 1.67 – 0.18 (DC) – 0.32(DC)2 for 50% < DC < 90%
DC = Duty cycle = VOUT/VIN
Example: with VOUT = 5V, VIN = 8V; DC = 5/8 = 0.625, and;
ISW(MAX) = 1.67 – 0.18 (0.625) – 0.32(0.625)2 = 1.43A
Current rating decreases with duty cycle because the
LT1578 has internal slope compensation to prevent cur-
rent mode subharmonic switching. For more details, read
Application Note 19. The LT1578 is a little unusual in this
regardbecauseithasnonlinearslopecompensationwhich
gives better compensation with less reduction in current
limit.
Maximum load current would be equal to maximum
switch current for an infinitely large inductor, but with
finite inductor size, maximum load current is reduced by
one-half peak-to-peak inductor current. The following
formula assumes continuous mode operation, implying
that the term on the right is less than one-half of IP.
The internal circuitry which forces reduced switching
frequency also causes current to flow out of the feedback
pin when output voltage is low. The equivalent circuitry is
shown in Figure 2. Q1 is completely off during normal
operation. If the FB pin falls below 0.7V, Q1 begins to
conduct current and reduces frequency at the rate of
approximately 1kHz/µA. To ensure adequate frequency
foldback (under worst-case short-circuit conditions), the
external divider Thevinin resistance must be low enough
V
V − V
IN OUT
(
OUT)(
)
IOUT(MAX)
=
IP −
Continuous Mode
2 L f V
( )( )( )
IN
to pull 35µA out of the FB pin with 0.5V on the pin (RDIV
≤
For the conditions above and L = 15µH,
14.3k). The net result is that reductions in frequency and
current limit are affected by output voltage divider imped-
ance. Although divider impedance is not critical, caution
should be used if resistors are increased beyond the
suggested values and short-circuit conditions will occur
with high input voltage. High frequency pickup will
increase and the protection accorded by frequency and
current foldback will decrease.
5 8 − 5
( )(
)
I
= 1.43 −
OUT MAX
(
)
−6
3
2 15• 10
200• 10
8
( )
=1.43 − 0.31= 1.12A
AtVIN =15V, dutycycleis33%, soIP isjustequaltoafixed
1.5A, and IOUT(MAX) is equal to:
MAXIMUM OUTPUT LOAD CURRENT
Maximum load current for a buck converter is limited by
the maximum switch current rating (IP) of the LT1578.
This current rating is 1.5A up to 50% duty cycle (DC),
decreasing to 1.3A at 80% duty cycle. This is shown
5 15 − 5
( )(
)
1.5 −
−6
3
2 15• 10
200• 10 15
( )
= 1.5 − 0.56 = 0.94A
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physical size of the inductor. Higher values allow more
output current because they reduce peak current seen by
the LT1578 switch, which has a 1.5A limit. Higher values
also reduce output ripple voltage, and reduce core loss.
GraphsintheTypicalPerformanceCharacteristicssection
show maximum output load current versus inductor size
and input voltage.
Note that there is less load current available at the higher
input voltage because inductor ripple current increases.
This is not always the case. Certain combinations of
inductor value and input voltage range may yield lower
available load current at the lowest input voltage due to
reduced peak switch current at high duty cycles. If load
current is close to the maximum available, please check
maximum available current at both input voltage
extremes. To calculate actual peak switch current with a
given set of conditions, use:
When choosing an inductor you might have to consider
maximum load current, core and copper losses, allowable
component height, output voltage ripple, EMI, fault cur-
rent in the inductor, saturation, and of course, cost. The
following procedure is suggested as a way of handling
thesesomewhatcomplicatedandconflictingrequirements.
VOUT V − V
(
)
IN
OUT
ISW PEAK =IOUT
+
(
)
2 L f V
( )( )( )
IN
1. Choose a value in microhenries from the graphs of
maximumloadcurrentandcoreloss.Choosingasmall
inductor may result in discontinuous mode operation
at lighter loads, but the LT1578 is designed to work
well in either mode. Keep in mind that lower core loss
means higher cost, at least for closed core geometries
like toroids.
For lighter loads where discontinuous operation can be
used, maximum load current is equal to:
2
I
f L V
IN
( ) ( )( )(
)
IOUT(MAX)
Discontinuous mode
=
P
2 V
V − V
(
)(
)
OUT
IN
OUT
Assume that the average inductor current is equal to
load current and decide whether or not the inductor
must withstand continuous fault conditions. If maxi-
mum load current is 0.5A, for instance, a 0.5A inductor
may not survive a continuous 1.5A overload condition.
Dead shorts will actually be more gentle on the induc-
tor because the LT1578 has foldback current limiting.
Example: with L = 5µH, VOUT = 5V, and VIN(MAX) = 15V,
2
3
−6
1.5 200• 10 5• 10
15
( )
(
)
I
=
= 0.34A
OUT MAX
(
)
2 5 15 − 5
( )(
)
2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, espe-
cially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving
because they saturate softly, whereas ferrite cores
saturate abruptly. Other core materials fall somewhere
in between. The following formula assumes continu-
ous mode of operation, but it errs only slightly on the
high side for discontinuous mode, so it can be used for
all conditions.
The main reason for using such a tiny inductor is that it is
physically very small, but keep in mind that peak-to-peak
inductorcurrentwillbeveryhigh. Thiswillincreaseoutput
ripplevoltage.Iftheoutputcapacitorhastobemadelarger
to reduce ripple voltage, the overall circuit could actually
wind up larger.
CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR
For most applications the output inductor will fall in the
rangeof15µHto60µH. Lowervaluesarechosentoreduce
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Table 1
VOUT V − V
(
)
IN
OUT
SERIES
CORE
IPEAK =IOUT +
VENDOR/
PART NO.
VALUE
DC
CORE RESIS- MATER- HEIGHT
2 f L V
( )( )( )
VIN = Maximum input voltage
f = Switching frequency, 200kHz
IN
(µ
H) (Amps) TYPE TANCE(
Ω)
IAL
(mm)
Coiltronics
CTX15-2
15
33
68
15
33
68
1.7
1.4
1.2
1.4
1.3
1.1
Tor
Tor
Tor
Tor
Tor
Tor
0.059
KMµ
KMµ
KMµ
52
6.0
6.0
6.4
4.2
6.0
6.4
CTX33-2
0.106
0.158
0.087
0.126
0.238
3. Decide if the design can tolerate an “open” core geom-
etry like a rod or barrel, with high magnetic field
radiation, orwhetheritneedsaclosedcorelikeatoroid
to prevent EMI problems. One would not want an open
core next to a magnetic storage media, for instance!
Thisisatoughdecisionbecausetherodsorbarrelsare
temptingly cheap and small and there are no helpful
guidelines to calculate when the magnetic field radia-
tion will be a problem.
CTX68-4
CTX15-1P
CTX33-2P
52
CTX68-4P
52
Sumida
CDRH74-150
CDH115-330
CDRH125-680
CDH74-330
Coilcraft
15
33
68
33
1.47
1.68
1.5
SC
SC
SC
SC
0.081
0.082
0.12
Fer
Fer
Fer
Fer
4.5
5.2
6
1.45
0.17
5.2
4. Start shopping for an inductor (see representative
surface mount units in Table 1) which meets the
requirements of core shape, peak current (to avoid
saturation),averagecurrent(tolimitheating),andfault
current(iftheinductorgetstoohot, wireinsulationwill
melt and cause turn-to-turn shorts). Keep in mind that
allgoodthingslikehighefficiency,lowprofile,andhigh
temperature operation will increase cost, sometimes
dramatically. Get a quote on the cheapest unit first to
calibrate yourself on price, then ask for what you really
want.
DO3308P-153
DO3316P-333
DO3316P-683
Pulse
15
33
68
2
2
SC
SC
SC
0.12
0.1
Fer
Fer
Fer
3
5.21
5.21
1.4
0.18
PE-53602
35
73
22
40
1.4
1.3
2.7
2.7
Tor
Tor
Tor
Tor
0.166
0.290
0.063
0.085
Fer
Fer
Fer
Fer
9.1
9.1
9.1
10
PE-53604
PE-53632
PE-53633
Gowanda
SMP3316-152K
SMP3316-332K
SMP3316-682K
Tor = Toroid
SC = Semi-closed geometry
Fer = Ferrite core material
15
33
68
3.5
2.3
1.7
SC
SC
SC
0.041
0.092
0.178
Fer
Fer
Fer
6
6
6
5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the Linear Technology’s applica-
tions department if you feel uncertain about the final
choice. They have experience with a wide range of
inductor types and can tell you about the latest devel-
opments in low profile, surface mounting, etc.
52 = Type 52 powdered iron core material
KMµ = Kool Mµ
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Output Capacitor
Output Capacitor Ripple Current (RMS):
The output capacitor is normally chosen by its Effective
Series Resistance (ESR), because this is what determines
output ripple voltage. To get low ESR takes volume, so
physically smaller capacitors have high ESR. The ESR
range for typical LT1578 applications is 0.05Ω to 0.2Ω. A
typical output capacitor is an AVX type TPS, 100µF at 10V,
with a guaranteed ESR less than 0.1Ω. This is a “D” size
surface mount solid tantalum capacitor. TPS capacitors
are specially constructed and tested for low ESR, so they
give the lowest ESR for a given volume. The value in
microfarads is not particularly critical, and values from
22µF to greater than 500µF work well, but you cannot
cheat mother nature on ESR. If you find a tiny 22µF solid
tantalumcapacitor, itwillhavehighESR, andoutputripple
voltage will be terrible. Table 2 shows some typical solid
tantalum surface mount capacitors.
0.29 V
V − V
IN OUT
(
OUT)(
)
IRIPPLE RMS
=
(
)
L f V
( )( )(
)
IN
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becomingavailableinsmallercasesizes.Thesearetempt-
ing for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor’s ESR
generatesaloop“zero”at5kHzto50kHzthatisinstrumen-
tal in giving acceptable loop phase margin. Ceramic
capacitors remain capacitive to beyond 300kHz and usu-
ally resonate with their ESL before their ESR provides any
damping. They are appropriate for input bypassing be-
cause of their high ripple current ratings and tolerance of
turn-on surges.
Table 2. Surface Mount Solid Tantalum Capacitor ESR
and Ripple Current
E Case Size
ESR (Max.,
Ω
)
Ripple Current (A)
0.7 to 1.1
0.4
AVX TPS, Sprague 593D
AVX TAJ
0.1 to 0.3
0.7 to 0.9
OUTPUT RIPPLE VOLTAGE
D Case Size
Figure 3 shows a typical output ripple voltage waveform
for the LT1578. Ripple voltage is determined by the high
frequency impedance of the output capacitor, and ripple
current through the inductor. Peak-to-peak ripple current
through the inductor into the output capacitor is:
AVX TPS, Sprague 593D
C Case Size
0.1 to 0.3
0.2 (typ)
0.7 to 1.1
0.5 (typ)
AVX TPS
Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true, and type TPS capacitors are
speciallytestedforsurgecapability,butsurgeruggedness
is not a critical issue with the output capacitor. Solid
tantalum capacitors fail during very high turn-on surges,
which do not occur at the output of regulators. High
discharge surges, such as when the regulator output is
dead shorted, do not harm the capacitors.
V
V − V
IN OUT
(
OUT)(
)
IP-P
=
V
L f
IN)( )( )
(
For high frequency switchers, the sum of ripple current
slew rates may also be relevant and can be calculated
from:
dI
dt
V
IN
L
Σ
=
Unlike the input capacitor, RMS ripple current in the
output capacitor is normally low enough that ripple cur-
rent rating is not an issue. The current waveform is
triangular with a typical value of 200mARMS. The formula
to calculate this is:
12
LT1578/LT1578-2.5
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Peak-to-peak output ripple voltage is the sum of a triwave
created by peak-to-peak ripple current times ESR, and a
square wave created by parasitic inductance (ESL) and
ripple current slew rate. Capacitive reactance is assumed
to be small compared to ESR or ESL.
IOUT V − V
(
)
IN
OUT
ID(AVG
=
)
V
IN
This formula will not yield values higher than 1A with
maximumloadcurrentof1.25Aunlesstheratioofinputto
output voltage exceeds 5:1. The only reason to consider a
larger diode is the worst-case condition of a high input
voltageandoverloaded(notshorted)output. Undershort-
circuit conditions, foldback current limit will reduce diode
current to less than 1A, but if the output is overloaded and
does not fall to less than 1/3 of nominal output voltage,
foldback will not take effect. With the overloaded condi-
tion, output current will increase to a typical value of 1.8A,
determined by peak switch current limit of 2A. With
VIN = 15V, VOUT = 4V (5V overloaded) and IOUT = 1.8A:
dI
dt
VRIPPLE = I
ESR + ESL Σ
(
P-P)(
) (
)
Example: withVIN =10V,VOUT =5V,L=30µH,ESR=0.1Ω,
ESL = 10nH:
5 10 − 5
( )(
)
I
=
= 0.42A
P-P
−6
3
10 30• 10
200• 10
( )
dI
dt
10
6
Σ
=
= 0.33• 10
−6
1.8 15 − 4
30• 10
(
)
I
=
= 1.32A
D AVG
−9
6
(
)
V
= 0.42A 0.1 + 10• 10
0.33• 10
15
(
)( )
RIPPLE
This is safe for short periods of time, but it would be
prudent to check with the diode manufacturer if continu-
ous operation under these conditions must be tolerated.
= 0.042 + 0.003 = 45mV
P-P
20mV/DIV
VOUT AT
IOUT = 1A
BOOST PIN CONSIDERATIONS
INDUCTOR
CURRENT
AT IOUT = 1A
200mA/DIV
Formostapplications, theboostcomponentsarea0.33µF
capacitor and a 1N914 or 1N4148 diode. The anode is
connected to the regulated output voltage and this gener-
ates a voltage across the boost capacitor nearly identical
to the regulated output. In certain applications, the anode
may instead be connected to the unregulated input volt-
age. This could be necessary if the regulated output
voltage is very low (< 3V) or if the input voltage is less than
6V. Efficiencyisnotaffectedbythecapacitorvalue, butthe
capacitor should have an ESR of less than 1Ω to ensure
that it can be recharged fully under the worst-case condi-
tion of minimum input voltage. Almost any type of film or
ceramic capacitor will work fine.
20mV/DIV
VOUT AT
IOUT = 50mA
INDUCTOR
200mA/DIV
CURRENT
AT IOUT = 50mA
2µs/DIV
1578 F03
Figure 3. LT1578 Ripple Voltage Waveform
CATCH DIODE
The suggested catch diode (D1) is a 1N5818 Schottky, or
its Motorola equivalent, MBR130. It is rated at 1A average
forward current and 30V reverse voltage. Typical forward
voltage is 0.42V at 1A. The diode conducts current only
during switch off time. Peak reverse voltage is equal to
regulatorinputvoltage.Averageforwardcurrentinnormal
operation can be calculated from:
WARNING! Peak voltage on the BOOST pin is the sum of
unregulated input voltage plus the voltage across the
13
LT1578/LT1578-2.5
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boost capacitor. This normally means that peak BOOST
pin voltage is equal to input voltage plus output voltage,
but when the boost diode is connected to the regulator
input, peak BOOST pin voltage is equal to twice the input
voltage. Be sure that BOOST pin voltage does not exceed
its maximum rating.
I
/50 VOUT / V
(
)(
)
OUT
IN
CMIN
=
f V −3V
( )(
)
OUT
f = Switching frequency
V
OUT = Regulated output voltage
VIN = Minimum input voltage
Fornearlyallapplications, a0.33µFboostcapacitorworks
just fine, but for the curious, more details are provided
here. The size of the boost capacitor is determined by
switch drive current requirements. During switch on time,
draincurrentonthecapacitorisapproximatelyIOUT/50.At
peakloadcurrentof1.25A,thisgivesatotaldrainof25mA.
Capacitor ripple voltage is equal to the product of on time
and drain current divided by capacitor value;
∆V = (tON)(25mA/C). To keep capacitor ripple voltage to
less than 0.5V (a slightly arbitrary number) at the worst-
case condition of tON = 4.7µs, the capacitor needs to be
0.24µF. Boost capacitor ripple voltage is not a critical
parameter, but if the minimum voltage across the capaci-
tor drops to less than 3V, the power switch may not
saturate fully and efficiency will drop. An approximate
formula for absolute minimum capacitor value is:
This formula can yield capacitor values substantially less
than 0.24µF, but it should be used with caution since it
does not take into account secondary factors such as
capacitor series resistance, capacitance shift with tem-
perature and output overload.
SHUTDOWN FUNCTION AND
UNDERVOLTAGE LOCKOUT
Figure 4 shows how to add undervoltage lockout (UVLO)
to the LT1578. Typically, UVLO is used in situations where
the input supply is current limited, or has a relatively high
source resistance. It is particularly useful for input sup-
plies with foldback current limiting. A switching regulator
draws constant power from the source, so source current
increases as source voltage drops. This looks like a
negative resistance load to the source and can cause the
source to current limit and latch under low source voltage
R
FB
LT1578
OUTPUT
V
SW
IN
INPUT
2.42V
–
+
STANDBY
R
HI
3.5µA
+
SHDN
+
–
TOTAL
SHUTDOWN
R
LO
C1
0.4V
GND
1578 F04
Figure 4. Undervoltage Lockout
14
LT1578/LT1578-2.5
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conditions. UVLO helps prevent the regulator from oper-
ating at source voltages where these problems might
occur.
input rises back to 13.5V. ∆V is therefore 1.5V and
VIN = 12V. Let RLO = 25k.
Threshold voltage for lockout is about 2.42V. A 3.5µA bias
current flows out of the pin at threshold. This internally
generated current is used to force a default high state on
the shutdown pin if the pin is left open. When low shut-
down current is not an issue, the error due to this current
can be minimized by making RLO 10k or less. If shutdown
currentisanissue, RLO canberaisedto100k, buttheerror
due to initial bias current and changes with temperature
should be considered.
25k 12 − 2. 1.5 /5 +1 + 1.5
42
(
)
[
]
RHI =
2.42 − 25k 3.5µA
(
)
25k 10.35
(
)
=
=111k
2.33
RFB = 111k 5 /1.5 = 370k
(
)
SWITCH NODE CONSIDERATIONS
R
LO = 10k to 100k 25k suggested
(
)
For maximum efficiency, switch rise and fall times are
made as short as possible. To prevent radiated EMI and
high frequency resonance problems, proper layout of the
components connected to the switch node is essential. B
field (magnetic) radiation is minimized by keeping catch
diode, switch pin, and input bypass capacitor leads as
short as possible. E field radiation is kept low by minimiz-
ingthelengthandareaofalltracesconnectedtotheswitch
pinandBOOSTpin. Agroundplaneshouldalwaysbeused
under the switcher circuitry to prevent interplane cou-
pling. A suggested layout for the critical components is
shown in Figure 5. Note that the feedback resistors and
compensation components are kept as far as possible
from the switch node. Also note that the high current
groundpathofthecatchdiodeandinputcapacitorarekept
very short and separate from the analog ground line.
RLO V − 2.42V
(
IN
)
RHI =
2.42V − RLO 3.5µA
(
)
VIN = Minimum input voltage
Keep the connections from the resistors to the shutdown
pin short and make sure that interplane or surface capaci-
tance to the switching nodes are minimized. If high resis-
tor values are used, the shutdown pin should be bypassed
with a 1000pF capacitor to prevent coupling problems
from the switch node. If hysteresis is desired in the
undervoltage lockout point, a resistor RFB can be added to
the output node. Resistor values can be calculated from:
RLO V − 2. ∆V /V
+1 + ∆V
42
IN
(
OUT
)
[
]
Thehighspeedswitchingcurrentpathisshownschemati-
cally in Figure 6. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, catch diode, and input capacitor is
the only one containing nanosecond rise and fall times. If
you follow this path on the PC layout, you will see that it is
irreducibly short. If you move the diode or input capacitor
away from the LT1578, get your resumé in order. The
other paths contain only some combination of DC and
200kHz triwave, so are much less critical.
RHI =
R
2.42 −
3.5µA
LO
(
)
RFB = RHI VOUT
/
∆V
(
)(
)
25k suggested for RLO
VIN = Input voltage at which switching stops as input
voltage descends to trip level
∆V = Hysteresis in input voltage level
Example: output voltage is 5V, switching is to stop if input
voltage drops below 12V and should not restart unless
15
LT1578/LT1578-2.5
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TAKE OUTPUT DIRECTLY FROM END
CONNECT OUTPUT
CAPACITOR DIRECTLY
TO HEAVY GROUND
OF OUTPUT CAPACITOR TO AVOID
PARASITIC RESISTANCE AND
INDUCTANCE (KELVIN CONNECTION)
C1
V
OUT
MINIMIZE AREA
OF CONNECTIONS
TO SWITCH NODE
AND BOOST NODE
L1
D2
MINIMIZE SIZE
OF FEEDBACK PIN
CONNECTIONS
C2
TO AVOID PICKUP
SW
IN
KEEP INPUT
CAPACITOR
AND CATCH
SYNC
SHDN
D1
C3
TERMINATE
FEEDBACK
V
DIODE CLOSE
TO REGULATOR
AND TERMINATE
THEM TO THE
SAME POINT
RESISTORS AND
COMPENSATION
COMPONENTS
DIRECTLY TO
SWITCHER
BOOST
FB
R2
R
V
C
C
C
C
R1
GROUND PIN
GND
GND
GROUND RING NEED NOT BE AS SHOWN
(NORMALLY EXISTS AS INTERNAL PLANE)
1578 F05
Figure 5. Suggested Layout for LT1578
SWITCH NODE
L1
5V
HIGH
FREQUENCY
CIRCULATING
PATH
V
IN
LOAD
1578 F06
Figure 6. High Speed Switching Path
16
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PARASITIC RESONANCE
When looking at this, a >100MHz oscilloscope must be
used, and waveforms should be observed on the leads of
the package. This switch off spike will also cause the SW
nodetogobelowground.TheLT1578hasspecialcircuitry
insidewhichmitigatesthisproblem, butnegativevoltages
over 1V lasting longer than 10ns should be avoided. Note
that 100MHz oscilloscopes are barely fast enough to see
the details of the falling edge overshoot in Figure 7.
Resonance or “ringing” may sometimes be seen on the
switch node (see Figure 7). Very high frequency ringing
following the switch voltage rise time is caused by switch/
diode/input capacitance lead inductance and diode ca-
pacitance. Schottky diodes have very high “Q” junction
capacitance that can ring for many cycles when excited at
high frequency. If total lead length for the input capacitor,
diode and switch path is 1 inch, the inductance will be
approximately 25nH. At switch off, this will produce a
spike across the NPN output device in addition to the input
voltage. At higher currents this spike can be in the order of
10V to 20V or higher with a poor layout, potentially
exceeding the absolute max switch voltage. The path
around switch, catch diode and input capacitor must be
kept as short as possible to ensure reliable operation.
A second, much lower frequency ringing is seen during
switch off time if load current is low enough to allow the
inductor current to fall to zero during part of the switch off
time (see Figure 8). Switch and diode capacitance reso-
nate with the inductor to form damped ringing at 1MHz to
10 MHz. This ringing is not harmful to the regulator and it
hasnotbeenshowntocontributesignificantlytoEMI. Any
attempttodampitwithanRCsnubberwillslightlydegrade
efficiency.
INPUT BYPASSING AND VOLTAGE RANGE
Input Bypass Capacitor
RISE AND FALL
WAVEFORMS ARE
SUPERIMPOSED
(PULSE WIDTH IS
Step-down converters draw current from the input supply
in pulses. The average height of these pulses is equal to
load current, and the duty cycle is equal to VOUT/VIN. Rise
and fall times of the current are very fast. A local bypass
capacitor across the input supply is necessary to ensure
proper operation of the regulator and minimize the ripple
current fed back into the input supply. The capacitor also
forces switching current to flow in a tight local loop,
minimizing EMI.
NOT 350ns)
5V/DIV
50ns/DIV
1578 F07
Figure 7. Switch Node Response
Do not cheat on the ripple current rating of the input
bypasscapacitor,butalsodonotbeoverlyconcernedwith
the value in microfarads. The input capacitor is intended
to absorb all the switching current ripple, which can have
anRMSvalueashighasonehalfoftheloadcurrent.Ripple
current ratings on the capacitor must be observed to
ensure reliable operation. In many cases it is necessary to
parallel two capacitors to obtain the required ripple rating.
Both capacitors must be of the same value and manufac-
turer to guarantee power sharing. The actual value of the
capacitor in microfarads is not particularly important
5V/DIV
SWITCH NODE
VOLTAGE
50mA/DIV
INDUCTOR
CURRENT
1µs/DIV
1578 F08
Figure 8. Discontinuous Mode Ringing
17
LT1578/LT1578-2.5
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because at 200kHz, any value above 15µF is essentially
resistive. RMS ripple current rating is the critical param-
eter. Actual RMS current can be calculated from:
series for instance, see Table 3), but even these units may
fail if the input voltage surge approaches the maximum
voltage rating of the capacitor. AVX recommends derating
capacitor voltage by 2:1 for high surge applications. The
highest voltage rating is 50V, so 25V may be a practical
input voltage upper limit when using solid tantalum ca-
pacitors for input bypassing.
2
I
=I
V
V − V
/V
IN
(
)
RIPPLE RMS
OUT OUT IN
OUT
(
)
The term inside the radical has a maximum value of 0.5
when input voltage is twice output, and stays near 0.5 for
a relatively wide range of input voltages. It is common
practice therefore to simply use the worst-case value and
assumethatRMSripplecurrentisonehalfofloadcurrent.
At maximum output current of 1.5A for the LT1578, the
input bypass capacitor should be rated at 0.75A ripple
current. Note however, that there are many secondary
considerations in choosing the final ripple current rating.
These include ambient temperature, average versus peak
load current, equipment operating schedule, and required
product lifetime. For more details, see Application Notes
19 and 46, and Design Note 95.
Larger capacitors may be necessary when the input volt-
age is very close to the minimum specified on the data
sheet. Small voltage dips during switch on time are not
normallyaproblem, butatverylowinputvoltagetheymay
cause erratic operation because the input voltage drops
below the minimum specification. Problems can also
occur if the input-to-output voltage differential is near
minimum. The amplitude of these dips is normally a
function of capacitor ESR and ESL because the capacitive
reactance is small compared to these terms. ESR tends to
be the dominate term and is inversely related to physical
capacitor size within a given capacitor type.
SYNCHRONIZING
Input Capacitor Type
TheSYNCpinisusedtosynchronizetheinternaloscillator
to an external signal. The SYNC input must pass from a
logic level low, through the maximum synchronization
threshold with a duty cycle between 10% and 90%. The
input can be driven directly from a logic level output. The
synchronizing range is equal to initial operating frequency
up to 400kHz. This means that minimum practical sync
frequency is equal to the worst-case high self-oscillating
frequency(250kHz),notthetypicaloperatingfrequencyof
200kHz. Caution should be used when synchronizing
above 280kHz because at higher sync frequencies the
amplitude of the internal slope compensation used to
prevent subharmonic switching is reduced. This type of
subharmonic switching only occurs at input voltages less
than twice output voltage. Higher inductor values will tend
to eliminate this problem. See Frequency Compensation
section for a discussion of an entirely different cause of
subharmonic switching before assuming that the cause is
insufficient slope compensation. Application Note 19 has
more details on the theory of slope compensation.
Some caution must be used when selecting the type of
capacitor used at the input to regulators. Aluminum
electrolytics are lowest cost, but are physically large to
achieve adequate ripple current rating, and size con-
straints (especially height) may preclude their use.
Ceramic capacitors are now available in larger values, and
their high ripple current and voltage rating make them
ideal for input bypassing. Cost is fairly high and footprint
may also be somewhat large. Solid tantalum capacitors
would be a good choice, except that they have a history of
occasionalspectacularfailureswhentheyaresubjectedto
large current surges during power-up. The capacitors can
short and then burn with a brilliant white light and lots of
nasty smoke. This phenomenon occurs in only a small
percentage of units, but it has led some OEMs to forbid
their use in high surge applications. The input bypass
capacitors of regulators can see these high surges when
abatteryorhighcapacitancesourceisconnected. Several
manufacturers have developed a line of solid tantalum
capacitors specially tested for surge capability (AVX TPS
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At power-up, when VC is being clamped by the FB pin (see
Figure2,Q2),thesyncfunctionisdisabled.Thisallowsthe
frequency foldback to operate in the shorted output con-
dition. During normal operation, switching frequency is
controlledbytheinternaloscillatoruntiltheFBpinreaches
0.7V, after which the SYNC pin becomes operational. If no
synchronization is required, this pin should be connected
to ground.
2
0.2 1 5
(
)( ) ( )
−9
3
P
=
+ 60• 10
1 10 200• 10
( )( )
SW
10
= 0.1 + 0.12 = 0.22W
2
5 1/50
( ) (
)
P
=
= 0.05W
BOOST
10
2
5 0.004
( ) (
)
−3
−3
P =10 0.55• 10
+5 1.6• 10
+
Q
THERMAL CALCULATIONS
10
Power dissipation in the LT1578 chip comes from four
sources: switch DC loss, switch AC loss, boost circuit
current,andinputquiescentcurrent.Thefollowingformu-
las show how to calculate each of these losses. These
formulas assume continuous mode operation, so they
should not be used for calculating efficiency at light load
currents.
= 0.02W
Total power dissipation is 0.22 + 0.05 + 0.02 = 0.29W.
Thermal resistance for LT1578 package is influenced by
the presence of internal or backside planes. With a full
plane under the SO package, thermal resistance will be
about 80°C/W. No plane will increase resistance to about
120°C/W. To calculate die temperature, add in worst-case
ambient temperature:
Switch loss:
2
R
I
V
OUT
(
) (
)
SW OUT
TJ = TA + θJA (PTOT
)
P
=
+ 60ns I
V
f
(
)( )( )
SW
OUT IN
V
IN
With the SO-8 package (θJA = 80°C/W), at an ambient
temperature of 50°C,
Boost current loss:
TJ = 50 + 80 (0.29) = 73.2°C
2
V
I
/50
(
)
OUT OUT
Die temperature is highest at low input voltage, so use
lowest continuous input operating voltage for thermal
calculations.
P
=
BOOST
V
IN
Quiescent current loss:
FREQUENCY COMPENSATION
−3
−3
P = V 0.55• 10
+ V
1.6• 10
Q
IN
OUT
Loop frequency compensation of switching regulators
can be a rather complicated problem because the reactive
components used to achieve high efficiency also intro-
duce multiple poles into the feedback loop. The inductor
and output capacitor on a conventional step-down con-
verter actually form a resonant tank circuit that can exhibit
peaking and a rapid 180° phase shift at the resonant
frequency. Bycontrast, theLT1578usesa“currentmode”
architecture to help alleviate the phase shift created by the
inductor. The basic connections are shown in Figure 9.
Figure 10 shows a Bode plot of the phase and gain of the
power section of the LT1578, measured from the VC pin to
2
V
0.004
(
)
OUT
+
V
IN
RSW = Switch resistance (≈0.2Ω)
60ns = Equivalent switch current/voltage overlap time
f = Switch frequency
Example: with VIN = 10V, VOUT = 5V and IOUT = 1A:
19
LT1578/LT1578-2.5
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the output. Gain is set by the 1.5A/V transconductance of
the LT1578 power section and the effective complex
impedance from output to ground. Gain rolls off smoothly
above the 160Hz pole frequency set by the 100µF output
capacitor. Phase drop is limited to about 85°. Phase
recoversandgainlevelsoffatthezerofrequency(≈16kHz)
set by capacitor ESR (0.1Ω).
This means that the error amplifier characteristics them-
selvesdonotcontributeexcessphaseshifttotheloop,and
the phase/gain characteristics of the error amplifier sec-
tion are completely controlled by the external compensa-
tion network.
In Figure 12, full loop phase/gain characteristics are
shownwithacompensationcapacitorof100pF, givingthe
error amplifier a pole at 2.8kHz, with phase rolling off to
90° and staying there. The overall loop has a gain of 66dB
at low frequency, rolling off to unity-gain at 58kHz. The
phase plot shows a two-pole characteristic until the ESR
of the output capacitor brings it back to single pole above
16kHz. Phase margin is about 77° at unity-gain.
Erroramplifiertransconductancephaseandgainareshown
in Figure 11. The error amplifier can be modeled as a
transconductance of 1000µMho, with an output imped-
ance of 570kΩ in parallel with 2.4pF. In all practical
applications,thecompensationnetworkfromtheVC pinto
ground has a much lower impedance than the output
impedance of the amplifier at frequencies above 200Hz.
2000
1500
1000
500
200
150
100
50
LT1578
CURRENT MODE
POWER STAGE
V
SW
FB
PHASE
GAIN
OUTPUT
ERROR
g
= 1.5A/V
m
AMPLIFIER
R1
R2
–
V
C
ESR
C1
+
1.21V
C
R
OUT
2.4pF
–3
OUT
570k
V
1 × 10
(
)
+
FB
V
C
GND
0
ERROR AMPLIFIER EQUIVALENT CIRCUIT
= 50Ω
0
R
C
R
LOAD
C
F
–500
–50
C
10
100
1k
10k
100k
1M
C
FREQUENCY (Hz)
1578 F11
1578 F09
Figure 9. Model for Loop Response
Figure 11. Error Amplifier Gain and Phase
80
60
40
20
0
180
135
90
40
40
V
V
I
= 10V
IN
= 5V
OUT
OUT
= 500mA
20
0
0
GAIN
PHASE
PHASE
–40
–80
–120
V
V
= 10V
IN
45
= 5V
OUT
OUT
OUT
I
= 500mA
= 100µF
GAIN
C
–20
–40
0
10V, AVX TPS
C
= 100pF
C
L = 30µH
–20
–45
1M
10
100
1k
10k
100k
10
100
1k
FREQUENCY (Hz)
10k
100k
FREQUENCY (Hz)
1578 F12
1578 F07
Figure 10. Response from VC Pin to Output
Figure 12. Overall Loop Characteristics
20
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Analog experts will note that around 7kHz, phase dips
close to the zero phase margin line. This is typical of
switching regulators, especially those that operate over a
wide range of loads. This region of low phase is not a
problem as long as it does not occur near unity-gain. In
practice, the variability of output capacitor ESR tends to
dominate all other effects with respect to loop response.
Variations in ESR will cause unity-gain to move around,
but at the same time phase moves with it so that adequate
phase margin is maintained over a very wide range of ESR
(≥ ±3:1).
evidenced by alternating pulse widths seen at the switch
node. In more severe cases, the regulator squeals or
hisses audibly even though the output voltage is still
roughly correct. None of this will show on a Bode plot
since this is an amplitude insensitive measurement. Tests
have shown that if ripple voltage on the VC is held to less
than100mVP-P,theLT1578willgenerallybewellbehaved.
The formula below will give an estimate of VC ripple
voltage when RC is added to the loop, assuming that RC is
large compared to the reactance of CC at 200kHz.
R G
V − V
ESR 1.21
( )(
)(
)(
)(
)
C
MA IN
OUT
What About a Resistor in the Compensation Network?
V
=
C RIPPLE
(
)
V
L f
(
)( )( )
IN
It is common practice in switching regulator design to add
a “zero” to the error amplifier compensation to increase
loop phase margin. This zero is created in the external
network in the form of a resistor (RC) in series with the
compensation capacitor. Increasing the size of this resis-
tor generally creates better and better loop stability, but
there are two limitations on its value. First, the combina-
tion of output capacitor ESR and a large value for RC may
cause loop gain to stop rolling off altogether, creating a
gain margin problem. An approximate formula for RC
where gain margin falls to zero is:
GMA = Error amplifier transconductance (1000µMho)
If a series compensation resistor of 15k gave the best
overall loop response, with adequate gain margin, the
resulting VC pin ripple voltage with VIN = 10V, VOUT = 5V,
ESR = 0.1Ω, L = 30µH, would be:
15k 1•10−3 10 − 5 0.1 1.21
(
)
(
)( )(
)
(
)
VC(RIPPLE
=
= 0.151V
)
10 30 •10−6 200 •103
This ripple voltage is high enough to possibly create
subharmonic switching. In most situations a compromise
value (<10k in this case) for the resistor gives acceptable
phase margin and no subharmonic problems. In other
cases, the resistor may have to be larger to get acceptable
phaseresponse, andsomemeansmustbeusedtocontrol
ripple voltage at the VC pin. The suggested way to do this
istoaddacapacitor(CF)inparallelwiththeRC/CC network
on the VC pin. The pole frequency for this capacitor is
typically set at one-fifth of the switching frequency so that
it provides significant attenuation of the switching ripple,
but does not add unacceptable phase shift at the loop
unity-gain frequency. With RC = 15k,
V
OUT
R Loop Gain = 1 =
(
)
C
G
(
G
)(
ESR 1.21
)(
)(
)
MP MA
GMP = Transconductance of power stage = 1.5A/V
GMA = Error amplifier transconductance = 1(10–3)
ESR = Output capacitor ESR
1.21 = Reference voltage
With VOUT = 5V and ESR = 0.1Ω, a value of 27.5k for RC
would yield zero gain margin, so this represents an upper
limit. There is a second limitation however which has
nothing to do with theoretical small signal dynamics. This
resistor sets high frequency gain of the error amplifier,
including the gain at the switching frequency. If the
switching frequency gain is high enough, an excessive
amout of output ripple voltage will appear at the VC pin
resulting in improper operation of the regulator. In a
marginal case, subharmonic switching occurs, as
5
5
CF =
=
= 265pF
2π 200 •103 15k
2π f RC
(
)( )(
)
(
)
(
)
21
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How Do I Test Loop Stability?
One way to check switching regulator loop stability is by
pulse loading the regulator output while observing the
transient response at the output, using the circuit shown
in Figure 13. The regulator loop is “hit” with a small
transient AC load current at a relatively low frequency,
50Hz to 1kHz. This causes the output to jump a few
millivolts, then settle back to the original value, as shown
in Figure 14. A well behaved loop will settle back cleanly,
whereas a loop with poor phase or gain margin will “ring”
as it settles. The number of rings indicates the degree of
stability, and the frequency of the ringing shows the
approximate unity-gain frequency of the loop. Amplitude
of the signal is not particularly important, as long as the
amplitude is not so high that the loop behaves nonlinearly.
The “standard” compensation for LT1578 is a 100pF
capacitor for CC, with RC = 0. While this compensation will
work for most applications, the “optimum” value for loop
compensation components depends, to various extents,
on parameters which are not well controlled. These in-
clude inductor value (±30% due to production tolerance,
load current and ripple current variations), output capaci-
tance (±20% to ±50% due to production tolerance,
temperature, aging and changes at the load), output
capacitor ESR (±200% due to production tolerance,
temperature and aging), and finally, DC input voltage and
output load current . This makes it important for the
designer to check out the final design to ensure that it is
“robust” and tolerant of all these variations.
RIPPLE FILTER
TO X1
OSCILLOSCOPE
PROBE
470Ω
4.7k
SWITCHING
REGULATOR
+
100µF TO
1000µF
3300pF
330pF
50Ω
ADJUSTABLE
INPUT SUPPLY
ADJUSTABLE
DC LOAD
TO
OSCILLOSCOPE
SYNC
100Hz TO 1kHz
100mV TO 1V
P-P
1578 F13
Figure 13. Loop Stability Test Circuit
V
OUT AT
IOUT = 500mA
BEFORE FILTER
V
OUT AT
IOUT = 500mA
AFTER FILTER
VOUT AT
IOUT = 50mA
AFTER FILTER
LOAD PULSE
THROUGH 50Ω
f ≈ 780Hz
10mV/DIV
5A/DIV
0.2ms/DIV
1578 F14
Figure 14. Loop Stability Check
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The output of the regulator contains both the desired low
frequency transient information and a reasonable amount
of high frequency (200kHz) ripple. The ripple makes it
difficult to observe the small transient, so a two-pole,
100kHz filter has been added. This filter is not particularly
critical; even if it attenuated the transient signal slightly,
this wouldn’t matter because amplitude is not critical.
lightloadsisnotparticularlysensitivetocomponentvaria-
tion, so if it looks reasonable under a transient test, it will
probably not be a problem in production. Note that fre-
quency of the light load ringing may vary with component
tolerance but phase margin generally hangs in there.
POSITIVE-TO-NEGATIVE CONVERTER
After verifying that the setup is working correctly, start
varyingloadcurrentandinputvoltagetoseeifyoucanfind
any combination that makes the transient response look
suspiciously “ringy.” This procedure may lead to an ad-
justment for best loop stability or faster loop transient
response. Nearly always you will find that loop response
looks better if you add in several kΩ for RC. Do this only
if necessary, because as explained before, RC above 1k
may require the addition of CF to control VC pin ripple.
If everything looks OK, use a heat gun and cold spray on
the circuit (especially the output capacitor) to bring out
any temperature-dependent characteristics.
The circuit in Figure 15 is a classic positive-to-negative
topology using a grounded inductor. It differs from the
standard approach in the way the IC chip derives its
feedback signal. Because the LT1578 accepts only posi-
tive feedback signals, the ground pin must be tied to the
regulated negative output. A resistor divider to ground or,
in this case, the sense pin, then provides the proper
feedback voltage for the chip.
D1
1N4148
C2
0.33µF
L1*
15µH
Keep in mind that this procedure does not take initial
component tolerance into account. You should see fairly
cleanresponseunderallloadandlineconditionstoensure
that component variations will not cause problems. One
note here: according to Murphy, the component most
likely to be changed in production is the output capacitor,
because that is the component most likely to have manu-
facturer variations (in ESR) large enough to cause prob-
lems. It would be a wise move to lock down the sources of
the output capacitor in production. Also, try varying com-
ponent values by a factor of 2 and see if the behavior is still
acceptable. Double and halve the values of RC and CC and
output capacitors. If the regulator still works correctly, it
will likely be good in production.
INPUT
5.5V TO
15V
BOOST
LT1578
V
V
IN
SW
R1
15.8k
FB
+
C3
10µF TO
50µF
GND
V
C
C1
+
R2
100µF
10V TANT
×2
4.99k
C
C
D2
1N5818
R
C
OUTPUT**
–5V, 0.5A
* INCREASE L1 TO 30µH OR 60µH FOR HIGHER CURRENT APPLICATIONS.
SEE APPLICATIONS INFORMATION
** MAXIMUM LOAD CURRENT DEPENDS ON MINIMUM INPUT VOLTAGE
AND INDUCTOR SIZE. SEE APPLICATIONS INFORMATION
1578 F15
Figure 15. Positive-to-Negative Converter
Inverting regulators differ from buck regulators in the
basicswitchingnetwork. Currentisdeliveredtotheoutput
as square waves with a peak-to-peak amplitude much
greater than load current. This means that maximum load
current will be significantly less than the LT1578’s 1.5A
maximumswitchcurrent, evenwithlargeinductorvalues.
The buck converter in comparison, delivers current to the
output as a triangular wave superimposed on a DC level
equal to load current, and load current can approach 1.5A
A possible exception to the “clean response” rule is at very
light loads, as evidenced in Figure 14 with ILOAD = 50mA.
Switching regulators tend to have dramatic shifts in loop
response at very light loads, mostly because the inductor
currentbecomesdiscontinuous.Onecommonresultisvery
slow but stable characteristics. A second possibility is low
phase margin, as evidenced by ringing at the output with
transients. The good news is that the low phase margin at
23
LT1578/LT1578-2.5
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withlargeinductors.Outputripplevoltageforthepositive-
to-negative converter will be much higher than a buck
converter. Ripple current in the output capacitor will also
be much higher. The following equations can be used to
calculateoperatingconditionsforthepositive-to-negative
converter.
This duty cycle is close enough to 50% that IP can be
assumed to be 1.5A.
OUTPUT DIVIDER
If the adjustable part is used, the resistor connected to
VOUT (R2) should be set to approximately 5k. R1 is
calculated from:
Maximum load current:
V
V
(
)(
)
IN OUT
R2 V
− 1.21
(
)
OUT
I −
V
V −
0.35
(
)(
)
P
OUT IN
R1=
2 V
+ V f L
(
)( )( )
OUT
IN
1.21
I
=
MAX
V
+V − 0.35 V
+ V
F
(
)(
)
OUT
IN
OUT
INDUCTOR VALUE
IP = Maximum rated switch current
VIN = Minimum input voltage
VOUT = Output voltage
VF = Catch diode forward voltage
0.35 = Switch voltage drop at 1.5A
Unlike buck converters, positive-to-negative converters
cannot use large inductor values to reduce output ripple
voltage. At 200kHz, values larger than 75µH make almost
no change in output ripple. The graph in Figure 16 shows
peak-to-peak output ripple voltage for a 5V to –5V con-
verter versus inductor value. The criteria for choosing the
Example: with VIN(MIN) = 5.5V, VOUT = 5V, L = 30µH,
VF = 0.5V, IP = 1.5A: IMAX = 0.6A. Note that this equation
does not take into account that maximum rated switch
current (IP) on the LT1578 is reduced slightly for duty
cyclesabove50%. Ifdutycycleisexpectedtoexceed50%
(input voltage less than output voltage), use the actual IP
value from the Electrical Characteristics table.
150
5V TO –5V CONVERTER
OUTPUT CAPACITOR’S
ESR = 0.1Ω
120
DISCONTINUOUS
I
= 0.1A
90
60
30
0
LOAD
DISCONTINUOUS
= 0.25A
I
Operating duty cycle:
LOAD
V
OUT + VF
V − 0.3 + VOUT + VF
DC =
CONTINUOUS
I
IN
> 0.38A
LOAD
(This formula uses an average value for switch loss, so it
may be several percent in error.)
0
15
30
45
60
75
INDUCTOR SIZE (µH)
1578 F16
With the conditions above:
Figure 16. Ripple Voltage on Positive-to-Negative Converter
5 + 0.5
5.5 − 0.3 + 5 + 0.5
DC =
= 51%
24
LT1578/LT1578-2.5
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inductor is therefore typically based on ensuring that peak
switch current rating is not exceeded. This gives the
lowest value of inductance that can be used, but in some
cases (lower output load currents) it may give a value that
creates unnecessarily high output ripple voltage. A com-
promise value is often chosen that reduces output ripple.
As you can see from the graph, large inductors will not
give arbitrarily low ripple, but small inductors can give
high ripple.
For the example above, with maximum load current of
0.25A:
2
2
)
5.5 1.5
(
) (
I
=
= 0.38A
CONT
4 5.5 + 5 5.5 +5 +0.5
(
)(
)
This says that discontinuous mode can be used and the
minimum inductor needed is found from:
The difficulty in calculating the minimum inductor size
needed is that you must first know whether the switcher
will be in continuous or discontinuous mode at the critical
point where switch current is 1.5A. The first step is to use
the following formula to calculate the load current where
the switcher must use continuous mode. If your load
current is less than this, use the discontinuous mode
formula to calculate the minimum inductor value needed.
If the load current is higher, use the continuous mode
formula.
2 5 0.25
( )(
)
L
=
= 5.6µH
MIN
2
)
3
200• 10 1.5
(
Inpractice,theinductorshouldbeincreasedbyabout30%
over the calculated minimum to handle losses and varia-
tionsinvalue. Thissuggestsaminimuminductorof7.3µH
for this application, but looking at the ripple voltage chart
showsthatoutputripplevoltagecouldbereducedbyafac-
toroftwobyusinga30µHinductor.Thereisnoruleofthumb
heretomakeafinaldecision.Ifmodestrippleisneededand
the larger inductor does the trick, this is probably the best
solution. If ripple is noncritical use the smaller inductor. If
ripple is extremely critical, a second stage filter may have
to be added in any case, and the lower value of inductance
can be used. Keep in mind that the output capacitor is the
other critical factor in determining output ripple voltage.
Rippleshownonthegraph(Figure16)iswithacapacitor’s
ESR of 0.1Ω. This is reasonable for AVX type TPS “D” or
“E” size surface mount solid tantalum capacitors, but the
final capacitor chosen must be looked at carefully for ESR
characteristics.
Output current where continuous mode is needed:
2
V
2 I
( ) ( P)
IN
ICONT
=
4 V + V
V + V + V
IN OUT F
(
OUT)(
)
IN
Minimum inductor discontinuous mode:
2 V
I
(
OUT)( OUT
f I
)
LMIN
=
2
( )( P)
Minimum inductor continuous mode:
V
V
OUT
(
IN)(
)
LMIN
=
V
+ VF
(
)
OUT
2 f V + V
I −I
OUT
1+
( )(
)
IN
OUT
P
V
IN
25
LT1578/LT1578-2.5
U
W U U
APPLICATIONS INFORMATION
Diode Current
Ripple Current in the Input and Output Capacitors
Average diodecurrentisequaltoloadcurrent. Peak diode
current will be considerably higher.
Positive-to-negativeconvertershavehighripplecurrentin
both the input and output capacitors. For long capacitor
lifetime, the RMS value of this current must be less than
the high frequency ripple current rating of the capacitor.
The following formula will give an approximate value for
RMS ripple current. This formula assumes continuous
conduction mode and a large inductor value. Small induc-
tors will give somewhat higher ripple current, especially in
discontinuous mode. The exact formulas are very com-
plex and appear in Application Note 44, pages 30 and 31.
For our purposes here, a simple fudge factor (ff) is added.
The value for ff is about 1.2 for load currents above 0.38A
(in continuous conduction mode) and L ≥10µH. It in-
creases to about 2.0 for smaller inductors at lower load
currents (in discontinuous conduction mode).
Peak diode current:
Continuous Mode =
V + V
V
V
(
)
(
)(
)
IN
OUT
IN OUT
I
+
OUT
V
2 L f V + V
( )( )(
IN
)
IN
OUT
2 I
(
V
OUT
)(
)
OUT
Discontinuous Mode =
L f
( )( )
Keep in mind that during start-up and output overloads,
the average diode current may be much higher than with
normalloads.Careshouldbeusedifdiodesratedlessthan
1A are used, especially if continuous overload conditions
must be tolerated.
VOUT
Capacitor IRMS = ff I
( )( OUT
)
V
IN
ff = Fudge factor (1.2 to 2.0)
26
LT1578/LT1578-2.5
U
PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
7
5
8
6
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
3
4
2
0.010 – 0.020
(0.254 – 0.508)
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0°– 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.050
(1.270)
BSC
0.014 – 0.019
(0.355 – 0.483)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
SO8 1298
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
27
LT1578/LT1578-2.5
U
TYPICAL APPLICATION
Dual Output SEPIC Converter
coupling losses. C4 provides a low impedance path to
maintain an equal voltage swing in L1B, improving regu-
lation. In a flyback converter, during switch on time, all the
converter’s energy is stored in L1A only, since no current
flows in L1B. At switch off, energy is transferred by
magnetic coupling into L1B, powering the –5V rail. C4
pulls L1B positive during switch on time, causing current
to flow, and energy to build in L1B and C4. At switch off,
the energy stored in both L1B and C4 supply the –5V rail.
This reduces the current in L1A and changes L1B current
waveform from square to triangular. For details on this
circuit see Design Note 100.
The circuit in Figure 17 generates both positive and
negative 5V outputs with a single piece of magnetics. The
inductor L1 is a 33µH surface mount inductor from
Coiltronics. Itismanufacturedwithtwoidenticalwindings
thatcanbeconnectedinseriesorparallel.Thetopologyfor
the 5V output is a standard buck converter. The –5V
topologywouldbeasimpleflybackwindingcoupledtothe
buck converter if C4 were not present. C4 creates the
SEPIC (Single-Ended Primary Inductance Converter) to-
pology which improves regulation and reduces ripple
current in L1. Without C4, the voltage swing on L1B
compared to L1A would vary due to relative loading and
C2
0.33µF
D2
1N914
BOOST
INPUT
OUTPUT
5V
V
V
IN
SW
FB
6V TO 15V
L1A*
LT1578
33µH
R1
SHDN
GND
15.8k
V
C
+
C1**
100µF
R2
4.99k
10V TANT
+
C3
22µF
C
C
100pF
D1
35V TANT
1N5818
GND
+
+
C5**
C4**
100µF
* L1 IS A SINGLE CORE WITH TWO WINDINGS
COILTRONICS CTX33-2
100µF
L1B*
D3
1N5818
10V TANT
** AVX TSPD107M010
†
OUTPUT
IF LOAD CAN GO TO ZERO, AN OPTIONAL
†
–5V
PRELOAD OF 1k TO 5k MAY BE USED TO
IMPROVE LOAD REGULATION
1578 F17
Figure 17. Dual Output SEPIC Converter
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PART NUMBER DESCRIPTION
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LT1371
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Burst Mode is a trademark of Linear Technology Corporation.
1578f LT/TP 0100 4K • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
28
●
●
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
LINEAR TECHNOLOGY CORPORATION 1999
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