LT1372IN8 [Linear]
500kHz and 1MHz High Efficiency 1.5A Switching Regulators; 在500kHz至1MHz的高效率1.5A开关稳压器型号: | LT1372IN8 |
厂家: | Linear |
描述: | 500kHz and 1MHz High Efficiency 1.5A Switching Regulators |
文件: | 总12页 (文件大小:277K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LT1372/LT1377
500kHz and 1MHz
High Efficiency
1.5A Switching Regulators
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DESCRIPTION
FEATURES
The LT®1372/LT1377 are monolithic high frequency
switching regulators. They can be operated in all standard
switching configurations including boost, buck, flyback,
forward,invertingand“Cuk.”A1.5Ahighefficiencyswitch
is included on the die, along with all oscillator, control and
protection circuitry. All functions of the LT1372/LT1377
are integrated into 8-pin SO/PDIP packages.
■
Faster Switching with Increased Efficiency
■
Uses Small Inductors: 4.7µH
■
All Surface Mount Components
■
Only 0.5 Square Inch of Board Space
■
Low Minimum Supply Voltage: 2.7V
■
Quiescent Current: 4mA Typ
■
Current Limited Power Switch: 1.5A
■
Regulates Positive or Negative Outputs
The LT1372/LT1377 typically consumes only 4mA quies-
centcurrent and has higher efficiencythan previous parts.
High frequency switching allows for very small inductors
to be used. All surface mount components consume less
than 0.5 square inch of board space.
■
Shutdown Supply Current: 12µA Typ
■
Easy External Synchronization
■
8-Pin SO or PDIP Packages
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APPLICATIONS
New design techniques increase flexibility and maintain
ease of use. Switching is easily synchronized to an exter-
nal logic level source. A logic low on the shutdown pin
reduces supply current to 12µA. Unique error amplifier
circuitry can regulate positive or negative output voltage
while maintaining simple frequency compensation tech-
niques. Nonlinear error amplifier transconductance re-
duces output overshoot on start-up or overload recovery.
Oscillator frequency shifting protects external compo-
nents during overload conditions.
■
Boost Regulators
■
CCFL Backlight Driver
■
Laptop Computer Supplies
Multiple Output Flyback Supplies
Inverting Supplies
■
■
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATION
12V Output Efficiency
5V-to-12V Boost Converter
100
D1
5V
V
= 5V
IN
L1*
MBRS120T3
4.7µH
†
V
OUT
90
80
70
60
50
12V
5
V
IN
R1
4
8
2
53.6k
1%
ON
V
SW
S/S
LT1372/LT1377
FB
OFF
*COILCRAFT DO1608-472 (4.7µH) OR
COILCRAFT DT3316-103 (10µH) OR
SUMIDA CD43-4R7 (4.7µH) OR
+
+
C1**
22µF
C4**
22µF
GND
6, 7
V
C
SUMIDA CD73-100KC (10µH) OR
R2
6.19k
1%
**AVX TPSD226M025R0200
1
†MAX I
OUT
C2
0.047µF
R3
2k
L1
I
OUT
C3
4.7µH 0.25A
10µH 0.35A
0.0047µF
0.01
0.1
1
OUTPUT CURRENT (A)
LT1372 • TA01
LT1372 • TA02
1
LT1372/LT1377
W W W
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W
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ABSOLUTE AXI U RATI GS
Supply Voltage ....................................................... 30V
Switch Voltage
/O
PACKAGE RDER I FOR ATIO
ORDER PART NUMBER
TOP VIEW
LT1372CN8 LT1372HVIN8
LT1372HVCN8 LT1372IS8
LT1372CS8 LT1372HVIS8
LT1372HVCS8 LT1377CS8
LT1372/LT1377 .................................................. 35V
LT1372HV .......................................................... 42V
S/S Pin Voltage....................................................... 30V
Feedback Pin Voltage (Transient, 10ms) .............. ±10V
Feedback Pin Current........................................... 10mA
Negative Feedback Pin Voltage
(Transient, 10ms)............................................. ±10V
Operating Junction Temperature Range
Commercial ........................................ 0°C to 125°C*
Industrial ......................................... –40°C to 125°C
Short Circuit ......................................... 0°C to 150°C
Storage Temperature Range ................ –65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
V
1
2
3
4
V
SW
8
7
6
5
C
FB
NFB
S/S
GND
GND S
V
IN
LT1372IN8
LT1377IS8
S8 PART MARKING
1377
1372I 1372HI 1377I
N8 PACKAGE
8-LEAD PDIP
S8 PACKAGE
8-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 100°C/ W (N8)
JMAX = 125°C, θJA = 120°C/ W (S8)
1372
1372H
T
Consult factory for Military grade parts.
*Units shipped prior to Date Code 9552 are rated at 100°C maximum
operating temperature.
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
V
REF
Reference Voltage
Measured at Feedback Pin
V = 0.8V
C
1.230
1.225
1.245
1.245
1.260
1.265
V
V
●
I
Feedback Input Current
V
= V
REF
250
550
900
nA
nA
FB
FB
●
●
Reference Voltage Line Regulation
Negative Feedback Reference Voltage
2.7V ≤ V ≤ 25V, V = 0.8V
0.01
0.03
%/V
IN
C
V
Measured at Negative Feedback Pin
Feedback Pin Open, V = 0.8V
–2.540
–2.570
–2.490
–2.490
–2.440
–2.410
V
V
NFB
●
●
●
C
I
Negative Feedback Input Current
V
= V
–45
–30
0.01
–15
0.05
µA
NFB
NFB
NFR
Negative Feedback Reference Voltage
Line Regulation
2.7V ≤ V ≤ 25V, V = 0.8V
%/V
IN
C
g
m
Error Amplifier Transconductance
∆I = ±25µA
1100
700
1500
1900
2300
µmho
µmho
C
●
●
●
Error Amplifier Source Current
Error Amplifier Sink Current
Error Amplifier Clamp Voltage
V
V
= V – 150mV, V = 1.5V
120
200
350
µA
µA
FB
FB
REF
C
= V
+ 150mV, V = 1.5V
1400
2400
REF
C
High Clamp, V = 1V
Low Clamp, V = 1.5V
1.70
0.25
1.95
0.40
2.30
0.52
V
V
FB
FB
A
V
Error Amplifier Voltage Gain
500
1
V/V
V
V Pin Threshold
C
Duty Cycle = 0%
0.8
1.25
f
Switching Frequency
2.7V ≤ V ≤ 25V
LT1372
IN
450
430
400
0.90
0.86
0.80
500
500
550
580
580
1.10
1.16
1.16
kHz
kHz
kHz
MHz
MHz
MHz
0°C ≤ T ≤ 125°C
●
●
J
–40°C ≤ T < 0°C (I Grade)
J
LT1377
0°C ≤ T ≤ 125°C
1
1
J
–40°C ≤ T < 0°C (I Grade)
J
2
LT1372/LT1377
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
SYMBOL PARAMETER
Maximum Switch Duty Cycle
CONDITIONS
MIN
TYP
95
MAX
UNITS
%
●
90
Switch Current Limit Blanking Time
Output Switch Breakdown Voltage
130
47
260
ns
BV
LT1372/LT1377
LT1372HV
●
●
35
V
0°C ≤ T ≤ 125°C
42
40
47
V
V
J
–40°C ≤ T < 0°C (I Grade)
J
V
Output Switch “On” Resistance
Switch Current Limit
I
= 1A
SW
●
0.5
0.8
Ω
SAT
I
Duty Cycle = 50%
Duty Cycle = 80% (Note 1)
●
●
1.5
1.3
1.9
1.7
2.7
2.5
A
A
LIM
∆I
∆I
Supply Current Increase During Switch On-Time
15
25
mA/A
IN
SW
Control Voltage to Switch Current
Transconductance
2
A/V
Minimum Input Voltage
Supply Current
●
●
2.4
4
2.7
5.5
V
I
2.7V ≤ V ≤ 25V
mA
Q
IN
Shutdown Supply Current
2.7V ≤ V ≤ 25V, V ≤ 0.6V
IN S/S
0°C ≤ T ≤ 125°C
●
12
30
50
µA
µA
J
–40°C ≤ T < 0°C (I Grade)
J
Shutdown Threshold
2.7V ≤ V ≤ 25V
●
●
●
0.6
5
1.3
12
2
V
µs
µA
IN
Shutdown Delay
25
15
S/S Pin Input Current
Synchronization Frequency Range
0V ≤ V ≤ 5V
–10
S/S
LT1372
LT1377
●
●
600
1.2
800
1.6
kHz
MHz
The
●
denotes specifications which apply over the full operating
Note 1: For duty cycles (DC) between 50% and 90%, minimum
guaranteed switch current is given by I = 0.667 (2.75 – DC).
temperature range.
LIM
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TYPICAL PERFORMANCE CHARACTERISTICS
Switch Saturation Voltage
vs Switch Current
Switch Current Limit
vs Duty Cycle
Minimum Input Voltage
vs Temperature
3.0
2.5
2.0
1.5
1.0
0.5
0
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
3.0
2.8
2.6
2.4
2.2
2.0
1.8
150°C
100°C
25°C
25°C AND
125°C
–55°C
–55°C
75 100
125 150
0.2 0.4 0.6 0.8 1.0 1.2 1.4
SWITCH CURRENT (A)
1.6 1.8 2.0
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
–50 –25
0
25 50
0
TEMPERATURE (°C)
LT1372 • G02
LT1372 • G03
LT1372 • G01
3
LT1372/LT1377
TYPICAL PERFORMANCE CHARACTERISTICS
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Error Amplifier Output Current
vs Feedback Pin Voltage
Shutdown Delay and Threshold
vs Temperature
Minimum Synchronization
Voltage vs Temperature
20
18
16
14
12
10
8
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
400
300
3.0
2.5
2.0
1.5
1.0
0.5
0
f
f
= 700kHz (LT1372)
= 1.4MHz (LT1377)
SYNC
SYNC
25°C
–55°C
SHUTDOWN THRESHOLD
SHUTDOWN DELAY
200
125°C
100
LT1377
LT1372
0
6
–100
–200
–300
4
2
0
75 100
–0.3
–0.2
–0.1
V
0.1
–50
50
100 125
150
–50 –25
0
25 50
125 150
–25
0
25
75
REF
TEMPERATURE (°C)
FEEDBACK PIN VOLTAGE (V)
TEMPERATURE (°C)
LT1372 • G04
LT1372 • G05
LT1372 • G06
S/S Pin Input Current
vs Voltage
Switching Frequency
Error Amplifier Transconductance
vs Temperature
vs Feedback Pin Voltage
2000
1800
1600
1400
1200
1000
800
5
110
100
90
80
70
60
50
40
30
20
10
V
IN
= 5V
∆I (V )
∆V (FB)
C
g
=
m
4
3
2
1
0
–1
–2
–3
–4
–5
600
400
200
0
–50 –25
100
125
150
–1
0
1
2
3
4
5
6
7
8
9
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
FEEDBACK PIN VOLTAGE (V)
25 50
0
75
TEMPERATURE (°C)
S/S PIN VOLTAGE (V)
LT1372 • G07
LT1372 • G08
LT1372 • G09
VC Pin Threshold and High
Clamp Voltage vs Temperature
Feedback Input Current
vs Temperature
Negative Feedback Input Current
vs Temperature
0
–10
–20
–30
–40
–50
2.4
2.2
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
800
700
600
500
400
300
200
100
0
V
=V
REF
V
=V
NFR
FB
NFB
V
HIGH CLAMP
C
V
THRESHOLD
C
–50
50
125
100
–50
50
100 125
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
LT1372 • G11
–25
0
25
75
150
–25
0
25
75
150
TEMPERATURE (°C)
TEMPERATURE (°C)
LT1372 • G10
LT1372 • G12
4
LT1372/LT1377
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PIN FUNCTIONS
VC (Pin 1): The Compensation pin is used for frequency
compensation, current limiting and soft start. It is the
output of the error amplifier and the input of the current
comparator. Loop frequency compensation can be per-
formed with an RC network connected from the VC pin to
ground.
VIN (Pin5):Bypassinputsupplypinwith10µFormore.The
part goes into undervoltage lockout when VIN drops below
2.5V. Undervoltage lockout stops switching and pulls the
VC pin low.
GND S (Pin 6): The ground sense pin is a “clean” ground.
The internal reference, error amplifier and negative feed-
back amplifier are referred to the ground sense pin. Con-
nect it to ground. Keep the ground path connection to the
output resistor divider and the VC compensation network
free of large ground currents.
FB (Pin 2): The Feedback pin is used for positive output
voltage sensing and oscillator frequency shifting. It is the
inverting input to the error amplifier. The noninverting
input of this amplifier is internally tied to a 1.245V
reference. Load on the FB pin should not exceed 250µA
when the NFB pin is used. See Applications Information.
GND (Pin 7): The ground pin is the emitter connection of
thepowerswitchandhaslargecurrentsflowingthroughit.
It should be connected directly to a good quality ground
plane.
NFB (Pin 3): The Negative Feedback pin is used for
negative output voltage sensing. It is connected to the
inverting input of the negative feedback amplifier through
a 100k source resistor.
VSW (Pin 8): The switch pin is the collector of the power
switch and has large currents flowing through it. Keep the
traces to the switching components as short as possible to
minimize radiation and voltage spikes.
S/S (Pin 4): Shutdown and Synchronization Pin. The S/S
pin is logic level compatible. Shutdown is active low and
the shutdown threshold is typically 1.3V. For normal
operation, pull the S/S pin high, tie it to VIN or leave it
floating. To synchronize switching, drive the S/S pin be-
tween 600kHz and 800kHz (LT1372) or 1.2MHz to 1.6MHz
(LT1377).
W
BLOCK DIAGRAM
V
SW
IN
SHUTDOWN
DELAY AND RESET
LOW DROPOUT
2.3V REG
S/S
ANTI-SAT
LOGIC
DRIVER
SWITCH
SYNC
OSC
5:1 FREQUENCY
SHIFT
+
–
NFBA
100k
50k
NFB
FB
COMP
–
+
+
–
EA
IA
0.08Ω
A
V
≈ 6
V
C
1.245V
REF
GND LT1372 • BD
GND SENSE
5
LT1372/LT1377
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OPERATION
The LT1372/LT1377 are current mode switchers. This
means that switch duty cycle is directly controlled by
switch current rather than by output voltage. Referring to
the block diagram, the switch is turned “On” at the start of
eachoscillatorcycle.Itisturned“Off”whenswitchcurrent
reachesapredeterminedlevel. Controlofoutputvoltageis
obtained by using the output of a voltage sensing error
amplifier to set current trip level. This technique has
several advantages. First, it has immediate response to
input voltage variations, unlike voltage mode switchers
which have notoriously poor line transient response.
Second, it reduces the 90° phase shift at mid-frequencies
in the energy storage inductor. This greatly simplifies
closed-loop frequency compensation under widely vary-
ing input voltage or output load conditions. Finally, it
allows simple pulse-by-pulse current limiting to provide
maximum switch protection under output overload or
short conditions. A low dropout internal regulator pro-
vides a 2.3V supply for all internal circuitry. This low
dropout design allows input voltage to vary from 2.7V to
25V with virtually no change in device performance. A
500kHz(LT1372)or1MHz(LT1377)oscillatoristhebasic
clock for all internal timing. It turns “On” the output switch
via the logic and driver circuitry. Special adaptive anti-sat
circuitry detects onset of saturation in the power switch
and adjusts driver current instantaneously to limit switch
saturation. Thisminimizesdriverdissipationandprovides
very rapid turn-off of the switch.
put overshoot on start-up or overload recovery. When
the feedback voltage exceeds the reference by 40mV,
error amplifier transconductance increases ten times,
whichreducesoutputovershoot.Thefeedbackinputalso
invokes oscillator frequency shifting, which helps pro-
tect components during overload conditions. When the
feedback voltage drops below 0.6V, the oscillator fre-
quencyisreduced5:1.Lowerswitchingfrequencyallows
full control of switch current limit by reducing minimum
switch duty cycle.
UniqueerroramplifiercircuitryallowstheLT1372/LT1377
to directly regulate negative output voltages. The negative
feedback amplifier’s 100k source resistor is brought out
fornegativeoutputvoltagesensing. TheNFBpinregulates
at –2.49V while the amplifier output internally drives the
FB pin to 1.245V. This architecture, which uses the same
main error amplifier, prevents duplicating functions and
maintainseaseofuse. ConsultLinearTechnologyMarket-
ing for units that can regulate down to –1.25V.
The error signal developed at the amplifier output is
brought out externally. This pin (VC) has three different
functions. Itisusedforfrequencycompensation, current
limit adjustment and soft starting. During normal regula-
tor operation this pin sits at a voltage between 1V (low
outputcurrent)and 1.9V(highoutputcurrent). Theerror
amplifierisacurrentoutput(gm)type, sothisvoltagecan
be externally clamped for lowering current limit. Like-
wise, acapacitorcoupledexternalclampwillprovidesoft
start. Switch duty cycle goes to zero if the VC pin is pulled
below the control pin threshold, placing the LT1372/
LT1377 in an idle mode.
A 1.245V bandgap reference biases the positive input of
the error amplifier. The negative input of the amplifier is
broughtoutforpositiveoutputvoltagesensing.Theerror
amplifier has nonlinear transconductance to reduce out-
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APPLICATIO S I FOR ATIO
Positive Output Voltage Setting
V
OUT
The LT1372/LT1377 develops a 1.245V reference (VREF
)
R1
R2
R1
V
= V
1 +
OUT
REF
(
)
from the FB pin to ground. Output voltage is set by
connecting the FB pin to an output resistor divider
(Figure 1). The FB pin bias current represents a small
errorandcanusuallybeignoredforvaluesofR2upto7k.
The suggested value for R2 is 6.19k. The NFB pin is
normally left open for positive output applications.
FB
PIN
V
OUT
1.245
R1 = R2
– 1
(
)
R2
V
REF
LT1372 • F01
Figure 1. Positive Output Resistor Divider
6
LT1372/LT1377
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APPLICATIO S I FOR ATIO
Positive fixed voltage versions are available (consult
Linear Technology marketing).
Shutdown and Synchronization
The dual function S/S pin provides easy shutdown and
synchronization. It is logic level compatible and can be
pulledhigh, tiedtoVIN orleftfloatingfornormaloperation.
A logic low on the S/S pin activates shutdown, reducing
the part’s supply current to 12µA. Typical synchronization
rangeisfrom1.05to1.8timesthepart’snaturalswitching
frequency, but is only guaranteed between 600kHz and
800kHz (LT1372) or 1.2MHz and 1.6MHz (LT1377). A
12µs resetable shutdown delay network guarantees the
part will not go into shutdown while receiving a synchro-
nization signal.
Negative Output Voltage Setting
The LT1372/LT1377 develops a –2.49V reference (VNFR
)
from the NFB pin to ground. Output voltage is set by
connecting the NFB pin to an output resistor divider
(Figure 2). The –30µA NFB pin bias current (INFB) can
cause output voltage errors and should not be ignored.
ThishasbeenaccountedforintheformulainFigure2. The
suggested value for R2 is 2.49k. The FB pin is normally left
open for negative output application. See Dual Polarity
Output Voltage Sensing for limitatins on FB pin loading
when using the NFB pin.
Cautionshouldbeusedwhensynchronizingabove700kHz
(LT1372) or 1.4MHz (LT1377) because at higher sync
frequenciestheamplitudeoftheinternalslopecompensa-
tion used to prevent subharmonic switching is reduced.
This type of subharmonic switching only occurs when the
duty cycle of the switch is above 50%. Higher inductor
values will tend to eliminate problems.
–V
OUT
R1
R2
–V
= V
NFB
1 +
+ I
(R1)
NFB
OUT
(
)
R1
I
NFB
NFB
PIN
V
– 2.49
OUT
R1 =
R2
2.49
R2
6
+
30 × 10–
V
NFR
(
)
(
)
LT1372 • F02
Thermal Considerations
Figure 2. Negative Output Resistor Divider
Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause exces-
sive die temperatures. The packages are rated at 120°C/W
for SO (S8) and 130°C/W for PDIP (N8).
Dual Polarity Output Voltage Sensing
Certain applications benefit from sensing both positive
and negative output voltages. One example is the “Dual
Output Flyback Converter with Overvoltage Protection”
circuit shown in the Typical Applications section. Each
output voltage resistor divider is individually set as de-
scribed above. When both the FB and NFB pins are used,
the LT1372/LT1377 acts to prevent either output from
going beyond its set output voltage. For example in this
application,ifthepositiveoutputweremoreheavilyloaded
than the negative, the negative output would be greater
and would regulate at the desired set-point voltage. The
positive output would sag slightly below its set-point
voltage. This technique prevents either output from going
unregulated high at no load. Please note that the load on
the FB pin should not exceed 250µA when the NFB pin is
used. This situation occurs when the resistor dividers are
used at both FB and NFB. True load on FB is not the full
divider current unless the positive output is shorted to
ground. See Dual Output Flyback Converter application.
Average supply current (including driver current) is:
IIN = 4mA + DC (ISW/60 + ISW × 0.004)
ISW = switch current
DC = switch duty cycle
Switch power dissipation is given by:
PSW = (ISW)2 × RSW × DC
RSW = output switch “On” resistance
Total power dissipation of the die is the sum of supply
current times supply voltage plus switch power:
PD(TOTAL) = (IIN × VIN) + PSW
7
LT1372/LT1377
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APPLICATIO S I FOR ATIO
Choosing the Inductor
radiation, or whether it needs a closed core like a toroid
to prevent EMI problems. One would not want an open
core next to a magnetic storage media for instance!
This is a tough decision because the rods or barrels are
temptingly cheap and small, and there are no helpful
guidelines to calculate when the magnetic field radia-
tion will be a problem.
For most applications the inductor will fall in the range of
2.2µH to 22µH. Lower values are chosen to reduce physi-
cal size of the inductor. Higher values allow more output
current because they reduce peak current seen by the
power switch, which has a 1.5A limit. Higher values also
reduce input ripple voltage and reduce core loss.
4. Start shopping for an inductor which meets the re-
quirements of core shape, peak current (to avoid
saturation), averagecurrent(tolimitheating)andfault
current.Iftheinductorgetstoohot,wireinsulationwill
melt and cause turn-to-turn shorts. Keep in mind that
allgoodthingslikehighefficiency,lowprofileandhigh
temperature operation will increase cost, sometimes
dramatically.
When choosing an inductor you might have to consider
maximum load current, core and copper losses, allowable
component height, output voltage ripple, EMI, fault
current in the inductor, saturation, and of course, cost.
Thefollowingprocedureissuggestedasawayofhandling
thesesomewhatcomplicatedandconflictingrequirements.
1. Assume that the average inductor current for a boost
converter is equal to load current times VOUT/VIN and
decide whether or not the inductor must withstand
continuous overload conditions. If average inductor
current at maximum load current is 0.5A, for instance,
a 0.5A inductor may not survive a continuous 1.5A
overload condition. Also be aware that boost convert-
ers are not short circuit protected, and that under
outputshortconditions, inductorcurrentislimitedonly
by the available current of the input supply.
5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the Linear Technology application
department if you feel uncertain about the final choice.
They have experience with a wide range of inductor
types and can tell you about the latest developments in
low profile, surface mounting, etc.
Output Capacitor
2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, espe-
cially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving be-
cause they saturate softly, whereas ferrite cores satu-
rate abruptly. Other core materials fall in between
somewhere. The following formula assumes continu-
ous mode operation but it errors only slightly on the
high side for discontinuous mode, so it can be used for
all conditions.
The output capacitor is normally chosen by its effective
seriesresistance,(ESR),becausethisiswhatdetermines
output ripple voltage. At 500kHz, any polarized capacitor
is essentially resistive. To get low ESR takes volume, so
physically smaller capacitors have high ESR. The ESR
range for typical LT1372 and LT1377 applications is
0.05Ω to 0.5Ω. A typical output capacitor is an AVX type
TPS, 22µF at 25V, with a guaranteed ESR less than 0.2Ω.
This is a “D” size surface mount solid tantalum capacitor.
TPS capacitors are specially constructed and tested for
low ESR, so they give the lowest ESR for a given volume.
To further reduce ESR, multiple output capacitors can be
used in parallel. The value in microfarads is not particu-
larly critical, and values from 22µF to greater than 500µF
work well, but you cannot cheat mother nature on ESR.
Ifyoufindatiny22µFsolidtantalumcapacitor,itwillhave
high ESR, and output ripple voltage will be terrible. Table
1 shows some typical solid tantalum surface mount
capacitors.
V
V
V (V
2(f)(L)(V
– V )
OUT
IN OUT IN
I
= I
×
OUT
+
PEAK
)
IN
OUT
V = Minimum Input Voltage
IN
f = 500kHz Switching Frequency (LT1372) or
1MHz Switching Frequency (LT1377)
3. Decide if the design can tolerate an “open” core geom-
etry like a rod or barrel, which have high magnetic field
8
LT1372/LT1377
U U
W
U
APPLICATIO S I FOR ATIO
Table 1. Surface Mount Solid Tantalum Capacitor
ESR and Ripple Current
0.3(V )(V
– V )
IN
)
IN OUT
I
=
RIPPLE
E CASE SIZE
ESR (MAX Ω)
RIPPLE CURRENT (A)
(f)(L)(V
OUT
AVX TPS, Sprague 593D
AVX TAJ
0.1 to 0.3
0.7 to 0.9
0.7 to 1.1
0.4
f = 500kHz Switching frequency (LT1372) or,
1MHz Switching frequency (LT1377)
D CASE SIZE
AVX TPS, Sprague 593D
AVX TAJ
0.1 to 0.3
0.9 to 2.0
0.7 to 1.1
0.36 to 0.24
Theinputcapacitorcanseeaveryhighsurgecurrentwhen
a battery or high capacitance source is connected “live”
andsolidtantalumcapacitorscanfailunderthiscondition.
Several manufacturers have developed a line of solid
tantalum capacitors specially tested for surge capability
(AVX TPS series, for instance), but even these units may
fail if the input voltage approaches the maximum voltage
rating of the capacitor. AVX recommends derating capaci-
torvoltageby2:1forhighsurgeapplications. Ceramicand
aluminum electrolytic capacitors may also be used and
have a high tolerance to turn-on surges.
C CASE SIZE
AVX TPS
AVX TAJ
0.2 (Typ)
1.8 to 3.0
0.5 (Typ)
0.22 to 0.17
B CASE SIZE
AVX TAJ
2.5 to 10
0.16 to 0.08
Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true and type TPS capacitors are
speciallytestedforsurgecapability,butsurgeruggedness
is not a critical issue with the output capacitor. Solid
tantalum capacitors fail during very high turn-on surges,
which do not occur at the output of regulators. High
discharge surges, such as when the regulator output is
dead shorted, do not harm the capacitors.
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becomingavailableinsmallercasesizes.Thesearetempt-
ing for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generatesaloop“zero”at5kHzto50kHzthatisinstrumen-
tal in giving acceptable loop phase margin. Ceramic ca-
pacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
They are appropriate for input bypassing because of their
highripplecurrentratingsandtoleranceofturn-onsurges.
Linear Technology plans to issue a Design Note on the use
of ceramic capacitors in the near future.
Single inductor boost regulators have large RMS ripple
current in the output capacitor, which must be rated to
handle the current. The formula to calculate this is:
Output Capacitor Ripple Current (RMS)
DC
1 – DC
I
(RMS) = I
= I
RIPPLE
OUT
V
OUT
– V
IN
IN
OUT
V
Input Capacitors
Output Diode
The input capacitor of a boost converter is less critical due
tothefactthattheinputcurrentwaveformistriangularand
does not contain large squarewave currents as is found in
the output capacitor. Capacitors in the range of 10µF to
100µFwithanESRof0.3Ω orlessworkwelluptofull1.5A
switch current. Higher ESR capacitors may be acceptable
at low switch currents. Input capacitor ripple current for
boost converter is :
The suggested output diode (D1) is a 1N5818 Schottky or
its Motorola equivalent, MBR130. It is rated at 1A average
forward current and 30V reverse voltage. Typical forward
voltage is 0.42V at 1A. The diode conducts current only
during switch off time. Peak reverse voltage for boost
converters is equal to regulator output voltage. Average
forward current in normal operation is equal to output
current.
9
LT1372/LT1377
U U
W
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APPLICATIO S I FOR ATIO
Frequency Compensation
(magnetic) radiation is minimized by keeping output di-
ode, switch pin, and output bypass capacitor leads as
short as possible. E field radiation is kept low by minimiz-
ingthelengthandareaofalltracesconnectedtotheswitch
pin. A ground plane should always be used under the
switcher circuitry to prevent interplane coupling.
Loopfrequencycompensationisperformedontheoutput
of the error amplifier (VC pin) with a series RC network.
The main pole is formed by the series capacitor and the
output impedance (≈500kΩ) of the error amplifier. The
pole falls in the range of 2Hz to 20Hz. The series resistor
creates a “zero” at 1kHz to 5kHz, which improves loop
stability and transient response. A second capacitor,
typically one-tenth the size of the main compensation
capacitor, is sometimes used to reduce the switching
frequency ripple on the VC pin. VC pin ripple is caused by
output voltage ripple attenuated by the output divider and
multiplied by the error amplifier. Without the second
capacitor, VC pin ripple is:
Thehighspeedswitchingcurrentpathisshownschemati-
cally in Figure 3. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, output diode, and output capacitor is
the only one containing nanosecond rise and fall times.
Keep this path as short as possible.
SWITCH
NODE
L1
V
OUT
1.245(V
)(g )(R )
m C
RIPPLE
(V
HIGH
FREQUENCY
CIRCULATING
PATH
V Pin Ripple =
C
V
LOAD
IN
)
OUT
V
m
= Output ripple (V
)
P–P
RIPPLE
g = Error amplifier transconductance
≈(1500µmho)
LT1372 • F03
Figure 3
R = Series resistor on V pin
V
C
OUT
C
= DC output voltage
More Help
To prevent irregular switching, VC pin ripple should be
kept below 50mVP–P. Worst-case VC pin ripple occurs at
maximum output load current and will also be increased
if poor quality (high ESR) output capacitors are used. The
addition of a 0.0047µF capacitor on the VC pin reduces
switching frequency ripple to only a few millivolts. A low
value for RC will also reduce VC pin ripple, but loop phase
margin may be inadequate.
For more detailed information on switching regulator
circuits, please see Application Note 19. Linear Technol-
ogyalsooffersacomputersoftwareprogram,SwitcherCAD,
to assist in designing switching converters. SwitcherCAD
will be updated in late 1995 for the LT1372 and LT1377. In
addition, our applications department is always ready to
lend a helping hand.
Switch Node Considerations
For maximum efficiency, switch rise and fall time are
made as short as possible. To prevent radiation and high
frequency resonance problems, proper layout of the com-
ponents connected to the switch node is essential. B field
10
LT1372/LT1377
U
TYPICAL APPLICATIONS N
Positive-to-Negative Converter with Direct Feedback
Dual Output Flyback Converter with Overvoltage Protection
V
IN
R2
1.21k
1%
R1
13k
1%
2.7V TO 16V
T1*
+
2
4
C1
D2
•
+
22µF
C4
P6KE-15A
D3
V
IN
5
47µF
MBRS140T3
T1*
2.7V TO 13V
•
V
IN
1N4148
V
OUT
†
1
3
4
8
3
–V
ON
OUT
V
15V
+
2, 3
5
S/S
SW
OFF
C1
22µF
–5V
+
R2
2.49k
1%
D1
MBRS130LT3
•
P6KE-20A
C4
LT1372/LT1377
NFB
GND
6, 7
47µF
2
5
4
8
1N4148
•
•
FB
S/S
V
IN
V
R3
2.49k
1%
4
8
3
6, 7
ON
+
V
SW
C
OFF
C5
47µF
1
LT1372/LT1377
NFB
GND
6, 7
1
–V
OUT
C2
0.047µF
R1
2k
*COILTRONICS CTX10-2 (407) 241-7876
†
–15V
R4
12.1k
1%
MAX I
OUT
MBRS140T3
C3
0.0047µF
V
C
I
V
IN
OUT
1
LT1372 • TA03
0.3A 3V
0.5A 5V
0.75A 9V
R5
2.49k
1%
C2
0.047µF
R3
2k
C3
0.0047µF
*DALE LPE-4841-100MB (605) 665-9301
LT1372 • TA04
90% Efficient CCFL Supply
Low Ripple 5V to –3V “Cuk”† Converter
5mA MAX
LAMP
C2
27pF
D1
1N4148
V
10
OUT
L1*
V
IN
–3V
5V
T1
2
1
3
4
250mA
V
IN
5
4
3
2
1
4.5V
•
•
+
TO 30V
10µF
C1
0.1µF
R1
1k
1%
C2
47µF
16V
5
4
7
6
8
V
V
SW
IN
Q1
Q2
+
+
C1
22µF
10V
C6
0.1µF
S/S
LT1372/LT1377
GND NFB
330Ω
3
1
D2
L1
33µH
2.7V TO
5.5V
GND S
V
C
C3
47µF
16V
1N4148
+
D1**
1N5818
8
2.2µF
+
5
C5
0.0047µF
R4
2k
V
R2
4.99k
1%
IN
562Ω*
10k
4
ON
S/S
V
SW
OFF
C4
0.047µF
20k
DIMMING
LT1372/LT1377
2
*SUMIDA CLS62-100L
LT1372 • TA05
V
FB
**MOTOROLA MBR0520LT3
†
GND
6, 7
V
C
PATENTS MAY APPLY
0.1µF
22k
1N4148
1
+
2µF
OPTIONAL REMOTE
DIMMING
LT1372 • TA06
C1 = WIMA MKP-20
L1 = COILCRAFT DT3316-333
Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001
T1 = COILTRONICS CTX 110609
* = 1% FILM RESISTOR
CCFL BACKLIGHT APPLICATION CIRCUITS
CONTAINED IN THIS DATA SHEET ARE
COVERED BY U.S. PATENT NUMBER 5408162
AND OTHER PATENTS PENDING
DO NOT SUBSTITUTE COMPONENTS
COILTRONICS (407) 241-7876
COILCRAFT (708) 639-6400
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of circuits as described herein will not infringe on existing patent rights.
11
LT1372/LT1377
U
TYPICAL APPLICATIONS N
2 Li-Ion Cell to 5V SEPIC Converter
V
IN
4V TO 9V
L1A*
10µH
5
MBRS130LT3
V
IN
†
•
4
8
2
V
OUT
5V
ON
V
C1 = AVX TPSD 336M020R0200
C2 = TOKIN 1E105ZY5U-C103-F
C3 = AVX TPSD107M010R0100
S/S
SW
OFF
C1
33µF
20V
R2
C2
1µF
LT1372/LT1377
18.7k
1%
+
FB
+
C3
*SINGLE INDUCTOR WITH TWO WINDINGS
•
100µF
GND
6, 7
V
C
COILTRONICS CTX10-1
†
10V
L1B*
10µH
MAX I
OUT
1
I
V
IN
R3
6.19k
1%
OUT
R1
2k
0.45A 4V
0.55A 5V
0.65A 7V
0.72A 9V
C5
0.0047µF
C4
0.047µF
LT1372 • TA07
U
PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.400*
(10.160)
MAX
0.130 ± 0.005
0.300 – 0.325
0.045 – 0.065
(3.302 ± 0.127)
(1.143 – 1.651)
(7.620 – 8.255)
8
7
6
5
0.065
(1.651)
TYP
0.255 ± 0.015*
(6.477 ± 0.381)
0.009 – 0.015
(0.229 – 0.381)
+0.025
0.125
(3.175)
MIN
0.005
(0.127)
MIN
0.015
(0.380)
MIN
0.325
1
2
4
3
–0.015
+0.635
8.255
N8 0695
0.100 ± 0.010
(2.540 ± 0.254)
0.018 ± 0.003
(0.457 ± 0.076)
(
)
–0.381
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
0.010 – 0.020
(0.254 – 0.508)
7
5
8
6
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0°– 8° TYP
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
0.016 – 0.050
0.406 – 1.270
0.050
(1.270)
BSC
0.014 – 0.019
(0.355 – 0.483)
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
SO8 0695
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
1
2
3
4
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
Good for Up to V = 40V
LT1172
100kHz 1.25A Boost Switching Regulator
12V 1.2A Monolithic Buck Converter
Micropower 2A Boost Converter
IN
LTC®1265
LT1302
Converts 5V to 3.3V at 1A with 90% Efficiency
Converts 2V to 5V at 600mA in SO8 Packages
Steps Down from Up to 25V Using 4.7µH Inductors
90% Efficient Boost Converter with Constant Frequency
LT1376
500kHz 1.5A Buck Switching Regulator
Low Supply Current 250kHz 1.5A Boost Switching Regulator
LT1373
LT/GP 0996 5K REV A • PRINTED IN THE USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
12
●
●
(408) 432-1900 FAX: (408) 434-0507 TELEX: 499-3977
LINEAR TECHNOLOGY CORPORATION 1995
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