LT1372IS8 [Linear]

500kHz and 1MHz High Efficiency 1.5A Switching Regulators; 在500kHz至1MHz的高效率1.5A开关稳压器


型号：	LT1372IS8
厂家：	Linear
描述：	500kHz and 1MHz High Efficiency 1.5A Switching Regulators 在500kHz至1MHz的高效率1.5A开关稳压器稳压器开关式稳压器或控制器电源电路开关式控制器光电二极管
文件：	总12页 (文件大小：277K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LT1372/LT1377

500kHz and 1MHz

High Efficiency

1.5A Switching Regulators

DESCRIPTION

FEATURES

The LT^®1372/LT1377 are monolithic high frequency

switching regulators. They can be operated in all standard

switching configurations including boost, buck, flyback,

forward,invertingand“Cuk.”A1.5Ahighefficiencyswitch

is included on the die, along with all oscillator, control and

protection circuitry. All functions of the LT1372/LT1377

are integrated into 8-pin SO/PDIP packages.

■

Faster Switching with Increased Efficiency

■

Uses Small Inductors: 4.7µH

■

All Surface Mount Components

■

Only 0.5 Square Inch of Board Space

■

Low Minimum Supply Voltage: 2.7V

■

Quiescent Current: 4mA Typ

■

Current Limited Power Switch: 1.5A

■

Regulates Positive or Negative Outputs

The LT1372/LT1377 typically consumes only 4mA quies-

centcurrent and has higher efficiencythan previous parts.

High frequency switching allows for very small inductors

to be used. All surface mount components consume less

than 0.5 square inch of board space.

■

Shutdown Supply Current: 12µA Typ

■

Easy External Synchronization

■

8-Pin SO or PDIP Packages

APPLICATIONS

New design techniques increase flexibility and maintain

ease of use. Switching is easily synchronized to an exter-

nal logic level source. A logic low on the shutdown pin

reduces supply current to 12µA. Unique error amplifier

circuitry can regulate positive or negative output voltage

while maintaining simple frequency compensation tech-

niques. Nonlinear error amplifier transconductance re-

duces output overshoot on start-up or overload recovery.

Oscillator frequency shifting protects external compo-

nents during overload conditions.

■

Boost Regulators

■

CCFL Backlight Driver

■

Laptop Computer Supplies

Multiple Output Flyback Supplies

Inverting Supplies

■

, LTC and LT are registered trademarks of Linear Technology Corporation.

TYPICAL APPLICATION

12V Output Efficiency

5V-to-12V Boost Converter

100

= 5V

L1*

MBRS120T3

4.7µH

†

OUT

12V

53.6k

S/S

LT1372/LT1377

OFF

*COILCRAFT DO1608-472 (4.7µH) OR

COILCRAFT DT3316-103 (10µH) OR

SUMIDA CD43-4R7 (4.7µH) OR

C1**

22µF

C4**

22µF

GND

6, 7

SUMIDA CD73-100KC (10µH) OR

6.19k

**AVX TPSD226M025R0200

^†MAX I

OUT

0.047µF

OUT

4.7µH 0.25A

10µH 0.35A

0.0047µF

0.01

0.1

OUTPUT CURRENT (A)

LT1372 • TA01

LT1372 • TA02

LT1372/LT1377

W W W

ABSOLUTE AXI U RATI GS

Supply Voltage ....................................................... 30V

Switch Voltage

PACKAGE RDER I FOR ATIO

ORDER PART NUMBER

TOP VIEW

LT1372CN8 LT1372HVIN8

LT1372HVCN8 LT1372IS8

LT1372CS8 LT1372HVIS8

LT1372HVCS8 LT1377CS8

LT1372/LT1377 .................................................. 35V

LT1372HV .......................................................... 42V

S/S Pin Voltage....................................................... 30V

Feedback Pin Voltage (Transient, 10ms) .............. ±10V

Feedback Pin Current........................................... 10mA

Negative Feedback Pin Voltage

(Transient, 10ms)............................................. ±10V

Operating Junction Temperature Range

Commercial ........................................ 0°C to 125°C*

Industrial ......................................... –40°C to 125°C

Short Circuit ......................................... 0°C to 150°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

NFB

S/S

GND

GND S

LT1372IN8

LT1377IS8

S8 PART MARKING

1377

1372I 1372HI 1377I

N8 PACKAGE

8-LEAD PDIP

S8 PACKAGE

8-LEAD PLASTIC SO

T_JMAX= 125°C, θ_JA= 100°C/ W (N8)

_JMAX= 125°C, θ_JA= 120°C/ W (S8)

1372

1372H

Consult factory for Military grade parts.

*Units shipped prior to Date Code 9552 are rated at 100°C maximum

operating temperature.

ELECTRICAL CHARACTERISTICS

V_IN= 5V, V_C= 0.6V, V_FB= V_REF, V_SW, S/S and NFB pins open, unless otherwise noted.

SYMBOL PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

REF

Reference Voltage

Measured at Feedback Pin

V = 0.8V

1.230

1.225

1.245

1.260

1.265

●

Feedback Input Current

= V

REF

250

550

900

●

Reference Voltage Line Regulation

Negative Feedback Reference Voltage

2.7V ≤ V ≤ 25V, V = 0.8V

0.01

0.03

%/V

Measured at Negative Feedback Pin

Feedback Pin Open, V = 0.8V

–2.540

–2.570

–2.490

–2.440

–2.410

NFB

●

Negative Feedback Input Current

= V

–45

–30

0.01

–15

0.05

µA

NFB

NFR

Negative Feedback Reference Voltage

Line Regulation

2.7V ≤ V ≤ 25V, V = 0.8V

%/V

Error Amplifier Transconductance

∆I = ±25µA

1100

700

1500

1900

2300

µmho

●

Error Amplifier Source Current

Error Amplifier Sink Current

Error Amplifier Clamp Voltage

= V – 150mV, V = 1.5V

120

200

350

µA

REF

= V

+ 150mV, V = 1.5V

1400

2400

REF

High Clamp, V = 1V

Low Clamp, V = 1.5V

1.70

0.25

1.95

0.40

2.30

0.52

Error Amplifier Voltage Gain

500

V/V

V Pin Threshold

Duty Cycle = 0%

0.8

1.25

Switching Frequency

2.7V ≤ V ≤ 25V

LT1372

450

430

400

0.90

0.86

0.80

500

550

580

1.10

1.16

kHz

MHz

0°C ≤ T ≤ 125°C

●

–40°C ≤ T < 0°C (I Grade)

LT1377

0°C ≤ T ≤ 125°C

–40°C ≤ T < 0°C (I Grade)

LT1372/LT1377

ELECTRICAL CHARACTERISTICS

V_IN= 5V, V_C= 0.6V, V_FB= V_REF, V_SW, S/S and NFB pins open, unless otherwise noted.

SYMBOL PARAMETER

Maximum Switch Duty Cycle

CONDITIONS

MIN

TYP

MAX

UNITS

●

Switch Current Limit Blanking Time

Output Switch Breakdown Voltage

130

260

LT1372/LT1377

LT1372HV

●

0°C ≤ T ≤ 125°C

–40°C ≤ T < 0°C (I Grade)

Output Switch “On” Resistance

Switch Current Limit

= 1A

●

0.5

0.8

Ω

SAT

Duty Cycle = 50%

Duty Cycle = 80% (Note 1)

●

1.5

1.3

1.9

1.7

2.7

2.5

LIM

∆I

Supply Current Increase During Switch On-Time

mA/A

Control Voltage to Switch Current

Transconductance

A/V

Minimum Input Voltage

Supply Current

●

2.4

2.7

5.5

2.7V ≤ V ≤ 25V

Shutdown Supply Current

2.7V ≤ V ≤ 25V, V ≤ 0.6V

IN S/S

0°C ≤ T ≤ 125°C

●

µA

–40°C ≤ T < 0°C (I Grade)

Shutdown Threshold

2.7V ≤ V ≤ 25V

●

0.6

1.3

µs

µA

Shutdown Delay

S/S Pin Input Current

Synchronization Frequency Range

0V ≤ V ≤ 5V

–10

S/S

LT1372

LT1377

●

600

1.2

800

1.6

kHz

MHz

The

●

denotes specifications which apply over the full operating

Note 1: For duty cycles (DC) between 50% and 90%, minimum

guaranteed switch current is given by I = 0.667 (2.75 – DC).

temperature range.

LIM

TYPICAL PERFORMANCE CHARACTERISTICS

Switch Saturation Voltage

vs Switch Current

Switch Current Limit

vs Duty Cycle

Minimum Input Voltage

vs Temperature

3.0

2.5

2.0

1.5

1.0

0.5

1.0

0.9

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1

3.0

2.8

2.6

2.4

2.2

2.0

1.8

150°C

100°C

25°C

25°C AND

125°C

–55°C

75 100

125 150

0.2 0.4 0.6 0.8 1.0 1.2 1.4

SWITCH CURRENT (A)

1.6 1.8 2.0

10 20 30 40 50 60 70 80 90 100

DUTY CYCLE (%)

–50 –25

25 50

TEMPERATURE (°C)

LT1372 • G02

LT1372 • G03

LT1372 • G01

LT1372/LT1377

TYPICAL PERFORMANCE CHARACTERISTICS

Error Amplifier Output Current

vs Feedback Pin Voltage

Shutdown Delay and Threshold

vs Temperature

Minimum Synchronization

Voltage vs Temperature

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

400

300

3.0

2.5

2.0

1.5

1.0

0.5

= 700kHz (LT1372)

= 1.4MHz (LT1377)

SYNC

25°C

–55°C

SHUTDOWN THRESHOLD

SHUTDOWN DELAY

200

125°C

100

LT1377

LT1372

–100

–200

–300

75 100

–0.3

–0.2

–0.1

0.1

–50

100 125

150

–50 –25

25 50

125 150

–25

REF

TEMPERATURE (°C)

FEEDBACK PIN VOLTAGE (V)

TEMPERATURE (°C)

LT1372 • G04

LT1372 • G05

LT1372 • G06

S/S Pin Input Current

vs Voltage

Switching Frequency

Error Amplifier Transconductance

vs Temperature

vs Feedback Pin Voltage

2000

1800

1600

1400

1200

1000

800

110

100

= 5V

∆I (V )

∆V (FB)

–1

–2

–3

–4

–5

600

400

200

–50 –25

100

125

150

–1

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0

FEEDBACK PIN VOLTAGE (V)

25 50

TEMPERATURE (°C)

S/S PIN VOLTAGE (V)

LT1372 • G07

LT1372 • G08

LT1372 • G09

V_CPin Threshold and High

Clamp Voltage vs Temperature

Feedback Input Current

vs Temperature

Negative Feedback Input Current

vs Temperature

–10

–20

–30

–40

–50

2.4

2.2

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

800

700

600

500

400

300

200

100

REF

NFR

NFB

HIGH CLAMP

THRESHOLD

–50

125

100

–50

100 125

–50 –25

25 50 75 100 125 150

TEMPERATURE (°C)

LT1372 • G11

–25

150

–25

150

TEMPERATURE (°C)

LT1372 • G10

LT1372 • G12

LT1372/LT1377

PIN FUNCTIONS

V_C(Pin 1): The Compensation pin is used for frequency

compensation, current limiting and soft start. It is the

output of the error amplifier and the input of the current

comparator. Loop frequency compensation can be per-

formed with an RC network connected from the V_Cpin to

ground.

V_IN(Pin5):Bypassinputsupplypinwith10µFormore.The

part goes into undervoltage lockout when V_INdrops below

2.5V. Undervoltage lockout stops switching and pulls the

V_Cpin low.

GND S (Pin 6): The ground sense pin is a “clean” ground.

The internal reference, error amplifier and negative feed-

back amplifier are referred to the ground sense pin. Con-

nect it to ground. Keep the ground path connection to the

output resistor divider and the V_Ccompensation network

free of large ground currents.

FB (Pin 2): The Feedback pin is used for positive output

voltage sensing and oscillator frequency shifting. It is the

inverting input to the error amplifier. The noninverting

input of this amplifier is internally tied to a 1.245V

reference. Load on the FB pin should not exceed 250µA

when the NFB pin is used. See Applications Information.

GND (Pin 7): The ground pin is the emitter connection of

thepowerswitchandhaslargecurrentsflowingthroughit.

It should be connected directly to a good quality ground

plane.

NFB (Pin 3): The Negative Feedback pin is used for

negative output voltage sensing. It is connected to the

inverting input of the negative feedback amplifier through

a 100k source resistor.

V_SW(Pin 8): The switch pin is the collector of the power

switch and has large currents flowing through it. Keep the

traces to the switching components as short as possible to

minimize radiation and voltage spikes.

S/S (Pin 4): Shutdown and Synchronization Pin. The S/S

pin is logic level compatible. Shutdown is active low and

the shutdown threshold is typically 1.3V. For normal

operation, pull the S/S pin high, tie it to V_INor leave it

floating. To synchronize switching, drive the S/S pin be-

tween 600kHz and 800kHz (LT1372) or 1.2MHz to 1.6MHz

(LT1377).

BLOCK DIAGRAM

SHUTDOWN

DELAY AND RESET

LOW DROPOUT

2.3V REG

S/S

ANTI-SAT

LOGIC

DRIVER

SWITCH

SYNC

OSC

5:1 FREQUENCY

SHIFT

–

NFBA

100k

50k

NFB

COMP

–

0.08Ω

≈ 6

1.245V

REF

GND LT1372 • BD

GND SENSE

LT1372/LT1377

OPERATION

The LT1372/LT1377 are current mode switchers. This

means that switch duty cycle is directly controlled by

switch current rather than by output voltage. Referring to

the block diagram, the switch is turned “On” at the start of

eachoscillatorcycle.Itisturned“Off”whenswitchcurrent

reachesapredeterminedlevel. Controlofoutputvoltageis

obtained by using the output of a voltage sensing error

amplifier to set current trip level. This technique has

several advantages. First, it has immediate response to

input voltage variations, unlike voltage mode switchers

which have notoriously poor line transient response.

Second, it reduces the 90° phase shift at mid-frequencies

in the energy storage inductor. This greatly simplifies

closed-loop frequency compensation under widely vary-

ing input voltage or output load conditions. Finally, it

allows simple pulse-by-pulse current limiting to provide

maximum switch protection under output overload or

short conditions. A low dropout internal regulator pro-

vides a 2.3V supply for all internal circuitry. This low

dropout design allows input voltage to vary from 2.7V to

25V with virtually no change in device performance. A

500kHz(LT1372)or1MHz(LT1377)oscillatoristhebasic

clock for all internal timing. It turns “On” the output switch

via the logic and driver circuitry. Special adaptive anti-sat

circuitry detects onset of saturation in the power switch

and adjusts driver current instantaneously to limit switch

saturation. Thisminimizesdriverdissipationandprovides

very rapid turn-off of the switch.

put overshoot on start-up or overload recovery. When

the feedback voltage exceeds the reference by 40mV,

error amplifier transconductance increases ten times,

whichreducesoutputovershoot.Thefeedbackinputalso

invokes oscillator frequency shifting, which helps pro-

tect components during overload conditions. When the

feedback voltage drops below 0.6V, the oscillator fre-

quencyisreduced5:1.Lowerswitchingfrequencyallows

full control of switch current limit by reducing minimum

switch duty cycle.

UniqueerroramplifiercircuitryallowstheLT1372/LT1377

to directly regulate negative output voltages. The negative

feedback amplifier’s 100k source resistor is brought out

fornegativeoutputvoltagesensing. TheNFBpinregulates

at –2.49V while the amplifier output internally drives the

FB pin to 1.245V. This architecture, which uses the same

main error amplifier, prevents duplicating functions and

maintainseaseofuse. ConsultLinearTechnologyMarket-

ing for units that can regulate down to –1.25V.

The error signal developed at the amplifier output is

brought out externally. This pin (V_C) has three different

functions. Itisusedforfrequencycompensation, current

limit adjustment and soft starting. During normal regula-

tor operation this pin sits at a voltage between 1V (low

outputcurrent)and 1.9V(highoutputcurrent). Theerror

amplifierisacurrentoutput(g_m)type, sothisvoltagecan

be externally clamped for lowering current limit. Like-

wise, acapacitorcoupledexternalclampwillprovidesoft

start. Switch duty cycle goes to zero if the V_Cpin is pulled

below the control pin threshold, placing the LT1372/

LT1377 in an idle mode.

A 1.245V bandgap reference biases the positive input of

the error amplifier. The negative input of the amplifier is

broughtoutforpositiveoutputvoltagesensing.Theerror

amplifier has nonlinear transconductance to reduce out-

U U

APPLICATIO S I FOR ATIO

Positive Output Voltage Setting

OUT

The LT1372/LT1377 develops a 1.245V reference (V_REF

)

= V

1 +

OUT

REF

(

)

from the FB pin to ground. Output voltage is set by

connecting the FB pin to an output resistor divider

(Figure 1). The FB pin bias current represents a small

errorandcanusuallybeignoredforvaluesofR2upto7k.

The suggested value for R2 is 6.19k. The NFB pin is

normally left open for positive output applications.

OUT

1.245

R1 = R2

– 1

(

)

REF

LT1372 • F01

Figure 1. Positive Output Resistor Divider

LT1372/LT1377

U U

APPLICATIO S I FOR ATIO

Positive fixed voltage versions are available (consult

Linear Technology marketing).

Shutdown and Synchronization

The dual function S/S pin provides easy shutdown and

synchronization. It is logic level compatible and can be

pulledhigh, tiedtoV_INorleftfloatingfornormaloperation.

A logic low on the S/S pin activates shutdown, reducing

the part’s supply current to 12µA. Typical synchronization

rangeisfrom1.05to1.8timesthepart’snaturalswitching

frequency, but is only guaranteed between 600kHz and

800kHz (LT1372) or 1.2MHz and 1.6MHz (LT1377). A

12µs resetable shutdown delay network guarantees the

part will not go into shutdown while receiving a synchro-

nization signal.

Negative Output Voltage Setting

The LT1372/LT1377 develops a –2.49V reference (V_NFR

)

from the NFB pin to ground. Output voltage is set by

connecting the NFB pin to an output resistor divider

(Figure 2). The –30µA NFB pin bias current (I_NFB) can

cause output voltage errors and should not be ignored.

ThishasbeenaccountedforintheformulainFigure2. The

suggested value for R2 is 2.49k. The FB pin is normally left

open for negative output application. See Dual Polarity

Output Voltage Sensing for limitatins on FB pin loading

when using the NFB pin.

Cautionshouldbeusedwhensynchronizingabove700kHz

(LT1372) or 1.4MHz (LT1377) because at higher sync

frequenciestheamplitudeoftheinternalslopecompensa-

tion used to prevent subharmonic switching is reduced.

This type of subharmonic switching only occurs when the

duty cycle of the switch is above 50%. Higher inductor

values will tend to eliminate problems.

–V

OUT

–V

= V

NFB

1 +

+ I

(R1)

NFB

OUT

(

)

NFB

– 2.49

OUT

R1 =

2.49

30 × 10^–

NFR

(

)

(

)

LT1372 • F02

Thermal Considerations

Figure 2. Negative Output Resistor Divider

Care should be taken to ensure that the worst-case input

voltage and load current conditions do not cause exces-

sive die temperatures. The packages are rated at 120°C/W

for SO (S8) and 130°C/W for PDIP (N8).

Dual Polarity Output Voltage Sensing

Certain applications benefit from sensing both positive

and negative output voltages. One example is the “Dual

Output Flyback Converter with Overvoltage Protection”

circuit shown in the Typical Applications section. Each

output voltage resistor divider is individually set as de-

scribed above. When both the FB and NFB pins are used,

the LT1372/LT1377 acts to prevent either output from

going beyond its set output voltage. For example in this

application,ifthepositiveoutputweremoreheavilyloaded

than the negative, the negative output would be greater

and would regulate at the desired set-point voltage. The

positive output would sag slightly below its set-point

voltage. This technique prevents either output from going

unregulated high at no load. Please note that the load on

the FB pin should not exceed 250µA when the NFB pin is

used. This situation occurs when the resistor dividers are

used at both FB and NFB. True load on FB is not the full

divider current unless the positive output is shorted to

ground. See Dual Output Flyback Converter application.

Average supply current (including driver current) is:

I_IN= 4mA + DC (I_SW/60 + I_SW× 0.004)

I_SW= switch current

DC = switch duty cycle

Switch power dissipation is given by:

P_SW= (I_SW)²× R_SW× DC

R_SW= output switch “On” resistance

Total power dissipation of the die is the sum of supply

current times supply voltage plus switch power:

P_D(TOTAL)= (I_IN× V_IN) + P_SW

LT1372/LT1377

U U

APPLICATIO S I FOR ATIO

Choosing the Inductor

radiation, or whether it needs a closed core like a toroid

to prevent EMI problems. One would not want an open

core next to a magnetic storage media for instance!

This is a tough decision because the rods or barrels are

temptingly cheap and small, and there are no helpful

guidelines to calculate when the magnetic field radia-

tion will be a problem.

For most applications the inductor will fall in the range of

2.2µH to 22µH. Lower values are chosen to reduce physi-

cal size of the inductor. Higher values allow more output

current because they reduce peak current seen by the

power switch, which has a 1.5A limit. Higher values also

reduce input ripple voltage and reduce core loss.

4. Start shopping for an inductor which meets the re-

quirements of core shape, peak current (to avoid

saturation), averagecurrent(tolimitheating)andfault

current.Iftheinductorgetstoohot,wireinsulationwill

melt and cause turn-to-turn shorts. Keep in mind that

allgoodthingslikehighefficiency,lowprofileandhigh

temperature operation will increase cost, sometimes

dramatically.

When choosing an inductor you might have to consider

maximum load current, core and copper losses, allowable

component height, output voltage ripple, EMI, fault

current in the inductor, saturation, and of course, cost.

Thefollowingprocedureissuggestedasawayofhandling

thesesomewhatcomplicatedandconflictingrequirements.

1. Assume that the average inductor current for a boost

converter is equal to load current times V_OUT/V_INand

decide whether or not the inductor must withstand

continuous overload conditions. If average inductor

current at maximum load current is 0.5A, for instance,

a 0.5A inductor may not survive a continuous 1.5A

overload condition. Also be aware that boost convert-

ers are not short circuit protected, and that under

outputshortconditions, inductorcurrentislimitedonly

by the available current of the input supply.

5. After making an initial choice, consider the secondary

things like output voltage ripple, second sourcing, etc.

Use the experts in the Linear Technology application

department if you feel uncertain about the final choice.

They have experience with a wide range of inductor

types and can tell you about the latest developments in

low profile, surface mounting, etc.

Output Capacitor

2. Calculate peak inductor current at full load current to

ensure that the inductor will not saturate. Peak current

can be significantly higher than output current, espe-

cially with smaller inductors and lighter loads, so don’t

omit this step. Powdered iron cores are forgiving be-

cause they saturate softly, whereas ferrite cores satu-

rate abruptly. Other core materials fall in between

somewhere. The following formula assumes continu-

ous mode operation but it errors only slightly on the

high side for discontinuous mode, so it can be used for

all conditions.

The output capacitor is normally chosen by its effective

seriesresistance,(ESR),becausethisiswhatdetermines

output ripple voltage. At 500kHz, any polarized capacitor

is essentially resistive. To get low ESR takes volume, so

physically smaller capacitors have high ESR. The ESR

range for typical LT1372 and LT1377 applications is

0.05Ω to 0.5Ω. A typical output capacitor is an AVX type

TPS, 22µF at 25V, with a guaranteed ESR less than 0.2Ω.

This is a “D” size surface mount solid tantalum capacitor.

TPS capacitors are specially constructed and tested for

low ESR, so they give the lowest ESR for a given volume.

To further reduce ESR, multiple output capacitors can be

used in parallel. The value in microfarads is not particu-

larly critical, and values from 22µF to greater than 500µF

work well, but you cannot cheat mother nature on ESR.

Ifyoufindatiny22µFsolidtantalumcapacitor,itwillhave

high ESR, and output ripple voltage will be terrible. Table

1 shows some typical solid tantalum surface mount

capacitors.

V (V

2(f)(L)(V

– V )

OUT

IN OUT IN

= I

OUT

PEAK

)

OUT

V = Minimum Input Voltage

f = 500kHz Switching Frequency (LT1372) or

1MHz Switching Frequency (LT1377)

3. Decide if the design can tolerate an “open” core geom-

etry like a rod or barrel, which have high magnetic field

LT1372/LT1377

U U

APPLICATIO S I FOR ATIO

Table 1. Surface Mount Solid Tantalum Capacitor

ESR and Ripple Current

0.3(V )(V

– V )

)

IN OUT

RIPPLE

E CASE SIZE

ESR (MAX Ω)

RIPPLE CURRENT (A)

(f)(L)(V

OUT

AVX TPS, Sprague 593D

AVX TAJ

0.1 to 0.3

0.7 to 0.9

0.7 to 1.1

0.4

f = 500kHz Switching frequency (LT1372) or,

1MHz Switching frequency (LT1377)

D CASE SIZE

AVX TPS, Sprague 593D

AVX TAJ

0.1 to 0.3

0.9 to 2.0

0.7 to 1.1

0.36 to 0.24

Theinputcapacitorcanseeaveryhighsurgecurrentwhen

a battery or high capacitance source is connected “live”

andsolidtantalumcapacitorscanfailunderthiscondition.

Several manufacturers have developed a line of solid

tantalum capacitors specially tested for surge capability

(AVX TPS series, for instance), but even these units may

fail if the input voltage approaches the maximum voltage

rating of the capacitor. AVX recommends derating capaci-

torvoltageby2:1forhighsurgeapplications. Ceramicand

aluminum electrolytic capacitors may also be used and

have a high tolerance to turn-on surges.

C CASE SIZE

AVX TPS

AVX TAJ

0.2 (Typ)

1.8 to 3.0

0.5 (Typ)

0.22 to 0.17

B CASE SIZE

AVX TAJ

2.5 to 10

0.16 to 0.08

Many engineers have heard that solid tantalum capacitors

are prone to failure if they undergo high surge currents.

This is historically true and type TPS capacitors are

speciallytestedforsurgecapability,butsurgeruggedness

is not a critical issue with the output capacitor. Solid

tantalum capacitors fail during very high turn-on surges,

which do not occur at the output of regulators. High

discharge surges, such as when the regulator output is

dead shorted, do not harm the capacitors.

Ceramic Capacitors

Higher value, lower cost ceramic capacitors are now

becomingavailableinsmallercasesizes.Thesearetempt-

ing for switching regulator use because of their very low

ESR. Unfortunately, the ESR is so low that it can cause

loop stability problems. Solid tantalum capacitor ESR

generatesaloop“zero”at5kHzto50kHzthatisinstrumen-

tal in giving acceptable loop phase margin. Ceramic ca-

pacitors remain capacitive to beyond 300kHz and usually

resonate with their ESL before ESR becomes effective.

They are appropriate for input bypassing because of their

highripplecurrentratingsandtoleranceofturn-onsurges.

Linear Technology plans to issue a Design Note on the use

of ceramic capacitors in the near future.

Single inductor boost regulators have large RMS ripple

current in the output capacitor, which must be rated to

handle the current. The formula to calculate this is:

Output Capacitor Ripple Current (RMS)

1 – DC

(RMS) = I

= I

RIPPLE

OUT

– V

OUT

Input Capacitors

Output Diode

The input capacitor of a boost converter is less critical due

tothefactthattheinputcurrentwaveformistriangularand

does not contain large squarewave currents as is found in

the output capacitor. Capacitors in the range of 10µF to

100µFwithanESRof0.3Ω orlessworkwelluptofull1.5A

switch current. Higher ESR capacitors may be acceptable

at low switch currents. Input capacitor ripple current for

boost converter is :

The suggested output diode (D1) is a 1N5818 Schottky or

its Motorola equivalent, MBR130. It is rated at 1A average

forward current and 30V reverse voltage. Typical forward

voltage is 0.42V at 1A. The diode conducts current only

during switch off time. Peak reverse voltage for boost

converters is equal to regulator output voltage. Average

forward current in normal operation is equal to output

current.

LT1372/LT1377

U U

APPLICATIO S I FOR ATIO

Frequency Compensation

(magnetic) radiation is minimized by keeping output di-

ode, switch pin, and output bypass capacitor leads as

short as possible. E field radiation is kept low by minimiz-

ingthelengthandareaofalltracesconnectedtotheswitch

pin. A ground plane should always be used under the

switcher circuitry to prevent interplane coupling.

Loopfrequencycompensationisperformedontheoutput

of the error amplifier (V_Cpin) with a series RC network.

The main pole is formed by the series capacitor and the

output impedance (≈500kΩ) of the error amplifier. The

pole falls in the range of 2Hz to 20Hz. The series resistor

creates a “zero” at 1kHz to 5kHz, which improves loop

stability and transient response. A second capacitor,

typically one-tenth the size of the main compensation

capacitor, is sometimes used to reduce the switching

frequency ripple on the V_Cpin. V_Cpin ripple is caused by

output voltage ripple attenuated by the output divider and

multiplied by the error amplifier. Without the second

capacitor, V_Cpin ripple is:

Thehighspeedswitchingcurrentpathisshownschemati-

cally in Figure 3. Minimum lead length in this path is

essential to ensure clean switching and low EMI. The path

including the switch, output diode, and output capacitor is

the only one containing nanosecond rise and fall times.

Keep this path as short as possible.

SWITCH

NODE

OUT

1.245(V

)(g )(R )

m C

RIPPLE

HIGH

FREQUENCY

CIRCULATING

PATH

V Pin Ripple =

LOAD

)

OUT

= Output ripple (V

)

P–P

RIPPLE

g = Error amplifier transconductance

≈(1500µmho)

LT1372 • F03

Figure 3

R = Series resistor on V pin

OUT

= DC output voltage

More Help

To prevent irregular switching, V_Cpin ripple should be

kept below 50mV_P–P. Worst-case V_Cpin ripple occurs at

maximum output load current and will also be increased

if poor quality (high ESR) output capacitors are used. The

addition of a 0.0047µF capacitor on the V_Cpin reduces

switching frequency ripple to only a few millivolts. A low

value for R_Cwill also reduce V_Cpin ripple, but loop phase

margin may be inadequate.

For more detailed information on switching regulator

circuits, please see Application Note 19. Linear Technol-

ogyalsooffersacomputersoftwareprogram,SwitcherCAD,

to assist in designing switching converters. SwitcherCAD

will be updated in late 1995 for the LT1372 and LT1377. In

addition, our applications department is always ready to

lend a helping hand.

Switch Node Considerations

For maximum efficiency, switch rise and fall time are

made as short as possible. To prevent radiation and high

frequency resonance problems, proper layout of the com-

ponents connected to the switch node is essential. B field

LT1372/LT1377

TYPICAL APPLICATIONS N

Positive-to-Negative Converter with Direct Feedback

Dual Output Flyback Converter with Overvoltage Protection

1.21k

13k

2.7V TO 16V

T1*

•

22µF

P6KE-15A

47µF

MBRS140T3

T1*

2.7V TO 13V

•

1N4148

OUT

†

–V

OUT

15V

2, 3

S/S

OFF

22µF

–5V

2.49k

MBRS130LT3

•

P6KE-20A

LT1372/LT1377

NFB

GND

6, 7

47µF

1N4148

•

S/S

2.49k

6, 7

OFF

47µF

LT1372/LT1377

NFB

GND

6, 7

–V

OUT

0.047µF

*COILTRONICS CTX10-2 (407) 241-7876

†

–15V

12.1k

MAX I

OUT

MBRS140T3

0.0047µF

OUT

LT1372 • TA03

0.3A 3V

0.5A 5V

0.75A 9V

2.49k

0.047µF

0.0047µF

*DALE LPE-4841-100MB (605) 665-9301

LT1372 • TA04

90% Efficient CCFL Supply

Low Ripple 5V to –3V “Cuk”^†Converter

5mA MAX

LAMP

27pF

1N4148

OUT

L1*

–3V

250mA

4.5V

•

TO 30V

10µF

0.1µF

47µF

16V

22µF

10V

0.1µF

S/S

LT1372/LT1377

GND NFB

330Ω

33µH

2.7V TO

5.5V

GND S

47µF

16V

1N4148

D1**

1N5818

2.2µF

0.0047µF

4.99k

562Ω*

10k

S/S

OFF

0.047µF

20k

DIMMING

LT1372/LT1377

*SUMIDA CLS62-100L

LT1372 • TA05

**MOTOROLA MBR0520LT3

†

GND

6, 7

PATENTS MAY APPLY

0.1µF

22k

1N4148

2µF

OPTIONAL REMOTE

DIMMING

LT1372 • TA06

C1 = WIMA MKP-20

L1 = COILCRAFT DT3316-333

Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001

T1 = COILTRONICS CTX 110609

* = 1% FILM RESISTOR

CCFL BACKLIGHT APPLICATION CIRCUITS

CONTAINED IN THIS DATA SHEET ARE

COVERED BY U.S. PATENT NUMBER 5408162

AND OTHER PATENTS PENDING

DO NOT SUBSTITUTE COMPONENTS

COILTRONICS (407) 241-7876

COILCRAFT (708) 639-6400

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tation that the interconnection of circuits as described herein will not infringe on existing patent rights.

LT1372/LT1377

TYPICAL APPLICATIONS N

2 Li-Ion Cell to 5V SEPIC Converter

4V TO 9V

L1A*

10µH

MBRS130LT3

†

•

OUT

C1 = AVX TPSD 336M020R0200

C2 = TOKIN 1E105ZY5U-C103-F

C3 = AVX TPSD107M010R0100

S/S

OFF

33µF

20V

1µF

LT1372/LT1377

18.7k

*SINGLE INDUCTOR WITH TWO WINDINGS

•

100µF

GND

6, 7

COILTRONICS CTX10-1

†

10V

L1B*

10µH

MAX I

OUT

6.19k

OUT

0.45A 4V

0.55A 5V

0.65A 7V

0.72A 9V

0.0047µF

0.047µF

LT1372 • TA07

PACKAGE DESCRIPTION ^{Dimensions in inches (millimeters) unless otherwise noted.}

N8 Package

8-Lead PDIP (Narrow 0.300)

(LTC DWG # 05-08-1510)

0.400*

(10.160)

MAX

0.130 ± 0.005

0.300 – 0.325

0.045 – 0.065

(3.302 ± 0.127)

(1.143 – 1.651)

(7.620 – 8.255)

0.065

(1.651)

TYP

0.255 ± 0.015*

(6.477 ± 0.381)

0.009 – 0.015

(0.229 – 0.381)

+0.025

0.125

(3.175)

MIN

0.005

(0.127)

MIN

0.015

(0.380)

MIN

0.325

–0.015

+0.635

8.255

N8 0695

0.100 ± 0.010

(2.540 ± 0.254)

0.018 ± 0.003

(0.457 ± 0.076)

(

)

–0.381

*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.

MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.189 – 0.197*

(4.801 – 5.004)

0.010 – 0.020

(0.254 – 0.508)

× 45°

0.053 – 0.069

(1.346 – 1.752)

0.004 – 0.010

(0.101 – 0.254)

0.008 – 0.010

(0.203 – 0.254)

0°– 8° TYP

0.150 – 0.157**

(3.810 – 3.988)

0.228 – 0.244

(5.791 – 6.197)

0.016 – 0.050

0.406 – 1.270

0.050

(1.270)

BSC

0.014 – 0.019

(0.355 – 0.483)

*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

SO8 0695

**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

RELATED PARTS

PART NUMBER

DESCRIPTION

COMMENTS

Good for Up to V = 40V

LT1172

100kHz 1.25A Boost Switching Regulator

12V 1.2A Monolithic Buck Converter

Micropower 2A Boost Converter

LTC^®1265

LT1302

Converts 5V to 3.3V at 1A with 90% Efficiency

Converts 2V to 5V at 600mA in SO8 Packages

Steps Down from Up to 25V Using 4.7µH Inductors

90% Efficient Boost Converter with Constant Frequency

LT1376

500kHz 1.5A Buck Switching Regulator

Low Supply Current 250kHz 1.5A Boost Switching Regulator

LT1373

LT/GP 0996 5K REV A • PRINTED IN THE USA

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

●

(408) 432-1900 FAX: (408) 434-0507 TELEX: 499-3977

LINEAR TECHNOLOGY CORPORATION 1995

LT1372IS8 [Linear]

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