LTC1772IS6 [Linear]
Constant Frequency Current Mode Step-Down DC/DC Controller in SOT-23; 恒定频率电流模式降压型DC / DC采用SOT -23控制器
| 型号： | LTC1772IS6 |
| 厂家： | 
Linear |
| 描述： | Constant Frequency Current Mode Step-Down DC/DC Controller in SOT-23 |
| 文件： | 总12页 (文件大小：160K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC1772
Constant Frequency
Current Mode Step-Down
DC/DC Controller in SOT-23
U
DESCRIPTIO
FEATURES
■
High Efficiency: Up to 94%
The LTC®1772 is a constant frequency current mode step-
downDC/DCcontrollerprovidingexcellentACandDCload
and line regulation. The device incorporates an accurate
undervoltagelockoutfeaturethatshutsdowntheLTC1772
when the input voltage falls below 2.0V.
■
High Output Currents Easily Achieved
■
Wide VIN Range: 2.5V to 9.8V
■
Constant Frequency 550kHz Operation
Burst Mode® Operation at Light Load
■
■
Low Dropout: 100% Duty Cycle
Tiny 6-Lead SOT-23 Package
0.8V Reference Allows Low Output Voltages
The LTC1772 provides a ±2.5% output voltage accuracy
and consumes only 270µA of quiescent current. For
applications where efficiency is a prime consideration, the
LTC1772 is configured for Burst Mode operation, which
enhances efficiency at low output current.
■
■
■
Current Mode Operation for Excellent Line and Load
Transient Response
■
Low Quiescent Current: 270µA
■
■
To further maximize the life of a battery source, the
external P-channel MOSFET is turned on continuously in
dropout (100%dutycycle).In shutdown, the device draws
a mere 8µA. High constant operating frequency of 550kHz
allows the use of a small external inductor.
Shutdown Mode Draws Only 8µA Supply Current
±2.5% Reference Accuracy
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APPLICATIO S
■
One or Two Lithium-Ion-Powered Applications
The LTC1772 is available in a small footprint 6-lead
SOT-23.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
■
Cellular Telephones
■
Wireless Modems
■
Portable Computers
■
Distributed 3.3V, 2.5V or 1.8V Power Systems
■
Scanners
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TYPICAL APPLICATIO
V
Efficiency vs Load Current
IN
2.5V
100
90
80
70
60
50
40
C1
TO 9.8V
R1
0.03Ω
V
= 4.2V
IN
10µF
V
= 3.3V
IN
10V
1
6
L1
4.7µH
I
/RUN PGATE
LTC1772
M1
TH
V
2.5V
2A
OUT
10k
220pF
V = 6V
IN
+
C2A
47µF
6V
C2B
1µF
10V
2
3
5
4
GND
V
D1
IN
–
V
= 8.4V
IN
174k
V
= 9.8V
IN
V
SENSE
FB
C1: TAIYO YUDEN LMK325BJ106K-T
C2A: SANYO 6TPA47M
C2B: AVX 0805ZC105KAT1A
D1: MOTOROLA MBRM120T3
L1: MURATA LQN6C-4R7
80.6k
V
= 2.5V
OUT
1
10
100
1000
10000
1772 F01a
M1: FAIRCHILD FDC638P
R1: IRC LRC-LR1206-01-R030F
LOAD CURRENT (mA)
1772 F01b
1772fb
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Figure 1. High Efficiency, High Output Current 2.5V/2A Regulator
LTC1772
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
ORDER PART NUMBER
LTC1772CS6
Input Supply Voltage (VIN).........................–0.3V to 10V
SENSE–, PGATE Voltages ............. –0.3V to (VIN + 0.3V)
VFB, ITH /RUN Voltages .............................–0.3V to 2.4V
PGATE Peak Output Current (<10µs) ......................... 1A
Storage Ambient Temperature Range ... –65°C to 150°C
Operating Temperature Range
LTC1772CS6 ........................................... 0°C to 70°C
LTC1772ES6 (Note 2) ........................ –40°C to 85°C
LTC1772IS6 (Note 2) ......................... –40°C to 85°C
LTC1772HS6 (Notes 2,3) ................. –40°C to 140°C
Junction Temperature (Note 3)............................. 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
TOP VIEW
LTC1772ES6
LTC1772IS6
LTC1772HS6
I
/RUN
GND
1
2
3
6
5
4
PGATE
TH
V
IN
–
V
FB
SENSE
S6 PART MARKING
S6 PACKAGE
6-LEAD PLASTIC SOT-23
JA = 230°C/ W
LTIL
θ
LTIM
LTB7
LTBRY
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
The
●
denotes specifications that apply over the full operating temperature
ELECTRICAL CHARACTERISTICS
range, otherwise specifications are at T = 25°C. V = 4.2V unless otherwise specified. (Note 2)
A
IN
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input DC Supply Current
Normal Operation
Sleep Mode
Shutdown
UVLO
Typicals at V = 4.2V (Note 4)
IN
2.4V ≤ V ≤ 9.8V
270
230
8
420
370
22
µA
µA
µA
µA
IN
2.4V ≤ V ≤ 9.8V
IN
2.4V ≤ V ≤ 9.8V, V /RUN = 0V
IN
ITH
V
< UVLO Threshold
6
10
IN
Undervoltage Lockout Threshold
V
V
V
Falling (LTC1772C)
Falling (LTC1772E, LTC1772I, LTC1772H)
Rising
●
●
1.60
1.55
1.85
2.00
2.00
2.10
2.30
2.35
2.40
V
V
V
IN
IN
IN
Shutdown Threshold (at I /RUN)
(LTC1772C)
(LTC1772E, LTC1772I, LTC1772H)
●
●
0.20
0.15
0.35
0.35
0.50
0.55
V
V
TH
Start-Up Current Source
V
/RUN = 0V
ITH
0.25
0.5
0.85
µA
Regulated Feedback Voltage
(Note 5) (LTC1772C)
(Note 5) (LTC1772E, LTC1772I, LTC1772H)
●
●
0.780
0.770
0.800
0.800
0.820
0.830
V
V
Output Voltage Line Regulation
Output Voltage Load Regulation
2.4V ≤ V ≤ 9.8V (Note 5)
0.05
mV/V
IN
I
I
/RUN Sinking 5µA (Note 5)
TH
/RUN Sourcing 5µA (Note 5)
TH
2.5
2.5
mV/µA
mV/µA
V
Input Current
(Note 5)
10
0.860
20
50
nA
V
FB
Overvoltage Protect Threshold
Overvoltage Protect Hysteresis
Oscillator Frequency
Measured at V
0.820
500
0.895
FB
mV
V
V
= 0.8V
= 0V
550
120
650
kHz
kHz
FB
FB
Gate Drive Rise Time
Gate Drive Fall Time
C
C
= 3000pF
= 3000pF
40
40
ns
ns
LOAD
LOAD
Peak Current Sense Voltage
(Note 6)
120
mV
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LTC1772
ELECTRICAL CHARACTERISTICS
Note 1: Absolute Maximum Ratings are those values beyond which the life
Operation at high junction temperatures degrades operating lifetimes.
of a device may be impaired.
Operating lifetimes at junction temperatures greater than 125°C is derated
to 1000 hours.
Note 2: The LTC1772E is guaranteed to meet specifications from 0°C to
70°C. Specifications over the –40°C to 85°C operating temperature range
are assured by design, characterization and correlation with statistical
process controls. The LTC1772I is guaranteed to meet specified
performance from –40°C to 85°C. The LTC1772H is guaranteed to meet
specified performance from –40°C to 140°C.
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC1772 is tested in a feedback loop that servos V to the
FB
output of the error amplifier.
Note 6: Peak current sense voltage is reduced dependent on duty cycle to
a percentage of value as given in Figure 2.
Note 3: T is calculated from the ambient temperature T and power
J
A
dissipation P according to the following formula:
D
T = T + (P • θ °C/W)
J
A
D
JA
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TYPICAL PERFOR A CE CHARACTERISTICS
Reference Voltage
vs Temperature
Normalized Frequency
vs Temperature
Undervoltage Lockout Trip
Voltage vs Temperature
2.14
2.10
2.06
2.02
1.98
1.94
1.90
1.86
1.82
1.78
1.74
825
820
815
810
805
800
795
790
785
780
775
10
8
V
FALLING
V
IN
= 4.2V
V
= 4.2V
IN
IN
6
4
2
0
–2
–4
–6
–8
–10
–55 –35 –15
5
25 45 65 85 105 125 145
–55 –35 –15
5
25 45 65 85 105 125 145
–55 –35 –15
5
25 45 65 85 105 125 145
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
1772 G03
1772 G01
1772 G02
–
Maximum (V – SENSE ) Voltage
Shutdown Threshold
vs Temperature
IN
vs Duty Cycle
130
550
510
470
430
390
350
310
270
230
190
150
V
A
= 4.2V
V
= 4.2V
IN
= 25°C
IN
T
120
110
100
90
80
70
60
50
20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
–55 –35 –15
5
25 45 65 85 105 125 145
TEMPERATURE (°C)
1772 G04
1772 G05
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LTC1772
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PIN FUNCTIONS
ITH/RUN (Pin 1): This pin performs two functions. It
VFB (Pin 3): Receives the feedback voltage from an exter-
servesastheerroramplifiercompensationpointaswellas nal resistive divider across the output.
the run control input. The current comparator threshold
SENSE– (Pin 4): The Negative Input to the Current Com-
increases with this control voltage. Nominal voltage range
for this pin is 0.7V to 1.9V. Forcing this pin below 0.35V
causes the device to be shut down. In shutdown all
functions are disabled and the PGATE pin is held high.
parator.
VIN (Pin5): SupplyPin. MustbecloselydecoupledtoGND
Pin 2.
PGATE (Pin 6): Gate Drive for the External P-Channel
MOSFET. This pin swings from 0V to VIN.
GND (Pin 2): Ground Pin.
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FUNCTIONAL DIAGRA
–
V
SENSE
4
IN
5
+
–
ICMP
V
IN
RS1
PGATE
6
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
SLOPE
COMP
R
Q
S
OSC
–
+
FREQ
OVP
FOLDBACK
BURST
CMP
+
–
0.3V
+
SHORT-CIRCUIT
DETECT
V
+
REF
0.15V
SLEEP
60mV
–
V
IN
EAMP
V
REF
0.8V
+
–
0.5µA
V
FB
I
TH
/RUN
3
+
–
1
V
IN
V
IN
0.3V
0.35V
+
–
SHDN
UV
SHDN
CMP
VOLTAGE
REFERENCE
V
REF
0.8V
GND
2
UNDERVOLTAGE
LOCKOUT
1.2V
1772 FD
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LTC1772
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(Refer to Functional Diagram)
OPERATIO
Main Control Loop
oscillator cycle will turn the external MOSFET on and the
switching cycle repeats.
TheLTC1772isaconstantfrequencycurrentmodeswitch-
ing regulator. During normal operation, the external
P-channel power MOSFET is turned on each cycle when
the oscillator sets the RS latch (RS1) and turned off when
the current comparator (ICMP) resets the latch. The peak
inductor current at which ICMP resets the RS latch is
controlled by the voltage on the ITH/RUN pin, which is the
output of the error amplifier EAMP. An external resistive
divider connected between VOUT and ground allows the
EAMPtoreceiveanoutputfeedbackvoltageVFB.Whenthe
load current increases, it causes a slight decrease in VFB
relative to the 0.8V reference, which in turn causes the
ITH/RUN voltage to increase until the average inductor
current matches the new load current.
Dropout Operation
When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the ON cycle decreases. This reduction means that
the external P-channel MOSFET will remain on for more
thanoneoscillatorcyclesincetheinductorcurrenthasnot
ramped up to the threshold set by EAMP. Further reduc-
tion in input supply voltage will eventually cause the
P-channel MOSFET to be turned on 100%, i.e., DC. The
outputvoltagewillthenbedeterminedbytheinputvoltage
minus the voltage drop across the MOSFET, the sense
resistor and the inductor.
ThemaincontrolloopisshutdownbypullingtheITH/RUN
pin low. Releasing ITH/RUN allows an internal 0.5µA
current source to charge up the external compensation
network. When the ITH/RUN pin reaches 0.35V, the main
control loop is enabled with the ITH/RUN voltage then
pulled up to its zero current level of approximately 0.7V.
Astheexternalcompensationnetworkcontinuestocharge
up, the corresponding output current trip level follows,
allowing normal operation.
Undervoltage Lockout
To prevent operation of the P-channel MOSFET below safe
input voltage levels, an undervoltage lockout is incorpo-
rated into the LTC1772. When the input supply voltage
drops below approximately 2.0V, the P-channel MOSFET
and all circuitry is turned off except the undervoltage block,
which draws only several microamperes.
Short-Circuit Protection
Comparator OVP guards against transient overshoots
>7.5% by turning off the external P-channel power
MOSFET and keeping it off until the fault is removed.
Whentheoutputisshortedtoground, thefrequencyofthe
oscillator will be reduced to about 120kHz. This lower
frequency allows the inductor current to safely discharge,
thereby preventing current runaway. The oscillator’s fre-
quency will gradually increase to its designed rate when
the feedback voltage again approaches 0.8V.
Burst Mode Operation
The LTC1772 enters Burst Mode operation at low load
currents. In this mode, the peak current of the inductor is
set as if VITH/RUN = 1V (at low duty cycles) even though
the voltage at the ITH/RUN pin is at a lower value. If the
inductor’saveragecurrentisgreaterthantheloadrequire-
ment, the voltage at the ITH/RUN pin will drop. When the
ITH/RUN voltage goes below 0.85V, the sleep signal goes
high, turning off the external MOSFET. The sleep signal
goes low when the ITH/RUN voltage goes above 0.925V
and the LTC1772 resumes normal operation. The next
Overvoltage Protection
As a further protection, the overvoltage comparator in the
LTC1772 will turn the external MOSFET off when the
feedback voltage has risen 7.5% above the reference
voltage of 0.8V. This comparator has a typical hysteresis
of 20mV.
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LTC1772
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(Refer to Functional Diagram)
OPERATIO
Slope Compensation and Inductor’s Peak Current
110
100
90
80
70
60
50
40
30
20
10
The inductor’s peak current is determined by:
V
10 R
ITH – 0.7
IPK
=
(
)
SENSE
I
= 0.4I
PK
when the LTC1772 is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope com-
pensation begins and effectively reduces the peak induc-
torcurrent. Theamountofreductionisgivenbythecurves
in Figure 2.
RIPPLE
AT 5% DUTY CYCLE
I
= 0.2I
RIPPLE
PK
AT 5% DUTY CYCLE
V
IN
= 4.2V
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
1772 F02
Figure 2. Maximum Output Current vs Duty Cycle
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APPLICATIONS INFORMATION
ThebasicLTC1772applicationcircuitisshownin Figure 1.
External component selection is driven by the load re-
quirement and begins with the selection of L1 and RSENSE
(= R1). Next, the power MOSFET, M1 and the output diode
D1 are selected followed by CIN (= C1) and COUT (= C2).
However,foroperationthatisabove40%dutycycle,slope
compensation effect has to be taken into consideration to
selecttheappropriatevaluetoprovidetherequiredamount
of current. Using Figure 2, the value of RSENSE is:
SF
RSENSE
=
RSENSE Selection for Output Current
(10)(IOUT )(100)
RSENSE is chosen based on the required output current.
Withthecurrentcomparatormonitoringthevoltagedevel-
oped across RSENSE, the threshold of the comparator
determines the inductor’s peak current. The output cur-
rent the LTC1772 can provide is given by:
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripplecurrent. However, thisisattheexpenseofefficiency
due to an increase in MOSFET gate charge losses.
0.12
RSENSE
IRIPPLE
IOUT
=
−
2
The inductance value also has a direct effect on ripple
current. The ripple current, IRIPPLE, decreases with higher
inductance or frequency and increases with higher VIN or
VOUT. The inductor’s peak-to-peak ripple current is given
by:
where IRIPPLE is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section).
A reasonable starting point for setting ripple current is
IRIPPLE = (0.4)(IOUT). Rearranging the above equation, it
becomes:
V − VOUT ⎛ VOUT + VD ⎞
IN
IRIPPLE
=
⎜
⎟
f(L) ⎝ V + VD ⎠
IN
1
RSENSE
=
for Duty Cycle < 40%
(10)(IOUT
)
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LTC1772
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APPLICATIONS INFORMATION
wherefistheoperatingfrequency.Acceptinglargervalues
of IRIPPLE allows the use of low inductances, but results in
higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is
Molypermalloy (from Magnetics, Inc.) is a very good, low
losscorematerialfortoroids,butitismoreexpensivethan
ferrite. A reasonable compromise from the same manu-
facturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Be-
cause they generally lack a bobbin, mounting is more
difficult. However, new designs for surface mount that do
not increase the height significantly are available.
I
RIPPLE =0.4(IOUT(MAX)).Remember,themaximumIRIPPLE
occurs at the maximum input voltage.
In Burst Mode operation on the LTC1772, the ripple
current is normally set such that the inductor current is
continuous during the burst periods. Therefore, the peak-
to-peak ripple current must not exceed:
Power MOSFET Selection
An external P-channel power MOSFET must be selected
for use with the LTC1772. The main selection criteria for
the power MOSFET are the threshold voltage VGS(TH) and
the “on” resistance RDS(ON), reverse transfer capacitance
CRSS and total gate charge.
0.03
IRIPPLE
≤
RSENSE
This implies a minimum inductance of:
Since the LTC1772 is designed for operation down to low
inputvoltages,asublogiclevelthresholdMOSFET(RDS(ON)
guaranteed at VGS = 2.5V) is required for applications that
workclosetothisvoltage.WhentheseMOSFETsareused,
makesurethattheinputsupplytotheLTC1772islessthan
the absolute maximum VGS rating, typically 8V.
V − VOUT ⎛ VOUT + VD ⎞
IN
LMIN
=
⎜
⎟
⎝
⎠
⎛ 0.03 ⎞ V + VD
IN
f
⎜
⎟
⎝RSENSE
⎠
(Use VIN(MAX) = VIN)
A smaller value than LMIN could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
The required minimum RDS(ON) of the MOSFET is gov-
erned by its allowable power dissipation. For applications
that may operate the LTC1772 in dropout, i.e., 100% duty
cycle, at its worst case the required RDS(ON) is given by:
Inductor Core Selection
PP
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot af-
ford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will in-
crease. Ferrite designs have very low core losses and are
preferred at high switching frequencies, so design goals
canconcentrateoncopperlossandpreventingsaturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design cur-
rent is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!
RDS(ON)
=
2
DC=100%
I
1+ δp
(
)
(
)
OUT(MAX)
where PP is the allowable power dissipation and δp is the
temperature dependency of RDS(ON). (1 + δp) is generally
given for a MOSFET in the form of a normalized RDS(ON) vs
temperature curve, but δp = 0.005/°C can be used as an
approximation for low voltage MOSFETs.
In applications where the maximum duty cycle is less than
100%andtheLTC1772isincontinuousmode,theRDS(ON)
is governed by:
PP
RDS(ON)
≅
2
DC I
1+ δp
(
)
(
)
OUT
where DC is the maximum operating duty cycle of the
LTC1772.
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LTC1772
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APPLICATIONS INFORMATION
Output Diode Selection
This formula has a maximum value at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is com-
monlyusedfordesignbecauseevensignificantdeviations
donotoffermuchrelief.Notethatcapacitormanufacturer’s
ripplecurrentratingsareoftenbasedon2000hoursoflife.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC1772, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As VIN approaches VOUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition the diode must
safelyhandleIPEAK atcloseto100%dutycycle. Therefore,
itisimportanttoadequatelyspecifythediodepeakcurrent
and average power dissipation so as not to exceed the
diode ratings.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:
Under normal load conditions, the average current con-
ducted by the diode is:
⎛ V − VOUT
⎞
⎟
IN
I =
D
I
OUT
⎜
⎝ V + VD ⎠
IN
⎛
⎝
1
⎞
∆VOUT ≈ IRIPPLE ESR +
⎜
⎟
⎠
4fCOUT
The allowable forward voltage drop in the diode is calcu-
lated from the maximum short-circuit current as:
where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the induc-
tor. The output ripple is highest at maximum input voltage
since ∆IL increases with input voltage.
PD
VF ≈
ISC(MAX)
where PD is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.
A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forwarddropandfastswitchingtimes. Remembertokeep
lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surfacemountconfigurations. Inthecaseoftantalum, itis
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS, AVX TPSV and KEMET T510 series of surface mount
tantalum, available in case heights ranging from 2mm to
4mm. Other capacitor types include Sanyo OS-CON,
CIN and COUT Selection
In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (VOUT + VD)/
(VIN + VD). To prevent large voltage transients, a low ESR
input capacitor sized for the maximum RMS current must
beused. ThemaximumRMScapacitorcurrentisgivenby:
1/2
]
V
V − V
OUT
(
)
[
OUT IN
CIN Required IRMS ≈ IMAX
V
IN
Nichicon PL series and Panasonic SP.
1772fb
8
LTC1772
U
W U U
APPLICATIONS INFORMATION
Low Supply Operation
Efficiency Considerations
Although the LTC1772 can function down to approxi-
mately 2V, the maximum allowable output current is
reducedwhenVIN decreasesbelow3V. Figure3showsthe
amount of change as the supply is reduced down to 2V.
Also shown in Figure 3 is the effect of VIN on VREF as VIN
goes below 2.3V.
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
oftenusefultoanalyzeindividuallossestodeterminewhat
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (η1 + η2 + η3 + ...)
105
where η1, η2, etc. are the individual losses as a percent-
V
REF
age of input power.
100
95
90
85
80
75
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1772 circuits: 1) LTC1772 DC bias current,
2) MOSFET gate charge current, 3) I2R losses and 4)
voltage drop of the output diode.
V
ITH
1. The VIN current is the DC supply current, given in the
electricalcharacteristics, thatexcludesMOSFETdriver
and control currents. VIN current results in a small loss
which increases with VIN.
2.0
2.2
2.4
2.6
2.8
3.0
INPUT VOLTAGE (V)
1772 F03
Figure 3. Line Regulation of V
and V
2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each time
a MOSFET gate is switched from low to high to low
again,apacketofchargedQmovesfromVIN toground.
The resulting dQ/dt is a current out of VIN which is
typically much larger than the DC supply current. In
continuous mode, IGATECHG = f(Qp).
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but
is “chopped” between the P-channel MOSFET (in se-
ries with RSENSE) and the output diode. The MOSFET
RDS(ON) plus RSENSE multiplied by duty cycle can be
summedwiththeresistancesofLandRSENSE toobtain
I2R losses.
REF
ITH
Setting Output Voltage
The LTC1772 develops a 0.8V reference voltage between
thefeedback(Pin3)terminalandground(seeFigure4).By
selecting resistor R1, a constant current is caused to flow
through R1 and R2 to set the overall output voltage. The
regulated output voltage is determined by:
R2
R1
⎛
⎝
⎞
⎟
⎠
VOUT = 0.8 1+
⎜
Formostapplications, an80kresistorissuggestedforR1.
To prevent stray pickup, locate resistors R1 and R2 close
to LTC1772.
V
OUT
4. The output diode is a major source of power loss at
high currents and gets worse at high input voltages.
The diode loss is calculated by multiplying the forward
voltage times the diode duty cycle multiplied by the
load current. For example, assuming a duty cycle of
50% with a Schottky diode forward voltage drop of
R2
R1
LTC1772
V
3
FB
1772 F04
Figure 4. Setting Output Voltage
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9
LTC1772
U
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APPLICATIONS INFORMATION
0.4V, the loss increases from 0.5% to 8% as the load
current increases from 0.5A to 2A.
will be reduced to approximately 50% of the maximum
output current.
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1772. These items are illustrated graphically in the
layout diagram in Figure 6. Check the following in your
layout:
Transition Loss = 2(VIN)2IO(MAX) RSS
(f)
C
Other losses including CIN and COUT ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
1. IstheSchottkydiodecloselyconnectedbetweenground
(Pin 2) and drain of the external MOSFET?
Foldback Current Limiting
2. Does the (+) plate of CIN connect to the sense resistor
as closely as possible? This capacitor provides AC
current to the MOSFET.
AsdescribedintheOutputDiodeSelection,theworst-case
dissipation occurs with a short-circuited output when the
diode conducts the current limit value almost continu-
ously. To prevent excessive heating in the diode, foldback
current limiting can be added to reduce the current in
proportion to the severity of the fault.
3. Is the input decoupling capacitor (0.1µF) connected
closely between VIN (Pin 5) and ground (Pin 2)?
4. Connect the end of RSENSE as close to VIN (Pin 5) as
possible. The VIN pin is the SENSE+ of the current
comparator.
Foldbackcurrentlimitingisimplementedbyaddingdiodes
DFB1 and DFB2 between the output and the ITH/RUN pin as
shown in Figure 5. In a hard short (VOUT = 0V), the current
5. Is the trace from SENSE– (Pin 4) to the Sense resistor
kept short? Does the trace connect close to RSENSE
?
V
OUT
LTC1772
/RUN V
6. Keep the switching node PGATE away from sensitive
small signal nodes.
R2
R1
+
I
TH
FB
D
D
FB1
FB2
7. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1 and R2 must be
connected between the (+) plate of COUT and signal
ground.
1772 F05
Figure 5. Foldback Current Limiting
V
IN
1
6
5
4
+
I
/RUN PGATE
LTC1772
TH
C
IN
L1
R
SENSE
2
3
R
ITH
GND
V
IN
V
OUT
M1
+
0.1µF
D1
C
OUT
–
V
SENSE
C
FB
ITH
R1
R2
1772 F06
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 6. LTC1772 Layout Diagram (See PC Board Layout Checklist)
1772fb
10
LTC1772
U
TYPICAL APPLICATIO
LTC1772 High Efficiency, Small Footprint 3.3V to 1.8V/0.5A Regulator
V
IN
3.3V
C1
10µF
10V
R1
0.15Ω
1
6
L1
10µH
I
/RUN PGATE
LTC1772
M1
TH
V
1.8V
0.5A
OUT
R4
10k
+
C2
47µF
6V
2
3
5
4
GND
V
D1
IN
–
C3
220pF
V
SENSE
R2
100k
FB
C1: TAIYO YUDEN CERAMIC
LMK325BJ106K-T
C2: SANYO POSCAP 6TPA47M R1: DALE 0.25W
D1: MOTOROLA MBRM120T3
L1: COILTRONICS UP1B-100
M1: Si3443DV
R3
80.6k
1772 TA02
U
PACKAGE DESCRIPTION
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
2.90 BSC
(NOTE 4)
0.62
MAX
0.95
REF
1.22 REF
1.4 MIN
1.50 – 1.75
2.80 BSC
3.85 MAX 2.62 REF
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
DATUM ‘A’
0.01 – 0.10
1.00 MAX
0.30 – 0.50 REF
1.90 BSC
0.09 – 0.20
(NOTE 3)
S6 TSOT-23 0302
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
1772fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
11
LTC1772
U
TYPICAL APPLICATIONS
LTC1772 3.3V to 5V/1A Boost Regulator
R1
0.033Ω
V
IN
3.3V
C1
L1
47µF
16V
× 2
4.7µH
D1
V
5V
1A
OUT
U1
C2
5
3
+
1
6
2
4
100µF
10V
× 2
I
/RUN PGATE
LTC1772
M1
TH
R4
10k
2
3
5
4
GND
V
IN
–
C3
220pF
V
SENSE
R2
FB
422k
R3
C1: AVXTPSE476M016R0047 L1: MURATA LQN6C-4R7 U1: FAIRCHILD NC7SZ04
80.6k
C2: AVXTPSE107M010R0100 M1: Si9804
ALSO SEE LTC1872
D1: IR10BQ015
R1: DALE 0.25W
FOR THIS APPLICATION
1772 TA03
LTC1772 5V/0.5A Flyback Regulator
V
IN
2.5V
TO 9.8V
R1
C2
0.033Ω
47µF
16V
×2
1
6
I
/RUN PGATE
LTC1772
M1
TH
C5
R4
10k
150pF
2
3
5
4
R6
100Ω
CERAMIC
GND
V
IN
–
C3
220pF
V
SENSE
FB
D1
V
OUT
T1
5V
R5
22Ω
C2
•
0.5A
+
100µF
10V
×2
C4
10µH
10µH
R2
52.3k
100pF
•
CERAMIC
R3
10k
C1: AVXTPSE476M016R0047 M1: Si9803
C2: AVXTPSE107M010R0100 R1: DALE 0.25W
D1: IR10BQ015
T1: COILTRONICS CTX10-4
1772 TA04
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1147 Series High Efficiency Step-Down Switching Regulator Controllers
LT1375/LT1376 1.5A, 500kHz Step-Down Switching Regulators
100% Duty Cycle, 3.5V ≤ V ≤ 16V
IN
High Frequency, Small Inductor, High Efficiency
LTC1622
Low Input Voltage Current Mode Step-Down DC/DC Controller
V
2V to 10V, I
Up to 4.5A, Synchronizable to
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IN
750kHz Optional Burst Mode Operation, 8-Lead MSOP
LTC1624
LTC1625
LTC1627
LTC1649
High Efficiency SO-8 N-Channel Switching Regulator Controller
No R
TM Synchronous Step-Down Regulator
N-Channel Drive, 3.5V ≤ V ≤ 36V
IN
97% Efficiency, No Sense Resistor
SENSE
Low Voltage, Monolithic Synchronous Step-Down Regulator
3.3V Input Synchronous Controller
Low Supply Voltage Range: 2.65V to 8V, I
= 0.5A
OUT
No Need for 5V Supply, Uses Standard Logic Gate
MOSFETs; I up to 15A
OUT
LTC1702
LTC1735
LTC1771
550kHz, 2 Phase, Dual Synchronous Controller
Two Channels; Minimum C and C , I
up to 15A
IN
OUT OUT
Single, High Efficiency, Low Noise Synchronous Switching Controller
Ultra-Low Supply Current Step-Down DC/DC Controller
High Efficiency 5V to 3.3V Conversion at up to 15A
10µA Supply Current, 93% Efficiency,
1.23V ≤ V
≤ 18V; 2.8V ≤ V ≤ 20V
OUT
IN
LTC1872
SOT-23 Step-Up Controller
2.5V ≤ V ≤ 9.8V; 550kHz; 90% Efficiency
IN
No R
is a trademark of Linear Technology Corporation.
SENSE
1772fb
LT/LT 0605 500 REV B • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
12
●
●
© LINEAR TECHNOLOGY CORPORATION 1999
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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