AN-37_技术文档

®

LinkSwitch-TN

Design Guide

Application Note AN-37

• Universal input – the same power supply/product can be

used worldwide

• High power density – smaller size, no µF’s of X class

capacitance needed

• High efficiency – Full load efficiencies >75% typical for

12 V output

• Excellent line and load regulation

• High efficiency at light load – ON/OFF control maintains

high efficiency even at light load

• Extremely energy efficient – input power <100 mW at no

load

• Entirely manufacturable in SMD

• More robust to drop test mechanical shock

• Fully fault protected (overload, short circuit and thermal

faults)

Introduction

LinkSwitch-TNcombinesahighvoltagepowerMOSFETswitch

with an ON/OFF controller in one device. It is completely self-

poweredfromtheDRAINpin,hasajitteredswitchingfrequency

for low EMI and is fully fault protected. Auto-restart limits

device and circuit dissipation during overload and output short

circuit while over temperature protection disables the internal

MOSFET during thermal faults. The high thermal shutdown

thresholdisidealforapplicationswheretheambienttemperature

is high while the large hysteresis protects the PCB and

surrounding components from high average temperatures.

LinkSwitch-TN is designed for any application where a non-

isolatedsupplyisrequiredsuchasappliances(coffeemachines,

rice cookers, dishwashers, microwave ovens etc.), nightlights,

emergency exit signs and LED drivers. LinkSwitch-TN can be

configured in all common topologies to give a line or neutral

referencedoutputandaninvertedornon-invertedoutputvoltage

- ideal for applications using triacs for AC load control. Using

a switching power supply rather than a passive dropper

(capacitive or resistive) gives a number of advantages, some of

which are listed below.

• Scalable – LinkSwitch-TN family allows the same basic

design to be used from <50 mA to 360 mA

Scope

This application note is for engineers designing a non-isolated

power supply using the LinkSwitch-TN family of devices. This

D_FB

R_FB

C_BP

R_BIAS

C_FB

R_FD_IN2

L_IN

FB

BP

S

+

D

L

LinkSwitch-TN

C_IN2

AC

Input

V_O

D_FW

C_O

R_PL

C_IN1

D_IN2

PI-3764-121003

1 (a)

D_FB

R_FB

C_BP

R_BIAS

C_FB

R_FD_IN1

L_IN

FB

BP

S

D

D_FW

LinkSwitch-TN

C_IN2

AC

Input

L

V_O

C_O

R_PL

C_IN1

+

D_IN2

PI-3765-121003

1 (b)

Figure 1 (a). Basic Configuration using LinkSwitch-TN in a Buck Converter. Figure 1 (b) Basic Configuration using LinkSwitch-TN in a

Buck-Boost Converter.

January 2004

AN-37

document describes the design procedure for buck and buck-

boost converters using the LinkSwitch-TN family of integrated

off-line switchers. The objective of this document is to provide

power supply engineers with guidelines in order to enable them

to quickly build efficient and low cost buck or buck-boost

converter based power supplies using low cost off-the-shelf

inductors. Complete design equations are provided for the

selection of the converter’s key components. Since the power

MOSFET and controller are integrated into a single IC the

design process is greatly simplified, the circuit configuration

has few parts and no transformer is required. Therefore a quick

startsectionisprovidedthatallowsoff-the-shelfcomponentsto

be selected for common output voltages and currents.

Quick Start

Readers wanting to start immediately can use the following

information to quickly select the components for a new design,

using Figure 1 and Tables 1 and 2 as references.

1) For AC input designs select the input stage (Table 9).

2) Select the topology (Tables 1 and 2).

- If better than ±10% output regulation is required,

then use opto coupler feedback with suitable reference.

3) Select the LinkSwitch-TN device, L, R_FBor V_Z, R_BIAS, C_FB,

R_Zand the reverse recovery time for D_FW

(Table 3: Buck, table 4:Buck-Boost).

4) Select freewheeling diode to meet t_rrdetermined in step 3

(Table 5).

In addition to this application note a design spreadsheet is

available within the PIXls tool in the PI Expert design software

suite. The reader may also find the LinkSwitch-TN DAK

engineering prototype board useful as an example of a working

supply. Further details of support tools and updates to this

document can be found at www.powerint.com.

5) For direct feedback designs, if the minimum load < 3 mA

then calculate R_PL= V_O/ 3 mA.

•

6) Select C_Oas 100 µF, 1.25 V_O, low ESR type.

7) Construct prototype and verify design.

TOPOLOGY

BASIC CIRCUIT SCHEMATIC

KEY FEATURES

High-Side

Buck –

Direct

1) Output referenced to input

2) Positive output (V_O) with respect to -V_IN

3) Step down – V_O< V_IN

FB

BP

S

Feedback

4) Low cost direct feedback (±10% typ.)

D

+

LinkSwitch-TN

V_IN

V_O

PI-3751-121003

High-Side

Buck Boost –

Direct

1) Output referenced to input

2) Negative output (V_O) with respect to -V_IN

3) Step down – V_O> V_INor V_O< V_IN

4) Low cost direct feedback (± 10% typ.)

5) Fail-safe – output is not subjected to input

voltage if the internal MOSFET fails

6) Ideal for driving LEDs – better accuracy and

temperature stability than low-side Buck

constant current LED driver

Feedback

FB

BP

S

D

+

LinkSwitch-TN

V_IN

V_O

+

PI-3794-121503

Notes

1. Low cost, directly sensed feedback typically achieves overall regulation tolerance of ± 10%.

2. To ensure output regulation a pre-load may be required to maintain a minimum load current of 3 mA (Buck and Buck-Boost only).

3. Boost topology (step up) also possible but not shown.

Table 1. LinkSwitch-TN Circuit Configurations Using Directly Sensed Feedback.

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TOPOLOGY

BASIC CIRCUIT SCHEMATIC

KEY FEATURES

High-Side

Buck –

Optocoupler

Feedback

1) Output referenced to input

2) Positive output (V_O) with respect to -V_IN

3) Step down – V_O< V_IN

FB

BP

S

D

+

R_Z

4) Optocoupler feedback

LinkSwitch-TN

- Accuracy only limited by reference choice

- Low cost non-safety rated optocoupler

- No pre-load required

V_IN

V_O

V_Z

5) Minimum no-load consumption

PI-3796-121903

Low-Side

Buck –

Optocoupler

Feedback

1) Output referenced to input

2) Negative output (V_O) with respect to +V_IN

3) Step down – V_O< V_IN

+

R_Z

LinkSwitch-TN

V_IN

V_O

4) Optocoupler feedback

V_Z

- Accuracy only limited by reference choice

- Low cost non-safety rated optocoupler

- No pre-load required

BP

FB

D

S

PI-3797-121903

Low-Side

1) Output referenced to input

+

Buck Boost –

Optocoupler

Feedback

V_Z

2) Positive output (V_O) with respect to +V_IN

3) Step up/down – V_O> V_INor V_O< V_IN

4) Optocoupler feedback

LinkSwitch-TN

V_IN

V_O

R_Z

- Accuracy only limited by reference choice

- Low cost non-safety rated optocoupler

- No pre-load required

BP

FB

+

D

S

PI-3798-121903

5) Fail-safe – output is not subjected to input

voltage if the internal MOSFET fails

6) Minimum no-load consumption

Notes

1. Performance of opto feedback only limited by accuracy of reference (Zener or IC).

2. Optocoupler does not need to be safety approved.

3. Reference bias current provides minimum load. The value of R_Zis determined by Zener test current or reference IC bias current.

Typically 470 Ω to 2 kΩ, 1/8 W, 5%

4. Boost topology (step-up) is also possible but not shown.

5. Optocoupler feedback provides lowest no-load consumption.

Table 2. LinkSwitch-TN Circuit Configurations Using Optocoupler Feedback.

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AN-37

INDUCTOR

TOKIN

V_OUTI_OUT(MAX)

LNK30X MODE DIODE t_rr

R_FB

V_Z

µH I_RMS(mA)

COILCRAFT

MDCM

CCM

≤120

160

175

225

280

360

680 220

680 230

680 320

680 340

680 440

680 430

SBC2-681-211

SBC3-681-211

SBC4-681-211

RFB0807-681

RFB0810-681

≤75 ns

≤35 ns

≤75 ns

≤35 ns

≤75 ns

≤35 ns

LNK304

LNK305

LNK306

MDCM

CCM

3.84 kΩ 3.9 V

5

MDCM

CCM

MDCM

CCM

≤85

120

160

175

225

280

360

680 180

1000 230

1500 320

680 340

1000 440

680 430

1500 400

SBC2-681-211

SBC3-102-281

SBC3-152-251

SBC3-681-361

SBC4-102-291

SBC4-681-431

SBC6-152-451

RFB0807-681

RFB0807-102

RFB0810-152

RFB0810-681

RFB0810-102

RFB0810-681

RFB1010-152

≤75 ns

≤35 ns

≤75 ns

≤35 ns

≤75 ns

≤35 ns

LNK304

MDCM

CCM

11.86 kΩ 11 V

12

LNK305

LNK306

MDCM

CCM

MDCM

CCM

≤70

120

160

175

225

280

360

680 160

1200 210

1800 210

820 310

1200 310

820 390

1500 390

SBC2-681-211

RFB0807-681

RFB0807-122

RFB0810-182

RFB0810-821

RFB1010-122

RFB1010-821

RFB1010-152

≤75 ns

≤35 ns

LNK304

LNK305

-

MDCM

15

24

≤75 ns 15.29 kΩ 13 V

CCM

MDCM

CCM

≤35 ns

≤75 ns

≤35 ns

LNK306

-

SBC6-152-451

MDCM

CCM

MDCM

CCM

≤50

120

160

175

225

280

360

680 130

1500 190

2200 180

1200 280

1500 280

1200 350

2200 360

SBC2-681-211

SBC4-152-221

SBC4-222-211

-

SBC6-152-451

-

RFB0807-681

RFB0810-152

RFB0810-222

RFB0810-122

RFB1010-152

RFB1010-122

-

≤75 ns

≤35 ns

LNK304

LNK305

25.6 kΩ 22 V

≤75 ns

≤35 ns

≤75 ns

≤35 ns

LNK306

MDCM

CCM

SBC6-222-351

Other Standard Components

R_BIAS: 2 kΩ, 1%, 1/8 W

C_BP: 0.1 µF, 50 V Ceramic

•

C_FB: 10 µF, 1.25 V_O

D_FB: 1N4005GP

R_Z:

470 Ω to 2 kΩ, 1/8 W, 5%

Table 3. Components Quick Select for Buck Converters.

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INDUCTOR

TOKIN

V_OUTI_OUT(MAX)

LNK30X MODE DIODE t_rr

R_FB

V_Z

µH I_RMS(mA)

COILCRAFT

MDCM

≤75 ns

≤120

160

175

225

280

360

680 220

680 230

680 340

680 320

680 440

680 430

SBC2-681-211

SBC3-681-361

SBC4-681-431

RFB0807-681

RFB0810-681

LNK304

CCM

≤35 ns

MDCM

CCM

≤75 ns

≤35 ns

≤75 ns

≤35 ns

LNK305

LNK306

3.84 kΩ 3.9 V

5

MDCM

CCM

MDCM

CCM

MDCM

CCM

≤70

120

160

175

225

280

360

680 180

1200 220

1800 210

820 320

1200 310

820 410

1800 410

SBC2-681-211

RFB0807-681

RFB1010-122

RFB0807-182

RFB0807-821

RFB0810-122

RFB0810-821

RFB1010-182

≤75 ns

≤35 ns

≤75 ns

≤35 ns

≤75 ns

≤35 ns

LNK304

-

11.86 kΩ 11 V

12

15

24

LNK305

LNK306

MDCM

CCM

MDCM

≤50

120

160

175

225

280

360

680 180

1500 220

2200 220

1000 320

1500 320

1200 400

2200 410

SBC2-681-211

SBC3-152-251

SBC4-222-211

SBC4-102-291

SBC4-152-251

-

RFB0807-681

RFB0807-152

RFB0810-222

RFB0810-102

RFB0810-152

RFB0810-122

RFB1010-222

≤75 ns

≤35 ns

≤75 ns

≤35 ns

≤75 ns

≤35 ns

LNK304

LNK305

CCM

MDCM

15.29 kΩ 13 V

CCM

LNK306 MDCM

CCM

SBC6-222-351

≤35

120

160

175

225

280

360

680 180

2200 210

3300 210

1800 300

2200 290

1800 370

3300 410

SBC2-681-211

SBC3-222-191

SBC4-332-161

RFB0807-681

RFB0810-222

RFB0810-332

RFB0810-182

RFB1010-222

RFB1010-182

-

MDCM

CCM

MDCM

CCM

MDCM

CCM

≤75 ns

≤35 ns

≤75 ns

≤35 ns

≤75 ns

≤35 ns

LNK304

25.6 kΩ 22 V

-

LNK305

LNK306

SBC4-222-211

-

Other Standard Components

R_BIAS: 2 kΩ, 1%, 1/8 W

C_BP: 0.1 µF, 50 V Ceramic

•

C_FB: 10 µF, 1.25 V_O

D_FB: 1N4005GP

R_Z:

470 Ω to 2 kΩ, 1/8 W, 5%

Table 4. Components Quick Select for Buck-Boost Converters.

V

I

t

rr

RRM

F

PART NO.

MANUFACTURER

PACKAGE

(V)

600

(A)

1

(ns)

50

75

30

35

20

75

MUR160

UF4005

BYV26C

FE1A

Leaded

SMD

Vishay

1

Vishay/Philips

Vishay

1

STTA10 6

STTA10 6U

US1J

1

ST Microelectronics

Vishay

1

SMD

Table 5. List of Ultra-Fast Diodes Suitable for use as the Freewheeling Diode.

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AN-37

To regulate the output, an ON/OFF control scheme is used as

illustrated in Table 6. As the decision to switch is made on a

cycle by cycle basis, the resultant power supply has extremely

good transient response and removes the need for control loop

compensationcomponents. Ifnofeedbackisreceivedfor50ms

then the supply enters auto restart.

LinkSwitch-TN Circuit Design

LinkSwitch-TN Operation

The basic circuit configuration for a Buck converter using

LinkSwitch-TN is shown in Figure 1a.

= MOSFET

Enabled

Reference

Schematic

and Key

FB

BP

S

D

+

LinkSwitch-TN

V_IN

V_O

= MOSFET

Disabled -

Cycle Skipped

PI-3784-121603

I_D

At beginning of each cycle the FEEDBACK

(FB) pin is sampled.

• If I_FB< 49 µA then next cycle occurs

• If I_FB> 49 µA then next switching cycle

is skipped

Is I_FB

>49 µA?

No

No Yes

No

Yes Yes No

Normal

Operation

High load – few cycles skipped

Low load – many cycles skipped

PI-3767-121903

I_FB< 49 µA, > 50 ms

= Auto Restart

If no feedback (I_FB< 49 µA) for > 50 ms

then output switching is disabled for

approximately 800 ms.

Auto Restart

50 ms

800 ms

Auto Restart = 50 ms ON / 800 ms OFF

PI-3768-121603

Table 6. LinkSwitch-TN Operation.

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AN-37

To allow direct sensing of the output voltage, without the need

forareference(ZenerdiodeorreferenceIC), theFBpinvoltage

is tightly toleranced over the entire operating temperature

range. For example, this allows a 12 V design with an overall

output tolerance of ±10%. For higher performance, an opto-

coupler can be used with a reference as shown in table 2. Since

the optocoupler just provides level shifting, it does not need to

be safety rated or approved. The use of an optocoupler also

allows flexibility in the location of the device, for example it

allows a buck converter configuration with the LinkSwitch-TN

in the low side return rail, reducing EMI as the SOURCE pins

and connected components are no longer part of the switching

node.

freewheeling diode, and the average current through the output

inductor are slightly lower in the Buck topology as compared to

the Buck-Boost topology.

Selecting the Operating Mode – MDCM and CCM

Operation

At the start of a design, select between mostly discontinuous

conduction mode (MDCM) and continuous conduction mode

(CCM) as this decides the selection of the LinkSwitch-TN

device, freewheeling diode and inductor. For maximum output

currentselectCCM,forallothercasesMDCMisrecommended.

Overall, select the operating mode and components to give the

lowest overall solution cost. Table 7 summarizes the trade-offs

between the two operating modes.

Selecting the Topology

IfpossibleusetheBucktopology, theBucktopologymaximizes

the available output power from a given LinkSwitch-TN and

inductor value. Also, the voltage stress on the power switch and

Additional differences between CCM and MDCM include

better transient response for DCM and lower output ripple (for

same capacitor ESR) for CCM. However these differences, at

COMPARISON OF CCM AND MDCM OPERATING MODES

OPERATING MODE

MDCM

CCM

I_L

I_O

Operating

Description

t

t_ON

t_OFF

t_IDLE

t_ON

t_OFF

PI-3769-121803

PI-3770-121503

Inductor current falls to zero during t_OFF

,

Current flows continuously in the inductor for

Borderline between MDCM and CCM when the entire duration of a switching cycle.

t_IDLE= 0.

Lower Cost

Lower value, smaller size.

Higher Cost

Higher value, larger size.

Inductor

Lower Cost

75 ns ultra-fast reverse recovery type

(≤35 ns for ambient >70 °C).

Higher Cost

35 ns ultra-fast recovery type required.

Freewheeling

Diode

Potentially Higher Cost

Potentially Lowest Cost

May require larger device to deliver required May allow smaller device to deliver required

LinkSwitch-TN

output current–depends on required output

current.

output current–depends on required output

current.

Higher Efficiency

Lower switching losses.

Lower Efficiency

Higher switching losses.

Efficiency

Overall

Typically Lower Cost

Typically Higher Cost

Table 7. Comparison of Mostly Discontinuous Conduction (MDCM) and Continuous Conduction (CCM) Modes of Operation.

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the low output currents of LinkSwitch-TN applications, are

Output Power, P_O: in Watts.

normally not significant.

Power supply efficiency, η: 0.7 for a 12 V output, 0.55 for a

5 V output if no better reference data available.

The conduction mode CCM or MDCM of a Buck or Buck-

Boost converter primarily depends on input voltage, output

voltage, output current and device current limit. The input

voltage, output voltage and output current are fixed design

parameters therefore the LinkSwitch-TN (current limit) is the

only design parameter that sets the conduction mode.

Total Capacitance C_IN(TOTAL)

µF/P_OUT(C_IN1+ C_IN2)

AC Input

Voltage (VAC)

Half Wave

Full Wave

Rectification

100/115

230

6-8

1-2

6-8

3-4

1

The phrase “mostly discontinuous” is used as with on-off

control, since a few switching cycles may exhibit continuous

inductor current, the majority of the switching cycles will be in

the discontinuous conduction mode. A design can be made

fully discontinuous but that will limit the available output

current, making the design less cost effective.

Universal

3-4

Table 10. Suggested Total Input Capacitance Values for Different

Input Voltage Ranges.

Step 2. Determine AC Input Stage

Step-by-Step Design Procedure

The input stage comprises fusible resistor(s) input rectification

diodes and line filter network. The fusible resistor should be

chosen as flame proof and depending on the differential line

input surge requirements, a wire wound type may be required.

The fusible resistor(s) provides fuse safety, inrush current

limiting and differential mode noise attenuation.

Step 1. Determine System Requirements VAC_MIN

VAC_MAX, P_O, V_O, f_L, η

,

Determine the input voltage range from Table 8.

Input (VAC)

100/115

230

VAC_MIN

85

VAC_MAX

132

For designs ≤1 W it is lower cost to use half-wave rectification,

>1 W full wave rectification (smaller input capacitors). The

EMI performance of half wave rectified designs is improved by

adding a second diode in the lower return rail. This provides

EMI gating (EMI currents only flow when the diode is

conducting)andalsodoublesdifferentialsurgewithstandasthe

surge voltage is shared across two diodes. Table 9 shows the

recommendedinputstagebasedonoutputpowerforauniversal

input design while Table 10 shows how to adjust the input

capacitance for other input voltage ranges.

195

85

265

Universal

265

Table 8. Standard Worldwide Input Line Voltage Ranges.

Line Frequency, f_L: 50 or 60 Hz, for half-wave rectification

use f_L/2.

Output Voltage, V_O: in Volts.

P_OUT

≤ 0.25 W

0.25-1 W

> 1 W

D_IN1-4

L_IN

**

+

L_IN

**

+

R_F1

D_IN1

R_F1

D_IN1

R_F1

D_IN1

R_F2

R_F1

**

C_IN

AC

IN

AC

IN

AC

IN

**

C_IN1

C_IN2

C_IN1

C_IN2

AC IN

C_IN1

C_IN2

*

D_IN2

R_F2

PI-3772-121603

PI-3771-121603

PI-3773-121603

PI-3774-121603

85-265 VAC

Input Stage

R_F1, R_F2: 100-470 Ω,

0.5 W, Fusible

R_F1: 8.2 Ω, 1 W Fusible

R_F2: 100 Ω, 0.5 W,

Flame proof

R_F1: 8.2 Ω, 1 W Fusible

L_IN: 470 µH-2.2 mH,

0.05 A-0.3 A

R_F1: 8.2 Ω, 1 W Fusible

L_IN: 470 µH-2.2 mH,

0.05 A-0.3 A

C_IN: ≥2.2 µF, 400 V

D_IN1, D_IN2: 1N4007, 1 A,

1000 V

C_IN1, C_IN2: ≥3.3 µF,

400 V each

C_IN1, C_IN2: ≥4 µF/W_OUT

400 V each

,

C_IN1, C_IN2: ≥2 µF/W_OUT,

400 V each

D_IN1, D_IN2: 1N4007, 1 A,

1000 V

D_IN1, D_IN2: 1N4007, 1 A,

1000 V

D_IN1, D_IN2: 1N4005, 1 A,

600 V

*Optional for improved EMI and line surge performance. Remove for designs requiring no impedance in return rail.

**Increase value to meet required differential line surge performance.

Comments

Table 9. Recommended AC Input Stages For Universal Input.

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Step 3. Determine Minimum and Maximum DC Input

Voltages V_MINand V_MAXBased on AC Input Voltage

Step 5. Select the Output Inductor

Tables3and4provideinductorvaluesandRMScurrentratings

for common output voltages and currents based on the

calculations in the design spreadsheet. Select the next nearest

higher voltage and/or current above the required output

specification. Alternatively the PIXls spreadsheet tool in the

PI Expert software design suite or Appendix A can be used to

calculate the exact inductor value (Eq. A7) and RMS current

rating (Eq. A20).

Calculate V_MAXas

(1)

V_MAX

= 2 ⋅V_ACMAX

Assuming that the value of input fusible resistor is small, the

voltage drop across it can be ignored.

Assume bridge diode conduction time of t_c= 3 ms if no other

data available.

It is recommended that the value of inductor chosen should be

closer to L_TYPrather than 1.5 L_TYPdue to lower DC resistance

•

Derive minimum input voltage V_MIN

and higher RMS rating. The lower limit of 680 µH limits the

maximum di/dt to prevent very high peak current values.

Tables 3 and 4 provide reference part numbers for standard

inductors from two suppliers.





1

2 ⋅ P

− t_C

O





2 ⋅ f





L

2

(2)

V_MIN

=

2 ⋅V

ACMIN

(

)

680 µH < L_TYP< L < 1.5⋅ L_TYP

η ⋅C_{IN (TOTAL)}

(5)

If V_MINis ≤70 V then increase value of C_IN(TOTAL)

.

For LinkSwitch-TN designs the mode of operation is not

dependent on the inductor value. The mode of operation is a

function of load current and current limit of the chosen device,

theinductorvaluemerelysetstheaverageswitchingfrequency.

Step 4. Select LinkSwitch-TN Device Based on

Output Current and Current Limit

Decide on operating mode - refer to Table 7.

Figure 2 shows a typical standard inductor manufacturer’s data

sheet. The value of off-the-shelf “drum core / dog bone / I core”

inductors will drop up to 20% in value as the current increases.

TheconstantK_{L_TOL}inequation(A7)andthedesignspreadsheet

adjusts for both this drop and the initial inductance value

tolerance.

For MDCM operation, the output current (I_O) should be less

than or equal to half the value of the minimum current limit of

the chosen device from the data sheet.

(3)

I_{LIMIT _ MIN}> 2 ⋅ I_O

Forexampleifa680µH, 360mAinductorisrequired, referring

to Figure 2, the tolerance is 10% and an estimated 9.5% for the

reduction in inductance at the operating current (approximately

For CCM operation, the device should be chosen such that the

output current I_O, is more than 50%, but less than 80% of the

•

minimum current limit I_{LIMIT_MIN}

.

[0.36/0.38] 10). ThereforethevalueofK_{L_TOL}=1.195(19.5%).

If no data is available assume a K_{L_TOL}of 1.15 (15%).

0.5⋅ I_{LIMIT _ MIN}< I_O< 0.8⋅ I_{LIMIT _ MIN}

(4)

Not all the energy stored in the inductor is delivered to the load,

due to losses in the inductor itself. To compensate for this a loss

Please see data sheet for LinkSwitch-TN current limit values.

Inductance and Current Rating

Current Rating

for 40 °C Rise

Current Rating

for Value -10%

Tolerance

for 20 °C Rise

SBC3 Series (SBC3-

Model

-

)

Inductance

Rdc

(W)

Rated Current

Current (Reference Value)

(A)

L(mH/ at 10 kHz

max.

∆T = 20 °C

∆T = 40 °C

L change rate -10%

681-361

102-281

152-251

222-191

332-151

680±10%

1000±10%

1500±10%

2200±10%

3300±10%

1.62

2.37

3.64

5.62

7.66

0.36

0.28

0.25

0.19

0.15

0.50

0.39

0.35

0.26

0.21

0.38

0.31

0.26

0.21

0.17

PI-3783-121003

Figure 2. Example of Standard Inductor Data Sheet.

A

1/04

9

AN-37

factor K_LOSSis used. This has a recommended value of between

50%and66%ofthetotalsupplylossesasgivenbyequation(5).

For example, a design with an overall efficiency (η) of 0.75

would have a K_LOSSvalue of between 0.875 and 0.833.

Let the value of R_BIAS= 2 kΩ; this biases the feedback network

at a current of ∼0.8 mA. Hence the value of R_FBis given by

V − V ⋅ R

V −1.65 V ⋅2 kΩ

V_O− V_FB

(

)

(

=

)

O

FB

BIAS

O

R_FB

=

V_FB

V_FB+ I_FB⋅ R_BIAS

1.748 V

(

)

+ I_FB

 1− η 

 2 1− η 

(

)

(

)

R_BIAS

(6)

K_LOSS= 1−

to 1−



(10)





2

3









Step 9. Select the Feedback Diode and Capacitor

Step 6. Select Freewheeling Diode

For the feedback capacitor, use a 10 µF general purpose

•

For MDCM operation at t_AMB≤70 °C, select an ultra-fast diode

with t_rr≤75 ns. At t_AMB>70 °C, t_rr≤ 35 ns.

electrolytic capacitor with a voltage rating ≥1.25 V_O.

For the feedback diode, use a glass passivated 1N4005GP or

•

For CCM operation, select an ultra-fast diode with t_rr≤35 ns.

1N4937GP device with a voltage rating of ≥1.25 V_MAX

Step 10. Select Bypass Capacitor

Use 0.1 µF, 50 V ceramic capacitor.

.

Allowing 25% design margin for the freewheeling diode,

(7)

V_PIV> 1.25⋅V_MAX

The diode must be able to conduct the full load current. Thus

Step 11. Select Pre-load Resistor

(8)

For direct feedback designs if the minimum load <3 mA then

calculate R_PL= V_O/ 3 mA.

I_F> 1.25⋅ I_O

Table 5 lists common freewheeling diode choices.

Other information

Step 7. Select Output Capacitor

Startup Into Non-Resistive Loads

The output capacitor should be chosen based on the output

voltage ripple requirement. Typically the output voltage ripple

is dominated by the capacitor ESR and can be estimated as:

If the total system capacitance is >100 µF or the output voltage

is >12 V, then the output may fail to reach regulation during

start-up. Thismayalsobetruewhentheloadisnotresistive, for

example the output is supplying a motor or fan.

V_RIPPLE

(9)

ESR_MAX

=

I_LIMIT

To increase the startup time, a soft-start capacitor can be added

across the feedback resistor, as shown in Figure 3. The value of

this soft-start capacitor is typically in the range of 0.47 µF to

where V_RIPPLEis the maximum output ripple specification and

I_LIMITis the LinkSwitch-TN current limit. The capacitor ESR

value should be specified approximately at the switching

frequency of 66 kHz.

•

47 µF with a voltage rating of 1.25 V_O. Figure 4 shows the

effect of C_SSused on a 12 V, 150 mA design driving a motor

load.

Capacitor values above 100 µF are not recommended as they

can prevent the output voltage from reaching regulation during

the 50 ms period prior to auto-restart. If more capacitance is

required, then a soft-start capacitor should be added (see Other

Information section).

C_SS

R_FB

Step 8. Select the Feedback Resistors

FB

BP

S

The values of R_FBand R_BIASare selected such that at the

regulated output voltage, the voltage on the FEEDBACK pin

(V_FB) is 1.65 V. This voltage is specified for a FEEDBACK pin

current (I_FB) of 49 µA.

D

+

LinkSwitch-TN

V_IN

V_O

PI-3775-121003

Figure 3. Example Schematic Showing Placement of Soft-Start

Capacitor.

A

1/04

10

AN-37

14

12

10

8

No soft-start capacitor. Output

never reaches regulation (in

auto-restart).

FB

BP

S

+7 V

D

+

LinkSwitch-TN

6V8

V_IN

RTN

-5 V

6

4

5V1

PI-3776-121003

2

0

Figure 5. Example Circuit – Generating Dual Output Voltages.

-2

2.5

Time (s)

5

0

Generating Negative and Positive Outputs

Inapplianceapplicationsthereisoftenarequirementtogenerate

both an AC line referenced positive and negative output. This

can be accomplished using the circuit in Figure 5. The two

zener diodes have a voltage rating close to the required output

voltage for each rail and ensure that regulation is maintained

when one rail is lightly and the other heavily loaded. The

LinkSwitch-TN circuit is designed as if it were a single output

voltage with an output current equal to the sum of both outputs.

The magnitude sum of the output voltages in this example being

12 V.

14

12

10

8

6

4

Soft-start capacitor value too

small – output still fails to reach

regulation before auto-restart.

2

Constant Current Circuit Configuration (LED Driver)

0

-2

The circuit shown in Figure 5 is ideal for driving constant

current loads such as LEDs. It uses the tight tolerance and

temperature stable FEEDBACK pin of LinkSwitch-TN as the

reference to provide an accurate output current.

2.5

Time (s)

5

0

14

12

10

8

R_FB

300 Ω

Optional

See Text

Correct value of soft-start

capacitor – output reaches

VR_FBD_FB

R_BIAS

regulation before auto-restart.

2 kΩ

FB

BP

S

R_SENSE

I_O

6

4

D

+

D_FW

LinkSwitch-TN

C_SENSE

L

V_IN

C_O

2

0

PI-3795-122403

-2

Figure 6. High-Side Buck-Boost Constant Current Output

Configuration.

2.5

Time (s)

5

0

To generate a constant current output, the average output

current is converted to a voltage by resistor R_SENSEand capacitor

Figure 4. Example of Using Soft-Start Capacitor to Enable Driving

a 12 V, 0.15 A Motor Load. All Measurements were made

at 85 VAC (worst case condition).

C_SENSEand fed into the FEEDBACK pin via R_FBand R_BIAS

.

With the values of R_BIASand R_FBas shown, the value of R_SENSE

should be chosen to generate a voltage drop of 2 V at the

A

1/04

11

AN-37

required output current. Capacitor C_SENSEfilters the voltage

across R_SENSE, which is modulated by inductor ripple current.

The value of C_SENSEshould be large enough to minimize the

ripple voltage, especially in MDCM designs. A value of C_SENSE

is selected such that the time constant (t) of R_SENSEand C_SENSEis

greater than 20 times that of the switching period (15 µs). The

•

Recommended Layout Considerations

Traces carrying high currents should be as short in length and

thick in width, as possible. These are the traces which connect

the input capacitor, LinkSwitch-TN, inductor, freewheeling

diode and the output capacitor.

peak voltage seen by C_SENSEis equal to R_SENSEI_LIMIT(MAX)

.

Most off-the-shelf inductors are drum core inductors or dog-

bone inductors. These inductors do not have a good closed

magneticpath,andareasourceofsignificantmagneticcoupling.

They are a source of differential mode noise and for this reason,

they should be placed as far away as possible from the AC input

lines.

The output capacitor is optional; however with no output

capacitor the load will see the full peak current (I_LIMIT) of the

selected LinkSwitch-TN. Increase the value of C_O(typically in

the range of 100 nF to 10 uF) to reduce the peak current to an

acceptable level for the load.

If the load is disconnected, feedback is lost and the large output

voltage which results may cause circuit failure. To prevent this,

a second voltage control loop, D_FBand VR_FB, can be added as

shown if Figure 6. This also requires that C_Ois fitted. The

voltage of the Zener is selected as the next standard value above

the maximum voltage across the LED string when it is in

constant current operation.

Appendix A

Calculations for Inductor Value for Buck and Buck-

Boost Topologies

There is a minimum value of inductance that is required to

deliver the specified output power, regardless of line voltage

and operating mode.

The same design equations / design spreadsheet can be used as

for a standard buck-boost design, with the following additional

considerations.

V_IN-V_O

V_L

•

1. V_O= LED V_FNumber of LEDs per string

t

•

2. I_O= LED I_FNumber of strings

3. Lower efficiency estimate due to R_SENSElosses (enter

V_O

R_SENSEinto design spreadsheet as inductor resistance)

4. Set R_BIAS= 2 kΩ and R_FB= 300 Ω

5. R_SENSE= 2/I_O

I_Limit

I_L

•

6. C_SENSE= 20 (15 µs/R_SENSE

)

7. Select C_Obased on acceptable output ripple current

through the load

I_O

t

8. If the load can be disconnected or for additional fault

protection, add voltage feedback components D_FBand

VR_FB, in addition to C_O.

t_ON

t_IDLE

t_OFF

PI-3778-121803

Figure 7. Inductor Voltage and Inductor Current of a Buck

Converter in DCM.

Thermal Environment

As a general case, Figure 7 shows the inductor current in

discontinuous conduction mode (DCM). The following

expressions are valid for both CCM as well as DCM operation.

There are three unique intervals in DCM as can be seen from

Figure 7. Interval t_ONis when the LinkSwitch-TN is ON and the

freewheeling diode is OFF. Current ramps up in the inductor

fromaninitialvalueofzero.Thepeakcurrentisthecurrentlimit

I_LIMITof the device. Interval t_OFFis when the LinkSwitch-TN is

OFF and the freewheeling diode is ON. Current ramps down to

zero during this interval. Interval t_IDLEis when both the

LinkSwitch-TNandfreewheelingdiodeareOFF,andtheinductor

current is zero.

To ensure good thermal performance, the SOURCE pin

temperature should be maintained below 100 °C, by providing

adequate heatsinking.

For applications with high ambient temperature (>50 °C), it is

recommendedtobuildandtestthepowersupplyatthemaximum

operatingambienttemperature,andensurethatthereisadequate

thermal margin. The figures for maximum output current

providedinthedatasheet correspondtoanambienttemperature

of 50 °C, and may need to be thermally derated. Also, it is

recommended to use ultra fast (≤35 ns) low reverse recovery

diodes at higher operating temperatures (>70 °C).

A

1/04

12

AN-37

In CCM this idle state does not exist and thus t_IDLE= 0.

1

2

I_RIPPLE⋅ L_MIN













⋅ I

+ I_INITIAL

(

)

LIMIT _ MIN

V_MIN− V_DS− V_O

_RIPPLE⋅ L_MIN

1

Neglecting the forward voltage drop of the freewheeling diode,

we can express the current swing at the end of interval t_ONin a

buck converter as

I_O=

1

2

I

T_{SW _ MAX}

+

⋅ I

(

+ I_INITIAL

LIMIT _ MIN

)

V_O

(A5)

V_MIN− V_DS− V_O

∆I t

= I

=

⋅t_ON

(

)

ON

RIPPLE

2 ⋅ V ⋅ I ⋅ V

− V_DS− V_O

(

_O) (

)

L_MIN

O

MIN

L_MIN

=

2

I

− I_INITIAL⋅ FS

⋅ V

− V_DS

(

)

(

)

LIMIT _ MIN

MIN

I_RIPPLE= 2 ⋅ I

− I_Ot_IDLE= 0 for CCM

(

)

(

)

LIMIT _ MIN

(A6)

I_RIPPLE= I_{LIMIT _ MIN}

,

t_IDLE> 0 for DCM

(

(A1)

Thishoweverdoesnotaccountforthelosseswithintheinductor

(resistance of winding and core losses) and the freewheeling

diode,whichwilllimitthemaximumpowerdeliveringcapability

and thus reduce the maximum output current. The minimum

inductance must compensate for these losses in order to deliver

specified full load power. An estimate of these losses can be

made by estimating the total losses in the power supply, and

then allocating part of these losses to the inductor and diode.

This is done by the loss factor K_LOSSwhich increases the size of

the inductor accordingly.

where

I_RIPPLE= Inductor Ripple Current

I_{LIMIT_MIN}= Minimum current limit

V_MIN= Minimum DC Bus Voltage

V_DS= On state Drain to Source Voltage drop

V_O= Output Voltage

L_MIN= Minimum Inductance

Similarly, we can express the current swing at the end of

interval t_OFFas

Furthermore, typical inductors for this type of application are

bobbin core or dog bone chokes. The specified current rating

refertoatemperatureriseof20°Cor40°Candtoaninductance

drop of 10%. We must incorporate an Inductance Tolerance

FactorK_{L_TOL}withintheexpressionforminimuminductance,to

accountforthismanufacturingtolerance. Thetypicalinductance

value thus can be expressed as

V_O

∆I t

(

= I

=

⋅t_OFF

)

OFF

RIPPLE

(A2)

L_MIN

The initial current through the inductor at the beginning of each

switching cycle can be expressed as

I_INITIAL= I_{LIMIT _ MIN}− I_RIPPLE

(A3)





V_O⋅ I_O

2 ⋅ K_{L _ TOL}

⋅

⋅ V

− V_DS− V_O

(

)

MIN





K



_LOSS

L_TYP

=

The average current through the inductor over one switching

cycle is equal to the output current I_O. This current can be

expressed as

2

I

− I_INITIAL⋅ FS

⋅ V

− V_DS

(

)

(

)

LIMIT _ MIN

MIN

(A7)

1

2

1

2





where

⋅ I

+ I_INITIAL⋅t

+

ON

⋅

(

)

LIMIT _ MIN

1









I_O=

T_{SW _ MAX}

K_LOSSisalossfactor,whichaccountsfortheoff-statetotallosses

of the inductor.

I

+ I_INITIAL⋅t

+ 0 ⋅t_IDLE

OFF

(

)

LIMIT _ MIN

(A4)

K_{L_TOL}is the Inductor Tolerance Factor and can be between 1.1

and 1.2. A typical value is 1.15.

where

I_O= Output Current.

T_{SW_MAX}= the switching interval corresponding to minimum

switching frequency FS_MIN

With this typical inductance we can express maximum output

power as

.

1

2

P

=

⋅ L_TYP⋅ I

− I_INITIAL

LIMIT _ MIN

⋅

(

)

O _ MAX

2

Substituting for t_ONand t_OFFfrom equations (A1) and (A2) we

have

V_MIN− V_DS

K_LOSS

(A8)

FS_MIN

⋅

V_MIN− V_DS− V_OK_{L _ TOL}

A

1/04

13

AN-37

SimilarlyforBuck-BoosttopologytheexpressionsforL_TYPand

P_{O_MAX}are

The current through the LinkSwitch-TN as a function of time is

given by





V_MIN− V_DS− V_O

V_O⋅ I_O

i_SWt = I

( )

+

⋅t ,0 < t ≤ t_ON

2 ⋅ K_{L _ TOL}

⋅

INITIAL





L

K



_LOSS

L_TYP

=

2

(A9)

I

− I_INITIAL⋅ FS

(

)

LIMIT _ MIN

MIN

i_SWt = 0 ,t < t ≤ t_ON

( )

ON

(A14)

The current through the Freewheeling diode as a function of

time is given by

1

2

P

=

⋅ L_TYP⋅(I_{LIMIT _ MIN}− I_INITIAL

)

O _ MAX

(A10)

2

i_Dt = 0, 0 < t ≤ t_ON

( )

Average Switching Frequency

V_O

i_Dt = I

( )

−

,t_ON< t ≤ t_SW

LIMIT _ MIN

(A15)

(A16)

L

SinceLinkSwitch-TNusesanon-offtypeofcontrol,thefrequency

of switching is non-uniform due to cycle skipping. We can

average this switching frequency by substituting the maximum

powerastheoutputpowerinequation(A8).Simplifying,wehave

V_O

L

i_Dt = 0, I_{LIMIT _ MIN}

( )

−

⋅t < 0

And the current through the inductor as a function of time is

given by

2 ⋅V_O⋅ I_O⋅ K_{L _ TOL}

V_MIN− V_DS− V_O

V_MIN− V_DS

FS_AVG

=

⋅

i t = i t + i

t

_L( ) _SW( ) _D( )

(A17)

2

L ⋅ I

− I_INITIALK

LOSS

(

)

LIMIT

(A11)

From the definition of RMS currents we can express the RMS

currentsthroughtheswitch, freewheelingdiodeandinductoras

follows

Similarly for Buck-Boost converter, simplifying equation (A9)

we have

t_ON

1

i_{SW _ RMS}

=

∫ i_SWt

²⋅ dt

( )

(A18)

K_{L _ TOL}

2 ⋅V_O⋅ I_O

T_AVG

0

FS_AVG

=

⋅

2

K_LOSS

L ⋅ I

− I_INITIALK

LOSS

(

)

LIMIT

(A12)

t_ON+t_OFF

1

i_{D _ RMS}

=

∫

i_Dt

²⋅ dt

( )

Calculation of RMS Currents

(A19)

(A20)

T_AVG

t_ON

The RMS current value through the inductor is mainly required

to ensure that the inductor is appropriately sized and will not

over heat. Also, RMS currents through the LinkSwitch-TN and

freewheeling diode are required to estimate losses in the power

supply.

T_AVG

1

i_{L _ RMS}

=

∫

i

t + i t

²⋅ dt

_SW( ) ( )

D

(

)

T_AVG

0

Since the switch and freewheeling diode currents fall to zero

during the turn off and turn on intervals respectively, the RMS

inductor current is simplified to

Assuming CCM operation, the initial current in the inductor in

steady state is given by

V_O

2

I_INITIAL= I_{LIMIT _ MIN}

−

⋅t_OFF

i_{L _ RMS}= i_{SW _ RMS}+ i_{D _ RMS}

(A13)

(A21)

L

For DCM operation this initial current will be zero.

A

1/04

14

AN-37

Table A1 lists the design equations for important parameters

using the Buck and Buck-Boost topologies.

PARAMETER BUCK

BUCK-BOOST

L_TYP











V_O⋅ I_O

2 ⋅ K_L⋅

⋅ V

− V_DS− V_O

(

)

MIN







K



_{L _ LOSS}



_{L _ LOSS}

L_TYP

=

L_TYP

=

2

I

− I_INITIAL⋅ FS

I

− I_INITIAL⋅ FS

⋅ V

− V_DS

(

)

(

)

(

)

LIMIT _ MIN

MIN

LIMIT _ MIN

MIN

2 ⋅V_O⋅ I_O⋅ K_L

V_MIN− V_DS− V_O

V_MIN− V_DS

2 ⋅V_O⋅ I_O

K_L

K_{L _ LOSS}

F_AVG

FS_TYP

=

⋅

FS_AVG

=

⋅

2

L ⋅ I

− I_INITIAL⋅ K

L ⋅ I

− I_INITIAL

(

)

(

)

LIMIT

L _ LOSS

LIMIT

i_SW(t)

V_MIN− V_DS

V_MIN− V_DS− V_O

i_SWt = i

( )

+

⋅t ,t ≤ t_ON

i_SWt = i

( )

+

⋅t ,t ≤ t_ON

INIT

LinkSwitch-TN

Current

L

i_SWt = 0 ,t > t

( )

i_SWt = 0 ,t > t

( )

ON

V_O

i_Dt = I

( )

−

⋅t ,t > t_ON

i_d(t)

i_Dt = I

( )

−

⋅t ,t > t_ON

LIMIT _ MIN

L

Diode

Forward

Current

V_O

L

i_Dt = 0 , I

( )

−

⋅t < 0

L

i_Dt = 0 , I

( )

−

⋅t < 0

LIMIT _ MIN

i_Dt = 0 ,t ≤ t

( )

i_Dt = 0 ,t ≤ t

( )

ON

i_L(t) Inductor

Current

i t = i t + i

t

_L( ) _SW( ) _D( )

i t = i t + i t

_L( ) _SW( ) _D( )

Max Drain

Voltage

V_MAX+ V_O

V_MAX

Table A1. Circuit Characteristics for Buck and Buck-Boost Topologies.

A

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AN-37

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and foreign patents or potentially by pending U.S. and foreign patent applications assigned to Power Integrations. A complete list of Power Integrations’ patents

may be found at www.powerint.com.

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POWER INTEGRATIONS' PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR

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properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user.

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support device or system, or to affect its safety or effectiveness.

The PI logo, TOPSwitch, TinySwitch, LinkSwitch and EcoSmart are registered trademarks of Power Integrations.

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D-80336, Muenchen, Germany

Phone: +49-895-527-3910

Fax: +49-895-527-3920

International Holdings, Inc.

5F-1, No. 316, Nei Hu Rd., Sec. 1

Nei Hu Dist.

Buford, GA 30518, USA

Phone:

Fax:

+1-678-714-6033

+1-678-714-6012

Taipei, Taiwan 114, R.O.C.

Phone:

Fax:

+886-2-2659-4570

+886-2-2659-4550

e-mail: eurosales@powerint.com

e-mail: usasales@powerint.com

Phone:

Fax:

+81-45-471-1021

+81-45-471-3717

e-mail: taiwansales@powerint.com

e-mail: japansales@powerint.com

UK (EUROPE & AFRICA

HEADQUARTERS)

Power Integrations (Europe) Ltd.

1st Floor, St. James’s House

East Street

CHINA (SHANGHAI)

Power Integrations

International Holdings, Inc.

Rm 807, Pacheer

INDIA (TECHNICAL SUPPORT)

Innovatech

261/A, Ground Floor

7th Main, 17th Cross,

Sadashivanagar

KOREA

Power Integrations

International Holdings, Inc.

8th Floor, DongSung Bldg.

17-8 Yoido-dong,

Commercial Centre

Farnham

Surrey

GU9 7TJ

United Kingdom

555 Nanjing West Road

Shanghai, 200041, China

Bangalore 560080

Youngdeungpo-gu,

Seoul, 150-874, Korea

Phone:

Fax:

+91-80-5113-8020

+91-80-5113-8023

Phone:

Fax:

+86-21-6215-5548

+86-21-6215-2468

Phone:

Fax:

+82-2-782-2840

+82-2-782-4427

e-mail: indiasales@powerint.com

Phone: +44 (0) 1252-730-140

Fax: +44 (0) 1252-727-689

e-mail: eurosales@powerint.com

e-mail: chinasales@powerint.com

e-mail: koreasales@powerint.com

APPLICATIONS HOTLINE

World Wide +1-408-414-9660

APPLICATIONS FAX

World Wide +1-408-414-9760

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型号：	AN-37
厂家：	ETC
描述：	LinkSwitch-TN Design Guide 使用LinkSwitch -TN设计指南\n
文件：	总16页 (文件大小：156K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

AN-37 [ETC]

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