AS1324_13 [AMSCO]
1.5MHz, 600mA Synchronous DC/DC Conver ter; 为1.5MHz ， 600mA同步DC / DC CONVER器
| 型号： | AS1324_13 |
| 厂家： | 
AMS(艾迈斯) |
| 描述： | 1.5MHz, 600mA Synchronous DC/DC Conver ter |
| 文件： | 总21页 (文件大小：1041K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
AS1324
1.5MHz, 600mA Synchronous DC/DC Converter
1 General Description
2 Key Features
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High Efficiency: Up to 96%
The AS1324 is a high-efficiency, constant-frequency synchronous
buck converter available in adjustable- and fixed-voltage versions.
The wide input voltage range (2.7V to 5.5V), automatic powersave
mode and minimal external component requirements make the
AS1324 perfect for any single Li-Ion battery-powered application.
Output Current: 600mA
Input Voltage Range: 2.7V to 5.5V
Constant Frequency Operation: 1.5MHz
Variable- and Fixed-Output Voltages
No Schottky Diode Required
Automatic Powersave Operation
Low Quiescent Current: 30µA
Internal Reference: 0.6V
Typical supply current with no load is 30µA and decreases to ≤1µA
in shutdown mode.
The AS1324 is available as the standard versions listed in Table 1.
Table 1. Standard Versions
Model
Output Voltage
Adjustable via External Resistors
Fixed: 1.2V
AS1324-AD
AS1324-12
AS1324-15
AS1324-18
Fixed: 1.5V
Fixed: 1.8V
Shutdown Mode Supply Current: ≤1µA
Thermal Protection
An internal synchronous switch increases efficiency and eliminates
the need for an external Schottky diode. The internally fixed
switching frequency (1.5MHz) allows for the use of small surface
mount external components.
5-pin TSOT-23 Package
Very low output voltages can be delivered with the internal 0.6V
feedback reference voltage.
3 Applications
The device is ideal for mobile communication devices, laptops and
PDAs, ultra-low-power systems, threshold detectors/discriminators,
telemetry and remote systems, medical instruments, or any other
space-limited application with low power-consumption requirements.
The AS1324 is available in a 5-pin TSOT-23 package.
Figure 1. Typical Application Diagram – High Efficiency Step
Down Converter
4.7µH
V
OUT = 1.8V, 600mA
V
IN = 2.7V to 5.5V
4
3
EN
GND
SW
1
2
3
5
4
VOUT
VIN
SW
CIN
C
OUT
10µF
10µF
AS1324-18
AS1324-18
5
1
VOUT
EN
VIN
GND
2
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Revision 1.06
1 - 21
AS1324
Datasheet - Pin Assignments
4 Pin Assignments
Figure 2. Pin Assignments (Top View)
EN
1
EN
5
VFB
1
2
3
5
VOUT
AS1324-12/
AS1324-15/
AS1324-18
AS1324
GND
SW
2
3
GND
SW
4
VIN
4
VIN
4.1 Pin Descriptions
Table 2. Pin Descriptions
Pin Number
Pin Name
Description
Enable Input. Driving this pin above 1.5V enables the device. Driving this pin below 0.3V puts the
device in shutdown mode. In shutdown mode all functions are disabled while SW goes high
impedance, drawing <1µA supply current.
1
EN
Note: This pin should not be left floating.
Ground.
2
3
GND
SW
Switch Node Connection to Inductor. This pin connects to the drains of the internal main and
synchronous power MOSFET switches.
Input Supply Voltage. This pin must be closely decoupled to GND with a ≥ 4.7µF ceramic capacitor.
Connect to any supply voltage between 2.7 to 5.5V.
4
5
VIN
Feedback Pin. This pin receives the feedback voltage from the external resistor divider across the
output. (Adjustable voltage variant only.)
V
FB
Output Voltage Feedback Pin. An internal resistor divider steps the output voltage down for
comparison to the internal reference voltage. (Fixed voltage variants only.)
V
OUT
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Revision 1.06
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AS1324
Datasheet - Absolute Maximum Ratings
5 Absolute Maximum Ratings
Stresses beyond those listed in Table 3 may cause permanent damage to the device. These are stress ratings only, and functional operation of
the device at these or any other conditions beyond those indicated in Section 6 Electrical Characteristics on page 4 is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
Table 3. Absolute Maximum Ratings
Parameter
Min
Max
Units
Comments
VIN to GND
-0.3
6
V
V
IN
+ 0.3
SW, EN, FB to GND
-0.3
V
Thermal Resistance ΘJA
ESD
207.4
ºC/W
kV
on PCB
HBM MIL-Std. 883E 3015.7 methods
JEDEC 78
2
Latch-Up
-100
-40
+100
+85
mA
ºC
Operating Temperature Range
Storage Temperature Range
-65
+125
ºC
The reflow peak soldering temperature (body
temperature) specified is in accordance with IPC/
JEDEC J-STD-020 “Moisture/Reflow Sensitivity
Classification for Non-Hermetic Solid State Surface
Mount Devices”.
Package Body Temperature
+260
125
ºC
ºC
The lead finish for Pb-free leaded packages is matte tin
(100% Sn).
Junction temperature (TJ) is calculated from the
ambient temperature (TAMB) and power dissipation
(PD) as:
Junction Temperature
T
J
= TAMB + (PD)(207.4ºC/W)
(EQ 1)
Moisture Sensitive Level
1
Represents an unlimited floor life time
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Revision 1.06
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AS1324
Datasheet - Electrical Characteristics
6 Electrical Characteristics
V
IN = EN = 3.6V, VOUT < VIN - 0.5V, TAMB = -40 to +85°C, typ. values @ TAMB = +25ºC (unless otherwise specified).
Table 4. Electrical Characteristics
Symbol
Parameter
Conditions
Min
Typ
Max
Units
V
IN
Input Voltage Range
2.7
5.5
V
Powersave Mode; VFB = 0.62V or VOUT = 103%,
I
Q
Quiescent Current
Shutdown Current
30
35
1
I
OUT = 0mA, TAMB = +25ºC
µA
I
SHDN
Shutdown Mode; VEN = 0V,
TAMB = +25ºC
0.1
Regulation
Regulated Feedback Voltage 1
V
FB
AS1324, IOUT = 100mA
IN = 2.7V to 5.5V
AMB = +25ºC
0.585
-30
0.6
0.1
0.615
V
Reference Voltage
Line Regulation
∆VFB
V
1
%/V
nA
IVFB
Feedback Current
T
30
AS1324-AD, IOUT = 100mA2
AS1324-12, IOUT = 100mA
AS1324-15, IOUT = 100mA
AS1324-18, IOUT = 100mA
V
FB
1.164
1.455
1.746
1.20
1.50
1.80
1.236
1.545
1.854
V
OUT
Regulated Output Voltage
V
Output Voltage
Line Regulation
∆VOUT
V
IN = 2.7 to 5.5V
0.1
1
%/V
Output Voltage
Load Regulation
V
LOADREG
IOUT = 0 to 100mA
0.02
%/mA
DC-DC Switches
V
IN = 3V, VFB = 0.5V or VOUT = 90%, TAMB =
IPK
Peak Inductor Current
0.5
0.75
1
A
25ºC
R
PFET
P-Channel FET RDS(ON)
N-Channel FET RDS(ON)
SW Leakage
I
LSW = 100mA
LSW = -100mA
EN = 0V, VSW = 0V or 5V
0.4
0.35
Ω
Ω
RNFET
LSW
I
I
V
±0.01
±1
µA
Control Inputs
V
EN
EN Threshold
0.3
1.2
1
1.5
±1
V
IEN
EN Leakage Current
±0.01
µA
Oscillator
V
FB = 0.6V or VOUT = 100%
1.5
1.8
MHz
kHz
fOSC
Oscillator Frequency
V
FB = 0V or VOUT = 0V, TAMB = 25ºC
115
1. The device is tested in a proprietary test mode where VFB is connected to the output of the error amplifier.
2. Please see Feedback Resistor Selection on page 13 for resistor values.
Note: All limits are guaranteed. The parameters with min and max values are guaranteed with production tests or SQC (Statistical Quality
Control) methods.
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Revision 1.06
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AS1324
Datasheet - Typical Operating Characteristics
7 Typical Operating Characteristics
Parts used for measurement: 4.7µH (MOS6020-472) Inductor, 10µF (GRM188R60J106ME47) CIN and COUT
.
Figure 3. Efficiency vs. Input Voltage; VOUT = 1.8V
Figure 4. Efficiency vs. Output Current; VOUT = 1.2V
95
100
95
90
85
80
75
70
65
60
55
50
90
85
80
75
70
65
60
55
50
IOUT = 600mA
IOUT = 100mA
IOUT = 10mA
IOUT = 1mA
VIN =2.5V
VIN =2.7V
VIN =3.7V
VIN =4.2V
VIN =5.5V
2.7
3.1
3.5
3.9
4.3
4.7
5.1
5.5
1
10
100
1000
Input Voltage (V)
Output Current (mA)
Figure 5. Efficiency vs. Output Current; VOUT = 1.5V
Figure 6. Efficiency vs. Output Current; VOUT = 1.8V
100
100
95
90
85
80
75
70
65
60
55
50
95
90
85
80
75
70
65
60
55
50
VIN = 2.5V
VIN = 2.7V
VIN = 3.7V
VIN = 4.2V
VIN = 5.5V
VIN = 2.5V
VIN = 2.7V
VIN = 3.7V
VIN = 4.2V
VIN = 5.5V
1
10
100
1000
1
10
100
1000
Output Current (mA)
Output Current (mA)
Figure 7. Efficiency vs. Output Current; VOUT = 2.5V
Figure 8. Efficiency vs. Output Current; VOUT = 3.3V
100
100
95
90
85
80
75
70
65
60
55
50
95
90
85
80
75
70
65
60
VIN = 3.7V
VIN = 4.2V
VIN = 5.5V
VIN =4.2V
VIN =5.5V
55
50
1
10
100
1000
1
10
100
1000
Output Current (mA)
Output Current (mA)
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Revision 1.06
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AS1324
Datasheet - Typical Operating Characteristics
Figure 9. Switching Frequency vs. Supply Voltage
Figure 10. Switching Frequency vs. Temperature
1.6
1.6
1.55
1.5
1.55
1.5
1.45
1.4
1.45
1.4
2.7
3.1
3.5
3.9
4.3
4.7
5.1
5.5
-45 -30 -15
0
15 30 45 60 75 90
Input Voltage (V)
Temperature (°C)
Figure 11. Feedback Voltage vs. Temperature
Figure 12. Output Voltage vs. Input Voltage
0.61
2
1.95
1.9
0.605
0.6
1.85
1.8
1.75
1.7
0.595
IOUT =600mA
IOUT =100mA
IOUT =10mA
IOUT =1mA
1.65
1.6
IOUT =100µA
0.59
2.7
3.1
3.5
3.9
4.3
4.7
5.1
5.5
-45 -30 -15
0
15 30 45 60 75 90
Temperature (C°)
Input Voltage (V)
Figure 13. VOUT vs. IOUT; VOUTNOM = 1.2V
Figure 14. VOUT vs. IOUT; VOUTNOM = 1.5V
1.3
1.6
Vin=2.5V
Vin=2.7V
Vin=5.5V
Vin=2.5V
Vin=2.7V
Vin=5.5V
1.25
1.2
1.55
1.5
1.15
1.1
1.45
1.4
0
100
200
300
400
500
600
0
100
200
300
400
500
600
Output Current (mA)
Output Current (mA)
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Revision 1.06
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AS1324
Datasheet - Typical Operating Characteristics
Figure 15. Quiescent Current vs. Input Voltage
Figure 16. Quiescent Current vs. Temperature
50
50
45
40
35
30
25
20
15
10
5
45
40
35
30
25
20
15
10
5
0
0
2.7
3.1
3.5
3.9
4.3
4.7
5.1
5.5
-45 -30 -15
0
15 30 45 60 75 90
Input Voltage (V)
Temperature (°C)
Figure 17. Load Step 0mA to 600mA
Figure 18. Load Step 10mA to 200mA
500µs/DIV
500µs/DIV
Figure 19. Startup
Figure 20. Powersave Mode
1ms/DIV
5µs/DIV
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Revision 1.06
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AS1324
Datasheet - Detailed Description
8 Detailed Description
The AS1324 is a high-efficiency buck converter that uses a constant-frequency current-mode architecture. The device contains two internal
MOSFET switches and is available in adjustable- and fixed-output voltage versions.
Figure 21. AS1324 - Block Diagram
Ramp
Compensator
–
ICOMP
VIN
4
OSC
V
IN
C
10µF
IN
+
OSCN
Frequency
Shift
5
AS1324
V
OUT/VFB
0.6V
+
R
R
1
2
Error
Amp
–
FB
–
PMOS
NMOS
OVDET
+
Digital
Logic
Anti-
Shoot
Through
0.6V +
∆VOVL
4.7µH
V
OUT
3
–
+
SW
C
OUT
1
10µF
0.6V
Reference
0.6V -
∆VOVL
EN
+
IRCMP
Shutdown
2
–
GND
Not applicable to AS1324
AS1324-12: R
AS1324-15: R
AS1324-18: R
1
1
1
+ R
+ R
+ R
2 = 600kΩ
2
= 750kΩ
= 900kΩ
2
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Revision 1.06
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AS1324
Datasheet - Detailed Description
8.1 Main Control Loop
During PWM operation the converters use a 1.5MHz fixed-frequency, current-mode control scheme. Basis of the current-mode PWM controller is
an open-loop, multiple input comparator that compares the error-amp voltage feedback signal against the sum of the amplified current-sense
signal and the slope-compensation ramp. At the beginning of each clock cycle, the internal high-side PMOS turns on until the PWM comparator
trips. During this time the current in the inductor ramps up, sourcing current to the output and storing energy in the inductor’s magnetic field.
When the PMOS turns off, the internal low-side NMOS turns on. Now the inductor releases the stored energy while the current ramps down, still
providing current to the output. The output capacitor stores charge when the inductor current exceeds the load and discharges when the inductor
current is lower than the load. Under overload conditions, when the inductor current exceeds the current limit, the high-side PMOS is turned off
and the low-side NMOS remains on until the next clock cycle.
When the PMOS is off, the NMOS is turned on until the inductor current starts to reverse (as indicated by the current reversal comparator
(IRCMP)), or the next clock cycle begins. The IRCMP detects the zero crossing.
The peak inductor current (IPK) is controlled by the error amplifier. When IOUT increases, VFB decreases slightly relative to the internal 0.6V
reference, causing the error amplifier’s output voltage to increase until the average inductor current matches the new load current.
The over-voltage detection comparator (OVDET) guards against transient overshoots by turning the main switch off and keeping it off until the
transient is removed.
8.2 Powersave Operation
The AS1324 uses an automatic powersave mode where the peak inductor current (IPK) is set to approximately 200mA while independent of the
output load. In powersave mode, load current is supplied solely from the output capacitor. As the output voltage drops, the error amplifier output
rises above the powersave threshold signaling to switch into PWM fixed frequency mode and turn the PMOS on. This process repeats at a rate
determined by the load demand.
Each burst event can last from a few cycles at light loads to almost continuous cycling (with short sleep intervals) at moderate loads. In between
bursts, the power MOSFETs are turned off, as is any unneeded circuitry, reducing quiescent current to 30µA.
8.3 Short-Circuit Protection
In cases where the AS1324 output is shorted to ground, the oscillator frequency (fOSC) is reduced to 1/13 the nominal frequency ( 115kHz).
This frequency reduction ensures that the inductor current has more time to decay, thus preventing runaway conditions. fOSC will progressively
increase to 1.5MHz when VFB/VOUT > 0V.
8.4 Shutdown
Connecting EN to GND or logic low places the AS1324 in shutdown mode and reduces the supply current to 0.1µA. In shutdown the control
circuitry and the internal NMOS and PMOS turn off and SW becomes high impedance disconnecting the input from the output. The output
capacitance and load current determine the voltage decay rate. For normal operation connect EN to VIN or logic high.
Note: Pin EN should not be left floating.
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Revision 1.06
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AS1324
Datasheet - Application Information
9 Application Information
The AS1324 is perfect for mobile communications equipment like cell phones and smart phones, digital cameras and camcorders, portable MP3
and DVD players, PDA’s and palmtop computers and any other handheld instruments.
Figure 22. Single Li-Ion 1.2V/600mA Regulator for High-Efficiency
4.7µH
V
1.2V
600mA
OUT
4
3
V
IN
2.7 to 4.2V
V
IN
SW
CIN
COUT
2.2µF
10µF
22pF
AS1324
301kΩ
5
1
R
2
EN
V
FB
301kΩ
R1
GND
2
Figure 23. 5V Input to 3.3V/600mA Buck Regulator
4.7µH
V
3.3V
600mA
OUT
4
3
V
5V
IN
C
4.7µF
IN
C
10µF
OUT
V
IN
SW
22pF
AS1324
301kΩ
5
V
1
R
2
EN
FB
R1
66.5kΩ
GND
2
Figure 24. Single Li-Ion 1.5V/600mA Regulator for High-Efficiency
4.7µH
V
OUT
4
3
SW
V
IN
1.5V
600mA
2.7 to 4.2V
V
IN
C
10µF
OUT
C
4.7µF
IN
AS1324-15
5
1
EN
VOUT
GND
2
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Revision 1.06
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AS1324
Datasheet - Application Information
Figure 25. Single Li-Ion 1.8V/600mA Regulator for Low Output Ripple
4.7µH
V
1.8V
600mA
OUT
4
3
V
IN
2.7 to 4.2V
V
IN
SW
C
10µF
IN
C
22µF
OUT
AS1324-18
5
1
EN
V
OUT
GND
2
9.1 External Component Selection
9.2 Inductor Selection
For most applications the value of the external inductor should be in the range of 2.2 to 6.8µH as the inductor value has a direct effect on the
ripple current. The selected inductor must be rated for its DC resistance and saturation current. The inductor ripple current (∆I ) decreases with
higher inductance and increases with higher VIN or VOUT
L
.
In Equation (EQ 2) the maximum inductor current in PWM mode under static load conditions is calculated. The saturation current of the inductor
should be rated higher than the maximum inductor current as calculated with Equation (EQ 3). This is recommended because the inductor
current will rise above the calculated value during heavy load transients.
VOUT
--------------
1 –
(EQ 2)
(EQ 3)
VIN
-----------------------
×
∆IL = VOUT
L × f
∆IL
-------
+
ILMAX = IOUTMAX
2
Where:
f = Switching Frequency (1.5 MHz typical)
L = Inductor Value
I
∆
Lmax = Maximum Inductor current
L = Peak to Peak inductor ripple current
The recommended starting point for setting ripple current is ∆IL = 240mA (40% of 600mA).
I
The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation.
Thus, a 720mA rated inductor should be sufficient for most applications (600mA + 120mA). A easy and fast approach is to select the inductor
current rating fitting to the maximum switch current limit of the converter.
Note: For highest efficiency, a low DC-resistance inductor is recommended.
Accepting larger values of ripple current allows the use of low inductance values, but results in higher output voltage ripple, greater core losses,
and lower output current capability.
The total losses of the coil have a strong impact on the efficiency of the dc/dc conversion and consist of both the losses in the dc resistance and
the following frequency-dependent components:
1. The losses in the core material (magnetic hysteresis loss, especially at high switching frequencies)
2. Additional losses in the conductor from the skin effect (current displacement at high frequencies)
3. Magnetic field losses of the neighboring windings (proximity effect)
4. Radiation losses
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AS1324
Datasheet - Application Information
Table 5. Recommended Inductors
Part Number
L
DCR
Current Rating
790mA
Dimensions (L/W/T)
3.2x2.5x2.0mm
3.2x2.5x2.0mm
3.1x3.1x0.8mm
3.1x3.1x1.5mm
6.0x6.8x2.4mm
6.0x6.8x2.4mm
4.0x4.0x1.8mm
4.0x4.0x1.8mm
Manufacturer
Murata
LQH32CN2R2M33
LQH32CN4R7M33
LPS3008-222MLC
2.2µH
4.7µH
2.2µH
2.2µH
2.2µH
4.7µH
2.2µH
4.7µH
97mΩ
150mΩ
175mΩ
110mΩ
35mΩ
50mΩ
72mΩ
105mΩ
www.murata.com
650mA
Coilcraft
www.coilcraft.com
1100mA
2000mA
3260mA
1820mA
1200mA
900mA
LPS3015-222MLC
MOS6020-222MLC
MOS6020-472MLC
CDRH3D16NP-2R2N
CDRH3D16ND-4R7N
Sumida
www.sumida.com
Figure 26. Efficiency Comparison of Different Inductors, VIN = 2.7V, VOUT = 1.8V and 1.2V
95
90
85
80
75
70
95
90
85
80
75
70
V
OUT = 1.8V
V
OUT = 1.2V
LQH32CN2R2
LPS3015-222
LQH32CN4R7
LPS3008-222
M OS6020-222
M OS6020-472
LQH32CN2R2
LPS3015-222
LQH32CN4R7
LPS3008-222
M OS6020-222
M OS6020-472
1
10
100
1000
1
10
100
1000
Output Current (mA)
Output Current (mA)
9.3 Output Capacitor Selection
The advanced fast-response voltage mode control scheme of the AS1324 allows the use of tiny ceramic capacitors. Because of their lowest
output voltage ripple low ESR ceramic capacitors are recommended. X7R or X5R dielectric output capacitor are recommended.
At high load currents, the device operates in PWM mode and the RMS ripple current is calculated as:
VOUT
--------------
1 –
(EQ 4)
VIN
----------------------- ----------------
1
IRMSC
= VOUT
×
×
OUT
L × f
2 ×
3
While operating in PWM mode the overall output voltage ripple is the sum of the voltage spike caused by the output capacitor ESR plus the
voltage ripple caused by charging and discharging the output capacitor:
VOUT
--------------
1 –
(EQ 5)
VIN
-----------------------
1
--------------------------------
∆VOUT = VOUT
×
×
+ ESR
L × f
8 × COUT × f
Higher value, low cost ceramic capacitors are available in very small case sizes, and their high ripple current, high voltage rating, and low ESR
make them ideal for switching regulator applications. Because the AS1324 control loop is not dependant on the output capacitor ESR for stable
operation, ceramic capacitors can be used to achieve very low output ripple and accommodate small circuit size.
At light loads, the converter operates in powersave mode and the output voltage ripple is in direct relation to the output capacitor and inductor
value used. Larger output capacitor and inductor values minimize the voltage ripple in powersave mode and tighten DC output accuracy in
powersave mode.
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Revision 1.06
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AS1324
Datasheet - Application Information
9.4 Input Capacitor Selection
In continuous mode, the source current of the PMOS is a square wave of the duty cycle VOUT/VIN. To prevent large voltage transients while
minimizing the interference with other circuits caused by high input voltage spikes, a low ESR input capacitor sized for the maximum RMS
current must be used. The maximum RMS capacitor current is given as:
(EQ 6)
V
OUT × (VIN – VOUT
-----------------------------------------------------------
×
)
IRMS = IMAX
VIN
where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX = ILIM - ∆I
L/2
This formula has a maximum at VIN = 2VOUT where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because
even significant deviations only provide negligible affects.
The input capacitor can be increased without any limit for better input voltage filtering. Take care when using small ceramic input capacitors.
When a small ceramic capacitor is used at the input, and the power is being supplied through long wires, such as from a wall adapter, a load step
at the output, or VIN step on the input, can induce ringing at the VIN pin. This ringing can then couple to the output and be mistaken as loop
instability, or could even damage the part by exceeding the maximum ratings.
9.4.1 Ceramic Input and Output Capacitors
When choosing ceramic capacitors for CIN and COUT, the X5R or X7R dielectric formulations are recommended. These dielectrics have the
best temperature and voltage characteristics for a given value and size. Y5V and Z5U dielectric capacitors, aside from their wide variation in
capacitance over temperature, become resistive at high frequencies and therefore should not be used.
Table 6. Recommended Input and Output Capacitor
Part Number
C
TC Code
Rated Voltage
Dimensions (L/W/T)
Manufacturer
Taiyo Yuden
www.t-yuden.com
JMK212BJ226MG-T
22µF
X5R
6.3V
0805
0603
0805
Murata
www.murata.com
GRM188R60J106ME47
GRM21BR71A475KA73
10µF
X5R
X7R
6.3V
10V
4.7µF
Because ceramic capacitors lose a lot of their initial capacitance at their maximum rated voltage, it is recommended that either a higher input
capacity or a capacitance with a higher rated voltage is used.
9.5 Feedback Resistor Selection
In the AS1324-AD, the output voltage is set by an external resistor divider connected to VFB (see Figure 27). This circuitry allows for remote
voltage sensing and adjustment.
Figure 27. Setting the AS1324 Output Voltage
0.6V ≤ VOUT ≤ 5.5V
R
R
2
5
R1<<R2
V
FB
1
AS1324
2
GND
Resistor values for the circuit shown in Figure 27 can be calculated as:
VOUT = 0,6 × 1 +
R2
------
R1
(EQ 7)
The output voltage can be adjusted by selecting different values for R
1 and R2. For R1 a value between 10kΩ and 500kΩ is recommended. A
higher resistance of R and R will result in a lower leakage current at the output. It is recommended to keep VIN 500mV higher than VOUT
1
2
.
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AS1324
Datasheet - Application Information
9.6 Efficiency
The efficiency of a switching regulator is equivalent to:
Efficiency = (POUT/PIN)100%
(EQ 8)
For optimum design, an analysis of the AS1324 is needed to determine efficiency limitations and to determine design changes for improved
efficiency. Efficiency can be expressed as:
Efficiency = 100% – (L
1
+ L2
+ L3
+ ...)
(EQ 9)
Where:
, L , L3, etc. are the individual losses as a percentage of input power.
L1
2
Although all dissipative elements in the circuit produce losses, those four main sources should be considered for efficiency calculation:
9.6.1 Input Voltage Quiescent Current Losses
The VIN current is the DC supply current given in the electrical characteristics which excludes MOSFET driver and control currents. VIN current
results in a small (<0.1%) loss that increases with VIN, even at no load. The VIN quiescent current loss dominates the efficiency loss at very low
load currents.
9.6.2 I²R Losses
Most of the efficiency loss at medium to high load currents are attributed to I²R loss, and are calculated from the resistances of the internal
switches (RSW) and the external inductor (RL). In continuous mode, the average output current flowing through inductor L is split between the
internal switches. Therefore, the series resistance looking into the SW pin is a function of both NMOS & PMOS RDS(ON) as well as the duty
cycle (DC) and can be calculated as follows:
R
SW = (RDS(ON)PMOS)(DC) + (RDS(ON)NMOS)(1 – DC)
(EQ 10)
The RDS(ON) for both MOSFETs can be obtained from the Electrical Characteristics on page 4. Thus, to obtain I²R losses calculate as follows:
I²R losses = IOUT²(RSW + R (EQ 11)
L
)
9.6.3 Switching Losses
The switching current is the sum of the control currents and the MOSFET driver. The MOSFET driver current results from switching the gate
capacitance of the power MOSFETs. If a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to
ground. The resulting dQ/dt is a current out of VIN that is typically much larger than the DC bias current. In continuous mode:
IGC = f(QPMOS + QNMOS
)
(EQ 12)
Where: QPMOS and QNMOS are the gate charges of the internal MOSFET switches.
The losses of the gate charges are proportional to VIN and thus their effects will be more visible at higher supply voltages.
9.6.4 Other Losses
Basic losses in the design of a system should also be considered. Internal battery resistances and copper trace can account for additional
efficiency degradations in battery operated systems. By making sure that CIN has adequate charge storage and very low ESR at the given
switching frequency, the internal battery and fuse resistance losses can be minimized. CIN and COUT ESR dissipative losses and inductor core
losses generally account for less than 2% total additional loss.
9.7 Thermal Shutdown
Due to its high-efficiency design, the AS1324 will not dissipate much heat in most applications. However, in applications where the AS1324 is
running at high ambient temperature, uses a low supply voltage, and runs with high duty cycles (such as in dropout) the heat dissipated may
exceed the maximum junction temperature of the device.
As soon as the junction temperature reaches approximately 150ºC the AS1324 goes in thermal shutdown. In this mode the internal PMOS &
NMOS switch are turned off. The device will power up again, as soon as the temperature falls below +145°C again.
9.8 Checking Transient Response
The main loop response can be evaluated by examining the load transient response. Switching regulators normally take several cycles to
respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equivalent to:
VDROP = ∆IOUT x ESR
(EQ 13)
Where:
ESR is the effective series resistance of COUT
.
∆IOUT also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its
steady-state value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem.
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AS1324
Datasheet - Application Information
9.9 Design Example
Figure 28 shows the AS1324 used in a single lithium-ion (3.7V typ) battery-powered mobile phone application. The load current requirement is
600mA (max) but most of the time the device will require only 2mA (standby mode current).
Figure 28. Design Example
2.2µH
4
3
V
OUT
V
IN
2.2V
3.7V
V
IN
SW
CIN
COUT
10µF
CER
4.7µF
CER
22pF
AS1324
1MΩ
5
1
R2
EN
VFB
R
1
375kΩ
GND
2
For the circuit shown in Figure 28, efficiency at low- and high-load currents is an important consideration when selecting the value for the
external inductor, which is calculated as:
VOUT
--------------
f∆IL
VOUT
--------------
VIN
(EQ 14)
L =
× 1 –
From (EQ 14), substituting VOUT = 2.2V, VIN = 3.7V, ∆IL = 240mA and f = 1.5MHz gives:
2,2V
2,2V
(EQ 15)
---------------------------------------------------
------------
= 2,48µH
L =
× 1 –
3,7V
(1,5MHz × 240mA)
Therefore, a standard 2.2µH inductor should be used for this design.
For best overall efficiency use an inductor with a rating of 720mA or greater and less than 0.2Ω series resistance. CIN will require an RMS
current rating of at least 0.3A
satisfy this requirement.
ILOAD(MAX)/2, whereas COUT will require an ESR of less than 0.25Ω. In most cases, a ceramic capacitor will
For the feedback resistors, select the value for R
1
= 375kΩ. R can then be calculated from (EQ 7) to be:
2
R2
= (VOUT/0.6 - 1)375k = 1000kΩ
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AS1324
Datasheet - Application Information
9.10 Layout Considerations
The AS1324 requires proper layout and design techniques for optimum performance.
ꢀ
ꢀ
The power traces (GND, SW, and VIN) should be kept as short, direct, and wide as is practical.
Pin VFB (AS1324 only) should be connected directly to the feedback resistors (R and R ). A potentiometer as replacement for R
should be avoided to minimize the output voltage ripple and to maintain the stability of the regulator.
The resistive divider (R /R ) must be connected between the positive plate of COUT and ground.
1
2
1 and R2
ꢀ
ꢀ
1
2
The positive plate of CIN should be connected as close to VIN as is practical since CIN provides the AC current to the internal power MOS-
FETs.
ꢀ
ꢀ
Switching node SW should be kept far away from the sensitive VFB node.
The negative plates of CIN and COUT should be kept as close to each other as is practical. A starpoint to Ground is recommended.
Figure 29. AS1324 Basic PCB Layout
R1
V
IN
Via to VIN
R2
Via to GND
1
5
AS1324
2
3
VOUT
L1
CFWD
Via to VOUT
SW
4
COUT
CIN
GND
Figure 30. AS1324 Basic Diagram
High Current Path
1
5
EN
VFB
AS1324
2
R
2
R
1
GND
C
OUT
CFWD
V
OUT
3
SW
4
VIN
L1
CIN
VIN
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AS1324
Datasheet - Application Information
Figure 31. AS1324-18 Basic PCB Layout
Via to VIN
V
IN
Via to VOUT
1
5
4
AS1324-18
2
3
VOUT
L
1
SW
COUT
CIN
GND
Figure 32. AS1324-18 Basic Diagram
High Current Path
1
5
EN
VOUT
AS1324-18
2
GND
C
OUT
V
OUT
3
4
VIN
SW
L1
CIN
V
IN
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AS1324
Datasheet - Package Drawings and Markings
10 Package Drawings and Markings
The device is available in an 5-pin TSOT-23 package.
Figure 33. 5-pin TSOT-23 Package
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AS1324
Datasheet - Package Drawings and Markings
Figure 34. 5-pin TSOT-23 Marking
Pin1
Bottom
Top
ZZZZ XXXX
Pin1
Package Code:
ZZZZ - Marking
XXXX - encoded Datecode
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AS1324
Datasheet
11 Ordering Information
The device is available as the following standard versions.
Table 7. Ordering Information
Ordering Code
Marking
Output
Description
Delivery Form
Package
1.5MHz, 600mA Synchronous DC/DC
Converter
5-pin TSOT-23
AS1324-BTTT-AD
ASKR
adjustable
Tape and Reel
1.5MHz, 600mA Synchronous DC/DC
Converter
5-pin TSOT-23
5-pin TSOT-23
5-pin TSOT-23
AS1324-BTTT-12
AS1324-BTTT-15
AS1324-BTTT-18
ASKT
ASKU
ASKS
1.2V
1.5V
1.8V
Tape and Reel
Tape and Reel
Tape and Reel
1.5MHz, 600mA Synchronous DC/DC
Converter
1.5MHz, 600mA Synchronous DC/DC
Converter
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AS1324
Datasheet - Ordering Information
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