EVAL-ADF7021-NDBIZ1 [ADI]
High Performance Narrow-Band Transceiver IC; 高性能窄带收发器IC型号: | EVAL-ADF7021-NDBIZ1 |
厂家: | ADI |
描述: | High Performance Narrow-Band Transceiver IC |
文件: | 总64页 (文件大小:2499K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
High Performance
Narrow-Band Transceiver IC
ADF7021-N
On-chip VCO and fractional-N PLL
FEATURES
On-chip, 7-bit ADC and temperature sensor
Fully automatic frequency control loop (AFC)
Digital received signal strength indication (RSSI)
Integrated Tx/Rx switch
Low power, narrow-band transceiver
Frequency bands using dual VCO
80 MHz to 650 MHz
842 MHz to 916 MHz
Programmable IF filter bandwidths of
9 kHz, 13.5 kHz, and 18.5 kHz
Modulation schemes: 2FSK, 3FSK, 4FSK, MSK
Spectral shaping: Gaussian and raised cosine filtering
Data rates supported: 0.05 kbps to 24 kbps
2.3 V to 3.6 V power supply
0.1 μA leakage current in power-down mode
APPLICATIONS
Narrow-band, short range device (SRD) standards
ARIB STD-T67, ETSI EN 300 220, Korean SRD standard,
FCC Part 15, FCC Part 90, FCC Part 95
Low cost, wireless data transfer
Remote control/security systems
Wireless metering
Wireless medical telemetry service (WMTS)
Home automation
Process and building control
Programmable output power
−16 dBm to +13 dBm in 63 steps
Automatic power amplifier (PA) ramp control
Receiver sensitivity
−130 dBm at 100 bps, 2FSK
−122 dBm at 1 kbps, 2FSK
Pagers
Patent pending, on-chip image rejection calibration
FUNCTIONAL BLOCK DIAGRAM
CE
RSET
MUXOUT
CREG(1:4)
TEMP
SENSOR
MUX
7-BIT ADC
R
LDO(1:4)
TEST MUX
LNA
2FSK
3FSK
4FSK
LNA
TxRxCLK
CLOCK
AND DATA
RECOVERY
R
FIN
RSSI/
LOG AMP
TxRxDATA
Tx/Rx
CONTROL
IF FILTER
R
FINB
DEMODULATOR
SWD
GAIN
AGC
CONTROL
SLE
SERIAL
PORT
SDATA
SREAD
SCLK
AFC
CONTROL
PA RAMP
2FSK
3FSK
4FSK
GAUSSIAN/
RAISED COSINE
FILTER
Σ-Δ
MODULATOR
÷1/÷2
DIV P
N/N + 1
RFOUT
MOD CONTROL
÷2
VCO1
3FSK
ENCODING
MUX
CP
PFD
VCO2
CLK
DIV
DIV R
OSC
L1 L2
VCOIN CPOUT
OSC1 OSC2
CLKOUT
Figure 1.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registeredtrademarks arethe property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.461.3113
www.analog.com
©2008 Analog Devices, Inc. All rights reserved.
ADF7021-N
TABLE OF CONTENTS
Features .............................................................................................. 1
Demodulation, Detection, and CDR....................................... 32
Receiver Setup............................................................................. 34
Demodulator Considerations................................................... 36
AFC Operation ........................................................................... 36
Automatic Sync Word Detection (SWD)................................ 37
Applications Information.............................................................. 38
IF Filter Bandwidth Calibration............................................... 38
LNA/PA Matching...................................................................... 39
Image Rejection Calibration..................................................... 40
Packet Structure and Coding.................................................... 42
Programming After Initial Power-Up ..................................... 42
Applications Circuit................................................................... 45
Serial Interface ................................................................................ 46
Readback Format........................................................................ 46
Interfacing to a Microcontroller/DSP ..................................... 48
Register 0—N Register............................................................... 49
Register 1—VCO/Oscillator Register...................................... 50
Register 2—Transmit Modulation Register ............................ 51
Register 3—Transmit/Receive Clock Register........................ 52
Register 4—Demodulator Setup Register ............................... 53
Register 5—IF Filter Setup Register......................................... 54
Register 6—IF Fine Cal Setup Register ................................... 55
Register 7—Readback Setup Register...................................... 56
Register 8—Power-Down Test Register .................................. 57
Register 9—AGC Register......................................................... 58
Register 10—AFC Register ....................................................... 59
Register 11—Sync Word Detect Register................................ 60
Register 12—SWD/Threshold Setup Register........................ 60
Register 13—3FSK/4FSK Demod Register ............................. 61
Register 14—Test DAC Register............................................... 62
Register 15—Test Mode Register ............................................. 63
Outline Dimensions....................................................................... 64
Ordering Guide .......................................................................... 64
Applications....................................................................................... 1
Functional Block Diagram .............................................................. 1
Revision History ............................................................................... 2
General Description......................................................................... 3
Specifications..................................................................................... 4
RF and PLL Specifications........................................................... 4
Transmission Specifications........................................................ 5
Receiver Specifications ................................................................ 6
Digital Specifications ................................................................... 9
General Specifications ............................................................... 10
Timing Characteristics .............................................................. 11
Timing Diagrams........................................................................ 12
Absolute Maximum Ratings.......................................................... 15
ESD Caution................................................................................ 15
Pin Configuration and Function Descriptions........................... 16
Typical Performance Characteristics ........................................... 18
Frequency Synthesizer ................................................................... 22
Reference Input........................................................................... 22
MUXOUT.................................................................................... 23
Voltage Controlled Oscillator (VCO)...................................... 24
Choosing Channels for Best System Performance................. 25
Transmitter ...................................................................................... 26
RF Output Stage.......................................................................... 26
Modulation Schemes.................................................................. 26
Spectral Shaping ......................................................................... 28
Modulation and Filtering Options........................................... 29
Transmit Latency........................................................................ 29
Test Pattern Generator............................................................... 29
Receiver Section.............................................................................. 30
RF Front End............................................................................... 30
IF Filter......................................................................................... 30
RSSI/AGC.................................................................................... 30
REVISION HISTORY
2/08—Revision 0: Initial Version
Rev. 0 | Page 2 of 64
ADF7021-N
GENERAL DESCRIPTION
The ADF7021-N is a high performance, low power, narrow-
band transceiver based on the ADF7021. The ADF7021-N has
IF filter bandwidths of 9 kHz, 13.5 kHz, and 18.5 kHz, making
it ideally suited to worldwide narrowband standards and
particularly those that stipulate 12.5 kHz channel separation.
The frequency-agile PLL allows the ADF7021-N to be used in
frequency-hopping, spread spectrum (FHSS) systems. Both
VCOs operate at twice the fundamental frequency to reduce
spurious emissions and frequency pulling problems.
The transmitter output power is programmable in 63 steps from
−16 dBm to +13 dBm and has an automatic power ramp control
to prevent spectral splatter and help meet regulatory standards.
The transceiver RF frequency, channel spacing, and modulation
are programmable using a simple 3-wire interface. The device
operates with a power supply range of 2.3 V to 3.6 V and can be
powered down when not in use.
It is designed to operate in the narrow-band, license-free ISM
bands and in the licensed bands with frequency ranges of 80
MHz to 650 MHz and 842 MHz to 916 MHz. The part has both
Gaussian and raised cosine transmit data filtering options to
improve spectral efficiency for narrow-band applications. It is
suitable for circuit applications targeted at the Japanese ARIB
STD-T67, the European ETSI EN 300 220, the Korean short
range device regulations, the Chinese short range device
regulations, and the North American FCC Part 15, Part 90, and
Part 95 regulatory standards. A complete transceiver can be
built using a small number of external discrete components,
making the ADF7021-N very suitable for price-sensitive and
area-sensitive applications.
A low IF architecture is used in the receiver (100 kHz), which
minimizes power consumption and the external component
count yet avoids dc offset and flicker noise at low frequencies.
The IF filter has programmable bandwidths of 9 kHz, 13.5 kHz,
and 18.5 kHz. The ADF7021-N supports a wide variety of pro-
grammable features including Rx linearity, sensitivity, and IF
bandwidth, allowing the user to trade off receiver sensitivity
and selectivity against current consumption, depending on the
application. The receiver also features a patent-pending automatic
frequency control (AFC) loop with programmable pull-in range
that allows the PLL to track out the frequency error in the
incoming signal.
The range of on-chip FSK modulation and data filtering options
allows users greater flexibility in their choice of modulation
schemes while meeting the tight spectral efficiency requirements.
The ADF7021-N also supports protocols that dynamically
switch among 2FSK, 3FSK, and 4FSK to maximize communica-
tion range and data throughput.
The receiver achieves an image rejection performance of 56 dB
using a patent-pending IR calibration scheme that does not
require the use of an external RF source.
The transmit section contains two voltage controlled oscillators
(VCOs) and a low noise fractional-N PLL with an output
resolution of <1 ppm. The ADF7021-N has a VCO using an
internal LC tank (421 MHz to 458 MHz, 842 MHz to 916 MHz)
and a VCO using an external inductor as part of its tank circuit
(80 MHz to 650 MHz). The dual VCO design allows dual-band
operation where the user can transmit and/or receive at any
frequency supported by the internal inductor VCO and can also
transmit and/or receive at a particular frequency band
supported by the external inductor VCO.
An on-chip ADC provides readback of the integrated tempera-
ture sensor, external analog input, battery voltage, and RSSI
signal, which provides savings on an ADC in some applications.
The temperature sensor is accurate to 10ꢀC over the full oper-
ating temperature range of −40ꢀC to +85ꢀC. This accuracy can
be improved by performing a 1-point calibration at room
temperature and storing the result in memory.
Rev. 0 | Page 3 of 64
ADF7021-N
SPECIFICATIONS
VDD = 2.3 V to 3.6 V, GND = 0 V, TA = TMIN to TMAX, unless otherwise noted. Typical specifications are at VDD = 3 V, TA = 25ꢀC.
All measurements are performed with the EVAL-ADF7021-NDBxx using the PN9 data sequence, unless otherwise noted.
RF AND PLL SPECIFICATIONS
Table 1.
Parameter
Min
Typ
Max Unit
Test Conditions/Comments
RF CHARACTERISTICS
See Table 9 for required VCO_BIAS and
VCO_ADJUST settings
Frequency Ranges (Direct Output)
160
842
80
650
916
325
458
24
MHz
MHz
MHz
MHz
MHz
External inductor VCO
Internal inductor VCO
External inductor VCO, RF divide-by-2 enabled
Internal inductor VCO, RF divide-by-2 enabled
Frequency Ranges (RF Divide-by-2 Mode)
421
Phase Frequency Detector (PFD) Frequency1 RF/256
PHASE-LOCKED LOOP (PLL)
VCO Gain2
868 MHz, Internal Inductor VCO
426 MHz, Internal Inductor VCO
426 MHz, External Inductor VCO
160 MHz, External Inductor VCO
Phase Noise (In-Band)
67
45
27
6
MHz/V VCO_ADJUST = 0, VCO_BIAS = 8
MHz/V VCO_ADJUST = 0, VCO_BIAS = 8
MHz/V VCO_ADJUST = 0, VCO_BIAS = 3
MHz/V VCO_ADJUST = 0, VCO_BIAS = 2
868 MHz, Internal Inductor VCO
−97
dBc/Hz 10 kHz offset, PA = 10 dBm, VDD = 3.0 V,
PFD = 19.68 MHz, VCO_BIAS = 8
433 MHz, Internal Inductor VCO
426 MHz, External Inductor VCO
Phase Noise (Out-of-Band)
−103
−95
dBc/Hz 10 kHz offset, PA = 10 dBm, VDD = 3.0 V,
PFD = 19.68 MHz, VCO_BIAS = 8
dBc/Hz 10 kHz offset, PA = 10 dBm, VDD = 3.0 V,
PFD = 9.84 MHz, VCO_BIAS = 3
dBc/Hz 1 MHz offset, fRF = 433 MHz, PA = 10 dBm,
VDD = 3.0 V, PFD = 19.68 MHz, VCO_BIAS = 8
−124
Normalized In-Band Phase Noise Floor3
PLL Settling
−203
40
dBc/Hz
μs
Measured for a 10 MHz frequency step to within
5 ppm accuracy, PFD = 19.68 MHz, loop bandwidth
(LBW) = 100 kHz
REFERENCE INPUT
Crystal Reference4
External Oscillator4, 5
Crystal Start-Up Time6
XTAL Bias = 20 μA
XTAL Bias = 35 μA
Input Level for External Oscillator7
OSC1
3.625
3.625
24
24
MHz
MHz
0.930
0.438
ms
ms
10 MHz XTAL, 33 pF load capacitors, VDD = 3.0 V
10 MHz XTAL, 33 pF load capacitors, VDD = 3.0 V
0.8
CMOS levels
V p-p
V
Clipped sine wave
OSC2
ADC PARAMETERS
INL
DNL
0.4
0.4
LSB
LSB
VDD = 2.3 V to 3.6 V, TA = 25°C
VDD = 2.3 V to 3.6 V, TA = 25°C
1 The maximum usable PFD at a particular RF frequency is limited by the minimum N divide value.
2 VCO gain measured at a VCO tuning voltage of 0.7 V. The VCO gain varies across the tuning range of the VCO. The software package ADIsimPLL™ can be used to model this
variation.
3 This value can be used to calculate the in-band phase noise for any operating frequency. Use the following equation to calculate the in-band phase noise performance
as seen at the power amplifier (PA) output: −203 + 10 log(fPFD) + 20 logN.
4 Guaranteed by design. Sample tested to ensure compliance.
5 A TCXO, VCXO, or OCXO can be used as an external oscillator.
6 Crystal start-up time is the time from chip enable (CE) being asserted to correct clock frequency on the CLKOUT pin.
7 Refer to the Reference Input section for details on using an external oscillator.
Rev. 0 | Page 4 of 64
ADF7021-N
TRANSMISSION SPECIFICATIONS
Table 2.
Parameter
Min
Typ
Max
Unit Test Conditions/Comments
DATA RATE
2FSK, 3FSK
4FSK
0.05
0.05
18.51 kbps IF_FILTER_BW = 18.5 kHz
24 kbps IF_FILTER_BW = 18.5 kHz
MODULATION
Frequency Deviation (fDEV
2
)
0.056
0.306
56
28.26 kHz
PFD = 3.625 MHz
PFD = 20 MHz
PFD = 3.625 MHz
156
kHz
Hz
Deviation Frequency Resolution
Gaussian Filter BT
0.5
Raised Cosine Filter Alpha
0.5/0.7
Programmable
TRANSMIT POWER
Maximum Transmit Power3
Transmit Power Variation vs.
Temperature
+13
1
dBm VDD = 3.0 V, TA = 25°C
dB
−40°C to +85°C
Transmit Power Variation vs. VDD
Transmit Power Flatness
Programmable Step Size
1
1
dB
dB
dB
2.3 V to 3.6 V at 915 MHz, TA = 25°C
902 MHz to 928 MHz, 3 V, TA = 25°C
−16 dBm to +13 dBm
0.3125
ADJACENT CHANNEL POWER (ACP)
426 MHz, External Inductor VCO
12.5 kHz Channel Spacing
PFD = 9.84 MHz
−50
−50
dBc
dBc
Gaussian 2FSK modulation, measured in a 4.25 kHz bandwidth
at 12.5 kHz offset, 2.4 kbps PN9 data, 1.2 kHz frequency deviation,
compliant with ARIB STD-T67
Gaussian 2FSK modulation, measured in a 8 kHz bandwidth at
25 kHz offset, 9.6 kbps PN9 data, 2.4 kHz frequency deviation,
compliant with ARIB STD-T67
25 kHz Channel Spacing
868 MHz, Internal Inductor VCO
12.5 kHz Channel Spacing
PFD = 19.68 MHz
−46
−43
dBm Gaussian 2FSK modulation, 10 dBm output power, measured in
a 6.25 kHz bandwidth at 12.5 kHz offset, 2.4 kbps PN9 data,
1.2 kHz frequency deviation, compliant with ETSI EN 300 220
dBm Gaussian 2FSK modulation, 10 dBm output power, measured in
a 12.5 kHz bandwidth at 25 kHz offset, 9.6 kbps PN9 data,
2.4 kHz frequency deviation, compliant with ETSI EN 300 220
25 kHz Channel Spacing
433 MHz, Internal Inductor VCO
12.5 kHz Channel Spacing
PFD = 19.68 MHz
−50
−47
dBm Gaussian 2FSK modulation, 10 dBm output power, measured in
a 6.25 kHz bandwidth at 12.5 kHz offset, 2.4 kbps PN9 data,
1.2 kHz frequency deviation, compliant with ETSI EN 300 220
dBm Gaussian 2FSK modulation, 10 dBm output power, measured in
a 12.5 kHz bandwidth at 25 kHz offset, 9.6 kbps PN9 data,
2.4 kHz frequency deviation, compliant with ETSI EN 300 220
25 kHz Channel Spacing
OCCUPIED BANDWIDTH
99.0% of total mean power; 12.5 kHz channel spacing (2.4 kbps
PN9 data, 1.2 kHz frequency deviation); 25 kHz channel spacing
(9.6 kbps PN9 data, 2.4 kHz frequency deviation)
2FSK Gaussian Data Filtering
12.5 kHz Channel Spacing
25 kHz Channel Spacing
2FSK Raised Cosine Data Filtering
12.5 kHz Channel Spacing
25 kHz Channel Spacing
3FSK Raised Cosine Filtering
12.5 kHz Channel Spacing
25 kHz Channel Spacing
4FSK Raised Cosine Filtering
25 kHz Channel Spacing
3.9
9.9
kHz
kHz
4.4
10.2
kHz
kHz
3.9
9.5
kHz
kHz
19.2 kbps PN9 data, 1.2 kHz frequency deviation
kHz
13.2
Rev. 0 | Page 5 of 64
ADF7021-N
Parameter
Min
Typ
Max
Unit Test Conditions/Comments
SPURIOUS EMISSIONS
Reference Spurs
HARMONICS4
−65
dBc
100 kHz loop bandwidth
13 dBm output power, unfiltered conductive/filtered conductive
Second Harmonic
Third Harmonic
All Other Harmonics
OPTIMUM PA LOAD IMPEDANCE5
fRF = 915 MHz
fRF = 868 MHz
fRF = 450 MHz
fRF = 426 MHz
fRF = 315 MHz
−35/−52
−43/−60
−36/−65
dBc
dBc
dBc
39 + j61
48 + j54
98 + j65
100 + j65
129 + j63
173 + j49
Ω
Ω
Ω
Ω
Ω
Ω
fRF = 175 MHz
1 Using Gaussian or raised cosine filtering. The frequency deviation should be chosen to ensure that the transmit-occupied signal bandwidth is within the receiver
IF filter bandwidth.
2 For the definition of frequency deviation, refer to the Register 2—Transmit Modulation Register section.
3 Measured as maximum unmodulated power.
4 Conductive filtered harmonic emissions measured on the EVAL-ADF7021-NDBxx, which includes a T-stage harmonic filter (two inductors and one capacitor).
5 For matching details, refer to the LNA/PA Matching section.
RECEIVER SPECIFICATIONS
Table 3.
Parameter
Min
Typ
Max
Unit
Test Conditions/Comments
Bit error rate (BER) = 10−3, low noise amplifier (LNA)
and power amplifier (PA) matched separately
SENSITIVITY
2FSK
Sensitivity at 0.1 kbps
−130
−127
−122
−115
dBm
dBm
dBm
dBm
fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz
fDEV = 4 kHz, high sensitivity mode, IF_FILTER_BW =
18.5 kHz
Sensitivity at 0.25 kbps
Sensitivity at 1 kbps
Sensitivity at 9.6 kbps
Gaussian 2FSK
Sensitivity at 0.1 kbps
−129
−127
−121
−114
dBm
dBm
dBm
dBm
fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz
fDEV = 4 kHz, high sensitivity mode, IF_FILTER_BW =
18.5 kHz
Sensitivity at 0.25 kbps
Sensitivity at 1 kbps
Sensitivity at 9.6 kbps
GMSK
Sensitivity at 9.6 kbps
−113
dBm
fDEV = 2.4 kHz, high sensitivity mode, IF_FILTER_BW =
18.5 kHz
Raised Cosine 2FSK
Sensitivity at 0.25 kbps
−127
−121
−114
dBm
dBm
dBm
fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz
fDEV = 1 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz
fDEV = 4 kHz, high sensitivity mode, IF_FILTER_BW =
18.5 kHz
Sensitivity at 1 kbps
Sensitivity at 9.6 kbps
Rev. 0 | Page 6 of 64
ADF7021-N
Parameter
Min
Typ
Max
Unit
Test Conditions/Comments
3FSK
Sensitivity at 9.6 kbps
−110
dBm
fDEV = 2.4 kHz, high sensitivity mode, IF_FILTER_BW =
18.5 kHz, Viterbi detection on
Raised Cosine 3FSK
Sensitivity at 9.6 kbps
−110
−112
−109
dBm
dBm
dBm
fDEV = 2.4 kHz, high sensitivity mode, IF_FILTER_BW =
13.5 kHz, alpha = 0.5, Viterbi detection on
4FSK
Sensitivity at 9.6 kbps
fDEV (inner) = 1.2 kHz, high sensitivity mode,
IF_FILTER_BW = 13.5 kHz
Raised Cosine 4FSK
Sensitivity at 9.6 kbps
fDEV (inner) = 1.2 kHz, high sensitivity mode,
IF_FILTER_BW = 13.5 kHz, alpha = 0.5
INPUT IP3
Two-tone test, fLO = 860 MHz, F1 = fLO + 100 kHz,
F2 = fLO − 800 kHz
Low Gain Enhanced Linearity
Mode
−3
dBm
LNA_GAIN = 3, MIXER_LINEARITY = 1
Medium Gain Mode
High Sensitivity Mode
ADJACENT CHANNEL REJECTION
868 MHz
−13.5
−24
dBm
dBm
LNA_GAIN = 10, MIXER_LINEARITY = 0
LNA_GAIN = 30, MIXER_LINEARITY = 0
Wanted signal is 3 dB above the sensitivity point
(BER = 10−3); unmodulated interferer is at the center
of the adjacent channel; rejection measured as the
difference between the interferer level and the
wanted signal level in dB
12.5 kHz Channel Spacing
25 kHz Channel Spacing
426 MHz
40
39
dB
dB
9 kHz IF_FILTER_BW
18.5 kHz IF_FILTER_BW
Wanted signal is 3 dB above the reference sensitivity
point (BER = 10−2); modulated interferer (same
modulation as wanted signal) at the center of the
adjacent channel; rejection measured as the
difference between the interferer level and reference
sensitivity level in dB
12.5 kHz Channel Spacing
25 kHz Channel Spacing
40
39
dB
dB
9 kHz IF_FILTER_BW, compliant with ARIB STD-T67
18.5 kHz IF_FILTER_BW, compliant with ARIB STD-T67
CO-CHANNEL REJECTION
Wanted signal (2FSK, 9.6 kbps, 4 kHz deviation) is
3 dB above the sensitivity point (BER = 10−3), modu-
lated interferer
868 MHz
−5
dB
IMAGE CHANNEL REJECTION
Wanted signal (2FSK, 9.6 kbps, 4 kHz deviation) is
10 dB above the sensitivity point (BER = 10−3); modu-
lated interferer (2FSK, 9.6 kbps, 4 kHz deviation) is
placed at the image frequency of fRF − 200 kHz; the
interferer level is increased until BER = 10−3
868 MHz
450 MHz, Internal Inductor
VCO
26/39
29/50
dB
dB
Uncalibrated/calibrated1, VDD = 3.0 V, TA = 25°C
Uncalibrated/calibrated1, VDD = 3.0 V, TA = 25°C
BLOCKING
Wanted signal is 10 dB above the input sensitivity
level; CW interferer level is increased until BER = 10−3
1 MHz
2 MHz
5 MHz
10 MHz
69
75
78
78.5
12
dB
dB
dB
dB
SATURATION
dBm
2FSK mode, BER = 10−3
(MAXIMUM INPUT LEVEL)
Rev. 0 | Page 7 of 64
ADF7021-N
Parameter
RSSI
Min
Typ
Max
Unit
Test Conditions/Comments
Range at Input2
−120 to −47
2
3
dBm
dB
dB
Linearity
Input power range = −100 dBm to −47 dBm
Input power range = −100 dBm to −47 dBm
See the RSSI/AGC section
Absolute Accuracy
Response Time
AFC
390
μs
Pull-In Range
0.5
1.5 × IF_
FILTER_BW
kHz
The range is programmable in Register 10
(R10_DB[24:31])
Response Time
Accuracy
64
0.5
Bits
kHz
Input power range = −100 dBm to +12 dBm
Rx SPURIOUS EMISSIONS3
Internal Inductor VCO
−91/−91
−52/−70
−62/−72
−64/−85
dBm
dBm
dBm
dBm
<1 GHz at antenna input, unfiltered conductive/filtered
conductive
>1 GHz at antenna input, unfiltered conductive/filtered
conductive
<1 GHz at antenna input, unfiltered conductive/filtered
conductive
>1 GHz at antenna input, unfiltered conductive/filtered
conductive
External Inductor VCO
LNA INPUT IMPEDANCE
fRF = 915 MHz
fRF = 868 MHz
fRF = 450 MHz
fRF = 426 MHz
RFIN to RFGND
24 − j60
26 − j63
63 − j129
68 − j134
96 − j160
178 − j190
Ω
Ω
Ω
Ω
Ω
Ω
fRF = 315 MHz
fRF = 175 MHz
1 Calibration of the image rejection used an external RF source.
2 For received signal levels < −100 dBm, it is recommended to average the RSSI readback value over a number of samples to improve the RSSI accuracy at low input powers.
3 Filtered conductive receive spurious emissions are measured on the EVAL-ADF7021-NDBxx, which includes a T-stage harmonic filter (two inductors and one
capacitor).
Rev. 0 | Page 8 of 64
ADF7021-N
DIGITAL SPECIFICATIONS
Table 4.
Parameter
Min
Typ
Max
Unit Test Conditions/Comments
TIMING INFORMATION
Chip Enabled to Regulator Ready
Chip Enabled to Tx Mode
TCXO Reference
10
μs
CREG (1:4) = 100 nF
32-bit register write time = 50 μs
1
2
ms
ms
XTAL
Chip Enabled to Rx Mode
32-bit register write time = 50 μs, IF filter coarse
calibration only
TCXO Reference
XTAL
1.2
2.2
ms
ms
Tx-to-Rx Turnaround Time
390 μs + (5 × tBIT)
Time to synchronized data out, includes AGC
settling (three AGC levels)and CDR synchronization;
see the AGC Information and Timing section for
more details; tBIT = data bit period
LOGIC INPUTS
Input High Voltage, VINH
Input Low Voltage, VINL
Input Current, IINH/IINL
Input Capacitance, CIN
Control Clock Input
LOGIC OUTPUTS
0.7 × VDD
V
V
μA
pF
MHz
0.2 × VDD
1
10
50
Output High Voltage, VOH
Output Low Voltage, VOL
CLKOUT Rise/Fall
DVDD − 0.4
V
V
ns
pF
IOH = 500 μA
IOL = 500 μA
0.4
5
10
CLKOUT Load
Rev. 0 | Page 9 of 64
ADF7021-N
GENERAL SPECIFICATIONS
Table 5.
Parameter
Min
Typ
Max
Unit
Test Conditions/Comments
TEMPERATURE RANGE (TA)
POWER SUPPLIES
Voltage Supply, VDD
TRANSMIT CURRENT CONSUMPTION1
868 MHz
−40
+85
°C
2.3
3.6
V
All VDD pins must be tied together
VDD = 3.0 V, PA is matched into 50 Ω
VCO_BIAS = 8
0 dBm
5 dBm
10 dBm
20.2
24.7
32.3
mA
mA
mA
450 MHz, Internal Inductor VCO
0 dBm
5 dBm
VCO_BIAS = 8
VCO_BIAS = 2
19.9
23.2
29.2
mA
mA
mA
10 dBm
426 MHz, External Inductor VCO
0 dBm
5 dBm
13.5
17
23.3
mA
mA
mA
10 dBm
RECEIVE CURRENT CONSUMPTION
868 MHz
VDD = 3.0 V
VCO_BIAS = 8
Low Current Mode
High Sensitivity Mode
433MHz, Internal Inductor VCO
Low Current Mode
High Sensitivity Mode
426 MHz, External Inductor VCO
Low Current Mode
High Sensitivity Mode
POWER-DOWN CURRENT CONSUMPTION
Low Power Sleep Mode
22.7
24.6
mA
mA
VCO_BIAS = 8
VCO_BIAS = 2
24.5
26.4
mA
mA
17.5
19.5
mA
mA
0.1
1
μA
CE low
1 The transmit current consumption tests used the same combined PA and LNA matching network as that used on the EVAL-ADF7021-NDBxx evaluation boards.
Improved PA efficiency is achieved by using a separate PA matching network.
Rev. 0 | Page 10 of 64
ADF7021-N
TIMING CHARACTERISTICS
VDD = 3 V 10%, DGND = AGND = 0 V, TA = 25ꢀC, unless otherwise noted. Guaranteed by design but not production tested.
Table 6.
Parameter
Limit at TMIN to TMAX
Unit
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
ns
μs
μs
Test Conditions/Comments
SDATA to SCLK setup time
SDATA to SCLK hold time
SCLK high duration
SCLK low duration
t1
t2
t3
t4
t5
t6
t8
t9
t10
t11
t12
t13
t14
t15
>10
>10
>25
>25
>10
>20
<25
<25
SCLK to SLE setup time
SLE pulse width
SCLK to SREAD data valid, readback
SREAD hold time after SCLK, readback
SCLK to SLE disable time, readback
TxRxCLK negative edge to SLE
TxRxDATA to TxRxCLK setup time (Tx mode)
TxRxCLK to TxRxDATA hold time (Tx mode)
TxRxCLK negative edge to SLE
SLE positive edge to positive edge of TxRxCLK
>10
5 < t11 < (¼ × tBIT)
>5
>5
>¼ × tBIT
>¼ × tBIT
Rev. 0 | Page 11 of 64
ADF7021-N
TIMING DIAGRAMS
Serial Interface
t3
t4
SCLK
t1
t2
DB1
DB0 (LSB)
(CONTROL BIT C1)
SDATA
SLE
DB31 (MSB)
DB30
DB2
(CONTROL BIT C2)
t6
t5
Figure 2. Serial Interface Timing Diagram
t1
t2
SCLK
SDATA
SLE
REG7 DB0
(CONTROL BIT C1)
t3
t10
RV2
RV1
X
X
RV16
RV15
SREAD
t9
t8
Figure 3. Serial Interface Readback Timing Diagram
2FSK/3FSK Timing
±1 × DATA RATE/32
1/DATA RATE
TxRxCLK
TxRxDATA
DATA
Figure 4. TxRxDATA/TxRxCLK Timing Diagram in Receive Mode
1/DATA RATE
TxRxCLK
TxRxDATA
DATA
FETCH
SAMPLE
Figure 5. TxRxDATA/TxRxCLK Timing Diagram in Transmit Mode
Rev. 0 | Page 12 of 64
ADF7021-N
4FSK Timing
In 4FSK receive mode, MSB/LSB synchronization should be guaranteed by SWD in the receive bit stream.
REGISTER 0 WRITE
SWITCH FROM Rx TO Tx
tSYMBOL
t13
t12
t11
tBIT
SLE
TxRxCLK
Rx SYMBOL
LSB
Rx SYMBOL
LSB
Tx SYMBOL
MSB
Rx SYMBOL
MSB
Rx SYMBOL
MSB
Tx SYMBOL
LSB
Tx SYMBOL
MSB
TxRxDATA
Tx/Rx MODE
Rx MODE
Tx MODE
Figure 6. Receive-to-Transmit Timing Diagram in 4FSK Mode
REGISTER 0 WRITE
SWITCH FROM Tx TO Rx
t15
tSYMBOL
t14
tBIT
SLE
TxRxCLK
Tx SYMBOL
LSB
Tx SYMBOL
LSB
Tx SYMBOL
MSB
Tx SYMBOL
MSB
Rx SYMBOL
MSB
Rx SYMBOL
LSB
TxRxDATA
Tx/Rx MODE
Tx MODE
Rx MODE
Figure 7. Transmit-to-Receive Timing Diagram in 4FSK Mode
Rev. 0 | Page 13 of 64
ADF7021-N
UART/SPI Mode
UART mode is enabled by setting R0_DB28 to 1. SPI mode is enabled by setting R0_DB28 to 1 and setting R15_DB[17:19] to 0x7.
The transmit/receive data clock is available on the CLKOUT pin.
tBIT
CLKOUT
(TRANSMIT/RECEIVE DATA
CLOCK IN SPI MODE.
NOT USED IN UART MODE.)
FETCH SAMPLE
TxRxCLK
(TRANSMIT DATA INPUT
IN UART/SPI MODE.)
Tx BIT
Tx BIT
Tx BIT
Tx BIT
Tx BIT
TxRxDATA
(RECEIVE DATA OUTPUT
IN UART/SPI MODE.)
HIGH-Z
Tx/Rx MODE
Tx MODE
Figure 8. Transmit Timing Diagram in UART/SPI Mode
tBIT
CLKOUT
(TRANSMIT/RECEIVE DATA
CLOCK IN SPI MODE.
FETCH SAMPLE
NOT USED IN UART MODE.)
TxRxCLK
(TRANSMIT DATA INPUT
IN UART/SPI MODE.)
HIGH-Z
TxRxDATA
(RECEIVE DATA OUTPUT
IN UART/SPI MODE.)
Rx BIT
Rx BIT
Rx BIT
Rx BIT
Rx BIT
Tx/Rx MODE
Rx MODE
Figure 9. Receive Timing Diagram in UART/SPI Mode
Rev. 0 | Page 14 of 64
ADF7021-N
ABSOLUTE MAXIMUM RATINGS
TA = 25ꢀC, unless otherwise noted.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Table 7.
Parameter
VDD to GND1
Rating
−0.3 V to +5 V
Analog I/O Voltage to GND
Digital I/O Voltage to GND
Operating Temperature Range
Industrial (B Version)
Storage Temperature Range
Maximum Junction Temperature
MLF θJA Thermal Impedance
Reflow Soldering
−0.3 V to AVDD + 0.3 V
−0.3 V to DVDD + 0.3 V
This device is a high performance RF integrated circuit with an
ESD rating of <2 kV and it is ESD sensitive. Proper precautions
should be taken for handling and assembly.
−40°C to +85°C
−65°C to +125°C
150°C
26°C/W
ESD CAUTION
Peak Temperature
Time at Peak Temperature
260°C
40 sec
1 GND = CPGND = RFGND = DGND = AGND = 0.
Rev. 0 | Page 15 of 64
ADF7021-N
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
1
36
35
34
33
32
31
30
29
28
27
26
25
CLKOUT
TxRxCLK
TxRxDATA
SWD
VCOIN
2
CREG1
3
VDD1
4
RFOUT
5
VDD2
RFGND
ADF7021-N
TOP VIEW
(Not to Scale)
6
7
RFIN
CREG2
ADCIN
GND2
RFINB
8
R
LNA
9
SCLK
VDD4
RSET
10
11
12
SREAD
SDATA
SLE
CREG4
GND4
Figure 10. Pin Configuration
Table 8. Pin Function Descriptions
Pin No.
Mnemonic
Description
1
VCOIN
The tuning voltage on this pin determines the output frequency of the voltage controlled oscillator (VCO).
The higher the tuning voltage, the higher the output frequency.
2
3
4
CREG1
VDD1
Regulator Voltage for PA Block. Place a series 3.9 Ω resistor and a 100 nF capacitor between this pin and
ground for regulator stability and noise rejection.
Voltage Supply for PA Block. Place decoupling capacitors of 0.1 μF and 100 pF as close as possible to this pin.
Tie all VDD pins together.
The modulated signal is available at this pin. Output power levels are from −16 dBm to +13 dBm. The output
should be impedance matched to the desired load using suitable components (see the Transmitter section).
RFOUT
5
6
RFGND
RFIN
Ground for Output Stage of Transmitter. All GND pins should be tied together.
LNA Input for Receiver Section. Input matching is required between the antenna and the differential LNA
input to ensure maximum power transfer (see the LNA/PA Matching section).
7
8
RFINB
RLNA
Complementary LNA Input. (See the LNA/PA Matching section.)
External Bias Resistor for LNA. Optimum resistor is 1.1 kΩ with 5% tolerance.
9
10
11
VDD4
RSET
CREG4
Voltage Supply for LNA/MIXER Block. This pin should be decoupled to ground with a 10 nF capacitor.
External Resistor. Sets charge pump current and some internal bias currents. Use a 3.6 kΩ resistor with 5% tolerance.
Regulator Voltage for LNA/MIXER Block. Place a 100 nF capacitor between this pin and GND for regulator
stability and noise rejection.
12, 19, 22 GND4
13 to 18
MIX_I, MIX_I,
Ground for LNA/MIXER Block.
Signal Chain Test Pins. These pins are high impedance under normal conditions and should be left unconnected.
MIX_Q, MIX_Q,
FILT_I, FILT_I
20, 21, 23 FILT_Q, FILT_Q, Signal Chain Test Pins. These pins are high impedance under normal conditions and should be left unconnected.
TEST_A
24
25
26
27
28
CE
Chip Enable. Bringing CE low puts the ADF7021-N into complete power-down. Register values are lost when
CE is low, and the part must be reprogrammed after CE is brought high.
Load Enable, CMOS Input. When SLE goes high, the data stored in the shift registers is loaded into one of the
four latches. A latch is selected using the control bits.
Serial Data Input. The serial data is loaded MSB first with the four LSBs as the control bits. This pin is a high
impedance CMOS input.
Serial Data Output. This pin is used to feed readback data from the ADF7021-N to the microcontroller. The
SCLK input is used to clock each readback bit (for example, AFC or ADC) from the SREAD pin.
SLE
SDATA
SREAD
SCLK
Serial Clock Input. This serial clock is used to clock in the serial data to the registers. The data is latched into
the 32-bit shift register on the CLK rising edge. This pin is a digital CMOS input.
Rev. 0 | Page 16 of 64
ADF7021-N
Pin No.
29
Mnemonic
GND2
Description
Ground for Digital Section.
30
ADCIN
Analog-to-Digital Converter Input. The internal 7-bit ADC can be accessed through this pin. Full scale is 0 V to
1.9 V. Readback is made using the SREAD pin.
31
CREG2
Regulator Voltage for Digital Block. Place a 100 nF capacitor between this pin and ground for regulator
stability and noise rejection.
32
33
VDD2
SWD
Voltage Supply for Digital Block. Place a decoupling capacitor of 10 nF as close as possible to this pin.
Sync Word Detect. The ADF7021-N asserts this pin when it has found a match for the sync word sequence
(see the Register 11—Sync Word Detect Register section). This provides an interrupt for an external
microcontroller indicating that valid data is being received.
34
35
TxRxDATA
TxRxCLK
Transmit Data Input/Received Data Output. This is a digital pin, and normal CMOS levels apply. In UART/SPI
mode, this pin provides an output for the received data in receive mode. In transmit UART/SPI mode, this pin
is high impedance (see the Interfacing to a Microcontroller/DSP section).
Outputs the data clock in both receive and transmit modes. This is a digital pin, and normal CMOS levels
apply. The positive clock edge is matched to the center of the received data. In transmit mode, this pin
outputs an accurate clock to latch the data from the microcontroller into the transmit section at the exact
required data rate. In UART/SPI mode, this pin is used to input the transmit data in transmit mode. In receive
UART/SPI mode, this pin is high impedance (see the Interfacing to a Microcontroller/DSP section).
36
37
CLKOUT
A divided-down version of the crystal reference with output driver. The digital clock output can be used to drive
several other CMOS inputs such as a microcontroller clock. The output has a 50:50 mark-space ratio and is inverted
with respect to the reference. Place a series 1 kΩ resistor as close as possible to the pin in applications where the
CLKOUT feature is being used.
Provides the DIGITAL_LOCK_DETECT signal. This signal is used to determine if the PLL is locked to the correct
frequency. It also provides other signals such as REGULATOR_READY, which is an indicator of the status of the
serial interface regulator (see the MUXOUT section for more information).
MUXOUT
38
39
OSC2
OSC1
Connect the reference crystal between this pin and OSC1. A TCXO reference can be used by driving this pin
with CMOS levels and disabling the internal crystal oscillator.
Connect the reference crystal between this pin and OSC2. A TCXO reference can be used by driving this pin
with ac-coupled 0.8 V p-p levels and by enabling the internal crystal oscillator.
40
41
VDD3
CREG3
Voltage Supply for the Charge Pump and PLL Dividers. Decouple this pin to ground with a 10 nF capacitor.
Regulator Voltage for Charge Pump and PLL Dividers. Place a 100 nF capacitor between this pin and ground
for regulator stability and noise rejection.
42
CPOUT
Charge Pump Output. This output generates current pulses that are integrated in the loop filter. The
integrated current changes the control voltage on the input to the VCO.
43
VDD
Voltage Supply for VCO Tank Circuit. Decouple this pin to ground with a 10 nF capacitor.
44, 46
L2, L1
External VCO Inductor Pins. If using an external VCO inductor, connect a chip inductor across these pins to set
the VCO operating frequency. If using the internal VCO inductor, these pins can be left floating. See the
Voltage Controlled Oscillator (VCO) section for more information.
45, 47
48
GND, GND1
CVCO
Grounds for VCO Block.
Place a 22 nF capacitor between this pin and CREG1 to reduce VCO noise.
Rev. 0 | Page 17 of 64
ADF7021-N
TYPICAL PERFORMANCE CHARACTERISTICS
–70
DR = 9.6kbps
RF FREQ = 900MHz
DATA = PRBS9
fDEV = 2.4kHz
RF FREQ = 869.5MHz
V
= 2.3V
DD
–80
–90
TEMPERATURE = 25°C
VCO_BIAS = 8
VCO_ADJUST = 3
I
= 0.8mA
CP
I
= 1.4mA
CP
2FSK
–100
–110
–120
–130
–140
–150
GFSK
I
= 2.2mA
CP
CENTER 869.5 25MHz
RES BW 300Hz
SPAN 50kHz
1
10
100
1000
10000
VBW 300Hz SWEEP 2.118s (601pts)
FREQUENCY OFFSET (kHz)
Figure 11. Phase Noise Response at 900 MHz, VDD = 2.3 V
Figure 14. Output Spectrum in 2FSK and GFSK Modes
16
12
DR = 9.6kbps
PA_BIAS = 11µA
PA_BIAS = 9µA
DATA = PRBS9
fDEV = 2.4kHz
RF FREQ = 869.5MHz
8
4
0
PA_BIAS = 5µA
–4
2FSK
PA_BIAS = 7µA
–8
–12
–16
–20
–24
–28
–32
–36
–40
RC2FSK
CENTER 869.5 25MHz
RES BW 300Hz
SPAN 50kHz
0
4
8
12 16 20 24 28 32 36 40 44 48 52 56 60
PA SETTING
VBW 300Hz SWEEP 2.118s (601pts)
Figure 12. RF Output Power vs. PA Setting
Figure 15. Output Spectrum in 2FSK and Raised Cosine 2FSK Modes
1R
RF FREQ = 440MHz
SR = 4.8ksym/s
DATA = PRBS9
fDEV = 2.4kHz
OUTPUT POWER = 10dBm
FILTER = T-STAGE LC FILTER
MARKER Δ = 52.2dB
RF FREQ = 869.5MHz
4FSK
1
RC4FSK
START 300MHz
RES BW 100Hz
STOP 3.5GHz
VBW 100Hz SWEEP 385.8ms (601pts)
CENTER 869.493 8MHz
RES BW 300Hz
SPAN 100kHz
VBW 300Hz SWEEP 4.237s (601pts)
Figure 13. PA Output Harmonic Response with T-Stage LC Filter
Figure 16. Output Spectrum in 4FSK and Raised Cosine 4FSK Modes
Rev. 0 | Page 18 of 64
ADF7021-N
REF 15dBm
0
–1
–2
–3
–4
–5
–6
–7
–8
ATTEN 25dB
SAMP LOG 10dB/
DATA RATE = 1kbps
fDEV = 1kHz
DR = 9.6kbps
DATA = PRS9
fDEV = 2.4kHz
RF FREQ = 135MHz
IF BW = 12.5kHz
RF FREQ = 869.5MHz
3.0V, +25°C
2.3V, +85°C
VAVG 100
3FSK
V1 V2
S3 FC
3.6V, –40°C
RC3FSK
–130 –128 –126 –124 –122 –120 –118 –116 –114 –112 –110 –108
RF INPUT POWER (dBm)
CENTER 869.5MHz
RES BW 300Hz
VBW 300Hz
SPAN 50kHz
SWEEP2.226s (401pts)
Figure 17. Output Spectrum in 3FSK and Raised Cosine 3FSK Modes
Figure 20. 2FSK Sensitivity vs. VDD and Temperature, fRF = 135 MHz
0
RAMP RATE:
CW ONLY
256 CODES/BIT
128 CODES/BIT
64 CODES/BIT
32 CODES/BIT
TRACE = MAX HOLD
3FSK MODULATION
DATA RATE = 9.6kbps
10
0
PA ON/OFF RATE = 3Hz
PA ON/OFF CYCLES = 10,000
fDEV = 2.4kHz
–1
–2
–3
–4
–5
–6
–7
–8
V
= 3.0V
DD
MOD INDEX = 0.5
RF FREQ = 440 MHz
–10
–20
–30
–40
–50
–60
2.3V +25°C
3.0V +25°C
3.6V +25°C
2.3V –40°C
3.0V –40°C
3.6V –40°C
2.3V +85°C
3.0V +85°C
3.6V +85°C
–120
–115
–110
–105
–100
–95
–100
–50
0
50
100
RF INPUT POWER (dBm)
FREQUENCY OFFSET (kHz)
Figure 18. Output Spectrum in Maximum Hold
for Various PA Ramp Rate Options
Figure 21. 3FSK Sensitivity vs. VDD and Temperature, fRF = 440 MHz
0
–1
–2
–3
–4
–5
–6
–7
–8
0
DATA RATE = 9.6kbps
fDEV = 4kHz
DATA RATE = 19.6kbps
SYMBOL RATE = 9.8ksym/s
fDEV (inner) = 2.4kHz
–1
RF FREQ = 868MHz
IF BW = 25kHz
MOD INDEX = 0.5
RF FREQ = 420MHz
–2
–3
–4
–5
–6
–7
–8
IF BW = 12.5kHz
3.0V, +25°C
2.3V, +85°C
3.6V, –40°C
2.3V +25°C
3.0V +25°C
3.6V +25°C
2.3V –40°C
3.0V –40°C
3.6V –40°C
2.3V +85°C
3.0V +85°C
3.6V +85°C
–122 –120 –118 –116 –114 –112 –110 –108 –106 –104
–120
–115
–110
–105
–100
–95
RF INPUT POWER (dBm)
RF INPUT POWER (dBm)
Figure 19. 2FSK Sensitivity vs. VDD and Temperature, fRF = 868 MHz
Figure 22. 4FSK Sensitivity vs. VDD and Temperature, fRF = 420 MHz
Rev. 0 | Page 19 of 64
ADF7021-N
90
80
70
60
50
40
30
20
10
0
2.5
0
+90°C
–2.5
–5.0
–7.5
–10.0
–12.5
–15.0
–17.5
–20.0
–22.5
–25.0
–27.5
–30.0
–32.5
–35.0
–37.5
RF FREQ = 868MHz
WANTED SIGNAL
(10dB ABOVE SENSITIVITY
POINT) = 2FSK,
–40°C
fDEV = 4kHz,
DATA RATE = 9.8kbps
BLOCKER = 2FSK,
fDEV = 4kHz,
DATA RATE = 9.8kbps
V
= 3.0V
DD
TEMPERATURE = 25°C
–10
–22 –18 –14 –10 –6
–2 0
2
6
10 14 18
22
90
92
94
96
98
100 102 104 106 108 110
FREQUENCY OFFSET (MHz)
IF FREQUENCY (kHz)
Figure 23. Wideband Interference Rejection
Figure 26. Variation of IF Filter Response with Temperature
(IF_FILTER_BW = 9 kHz, Temperature Range is −40°C to +90°C in 10° Steps)
–20
–40
–100
RF FREQ = 860MHz
2FSK MODULATION
DATA RATE = 9.6kbps
RSSI
READBACK LEVEL
–102
IF BW = 25kHz
= 3.0V
–104
V
DD
TEMPERATURE = 25°C
–60
–106
–108
–110
–112
–114
–116
–118
–80
DISCRIMINATOR BANDWIDTH =
2× FSK FREQUENCY DEVIATION
–100
–120
–140
ACTUAL RF INPUT LEVEL
DISCRIMINATOR BANDWIDTH =
1× FSK FREQUENCY DEVIATION
–122.5 –112.5 –102.5 –92.5 –82.5 –72.5 –62.5 –52.5 –42.5
RF INPUT (dBm)
0
0.2 0.4 0.6
0.8
1.0
1.2
MODULATION INDEX
Figure 27. 2FSK Sensitivity vs. Modulation Index vs. Correlator
Discriminator Bandwidth
Figure 24. Digital RSSI Readback Linearity
70
0
RF FREQ = 430MHz
CALIBRATED
EXTERNAL VCO INDUCTOR
DATA RATE = 9.6kbps
60
50
40
30
20
10
0
–1
TEMPERATURE = 25°C, V = 3.0V
DD
THRESHOLD DETECTION
–2
VITERBI DETECTION
–3
UNCALIBRATED
–4
–5
3FSK MODULATION
DD
V
= 3.0V, TEMP = 25°C
DATA RATE = 9.6kbps
–6
fDEV = 2.4kHz
RF FREQ = 868MHz
IF BW = 18.75kHz
–10
–7
429.80 429.85 429.90 429.95 430.00 430.05 430.10 430.15 430.20
–120 –118 –116 –114 –112 –110 –108 –106 –104 –102 –100
INPUT POWER (dBm)
RF FREQUENCY (MHz)
Figure 28. 3FSK Receiver Sensitivity Using Viterbi Detection and
Threshold Detection
Figure 25. Image Rejection, Uncalibrated vs. Calibrated
Rev. 0 | Page 20 of 64
ADF7021-N
–70
–80
MODULATION = 2FSK
DATA RATE = 9.6kbps
fDEV = 4kHz
+3
+1
HIGH MIXER
LINEARITY
IF BW = 12.5kHz
DEMOD = CORRELATOR
SENSITIVITY @ 1E-3 BER
IP3= –5dBm
IP3 = –3dBm
–90
0
–100
–110
–120
–130
IP3 = –9dBm
–1
IP3 = –20dBm
IP3 = –24dBm
DEFAULT
MIXER
LINEARITY
IP3 = –13.5dBm
–3
RF I/P LEVEL = –70dBm
DATA RATE = 9.7kbps
fDEV (inner) = 1.2kHz
IF BW = 25kHz
POST DEMOD BW = 12.4kHz
22452 ACQS
M 50µs
3, 72
(LOW GAIN MODE)
10, 72
30, 72
(HIGH GAIN MODE)
(MEDIUM GAIN MODE)
LNA GAIN, FILTER GAIN
Figure 31. Receive Sensitivity vs. LNA/IF Filter Gain and Mixer Linearity Settings
(The input IP3 at each setting is also shown)
Figure 29. 4FSK Receiver Eye Diagram Measured Using the Test DAC Output
+1
0
–1
RF I/P LEVEL = –70dBm
DATA RATE = 10kbps
fDEV = 2.5kHz
IF BW = 12.5kHz
POST DEMOD BW = 12.4kHz
4
20834 ACQS
M 20µs
C13
1.7V
Figure 30. 3FSK Receiver Eye Diagram Measured Using the Test DAC Output
Rev. 0 | Page 21 of 64
ADF7021-N
FREQUENCY SYNTHESIZER
REFERENCE INPUT
CLKOUT Divider and Buffer
The CLKOUT circuit takes the reference clock signal from the
oscillator section, shown in Figure 32, and supplies a divided-
down, 50:50 mark-space signal to the CLKOUT pin. The CLKOUT
signal is inverted with respect to the reference clock. An even
divide from 2 to 30 is available. This divide number is set in
R1_DB[7:10]. On power-up, the CLKOUT defaults to divide-by-8.
The on-board crystal oscillator circuitry (see Figure 32) can use
a quartz crystal as the PLL reference. Using a quartz crystal with
a frequency tolerance of ≤10 ppm for narrow-band applications
is recommended. It is possible to use a quartz crystal with >10 ppm
tolerance, but to comply with the absolute frequency error
specifications of narrow-band regulations (for example, ARIB
STD-T67 and ETSI EN 300 220), compensation for the
frequency error of the crystal is necessary.
DV
DD
CLKOUT
ENABLE BIT
The oscillator circuit is enabled by setting R1_DB12 high. It is
enabled by default on power-up and is disabled by bringing CE
low. Errors in the crystal can be corrected by using the automatic
frequency control feature or by adjusting the fractional-N value
(see the N Counter section).
DIVIDER
1 TO 15
OSC1
÷2
CLKOUT
Figure 33. CLKOUT Stage
To disable CLKOUT, set the divide number to 0. The output
buffer can drive up to a 20 pF load with a 10% rise time at
4.8 MHz. Faster edges can result in some spurious feedthrough
to the output. A series resistor (1 kΩ) can be used to slow the
clock edges to reduce these spurs at the CLKOUT frequency.
OSC1
OSC2
CP1
CP2
R Counter
Figure 32. Oscillator Circuit on the ADF7021-N
The 3-bit R counter divides the reference input frequency by an
integer between 1 and 7. The divided-down signal is presented
as the reference clock to the phase frequency detector (PFD). The
divide ratio is set in R1_DB[4:6]. Maximizing the PFD frequency
reduces the N value. This reduces the noise multiplied at a rate of
20 log(N) to the output and reduces occurrences of spurious
components.
Two parallel resonant capacitors are required for oscillation at
the correct frequency. Their values are dependent on the crystal
specification. They should be chosen to make sure that the
series value of capacitance added to the PCB track capacitance
adds up to the specified load capacitance of the crystal, usually
12 pF to 20 pF. Track capacitance values vary from 2 pF to 5 pF,
depending on board layout. When possible, choose capacitors
that have a very low temperature coefficient to ensure stable
frequency operation over all conditions.
Register 1 defaults to R = 1 on power-up.
PFD [Hz] = XTAL/R
Using a TCXO Reference
Loop Filter
A single-ended reference (TCXO, VCXO, or OCXO) can also be
used with the ADF7021-N. This is recommended for applications
having absolute frequency accuracy requirements of <10 ppm, such
as applications requiring compliance with ARIB STD-T67 or
ETSI EN 300 220. The following are two options for interfacing
the ADF7021-N to an external reference oscillator.
The loop filter integrates the current pulses from the charge
pump to form a voltage that tunes the output of the VCO to the
desired frequency. It also attenuates spurious levels generated by
the PLL. A typical loop filter design is shown in Figure 34.
CHARGE
VCO
PUMP OUT
•
An oscillator with CMOS output levels can be applied to
OSC2. The internal oscillator circuit should be disabled by
setting R1_DB12 low.
Figure 34. Typical Loop Filter Configuration
•
An oscillator with 0.8 V p-p levels can be ac-coupled through
a 22 pF capacitor into OSC1. The internal oscillator circuit
should be enabled by setting R1_DB12 high.
The loop should be designed so that the loop bandwidth (LBW)
is approximately 100 kHz. This provides a good compromise
between in-band phase noise and out-of-band spurious rejection.
Widening the LBW excessively reduces the time spent jumping
between frequencies, but it can cause insufficient spurious attenua-
tion. Narrow-loop bandwidths can result in the loop taking long
periods to attain lock and can also result in a higher level of power
falling into the adjacent channel. The loop filter design on the
Programmable Crystal Bias Current
Bias current in the oscillator circuit can be configured between 20
μA and 35 μA by writing to the XTAL_BIAS bits (R1_DB [13:14]).
Increasing the bias current allows the crystal oscillator to power
up faster.
Rev. 0 | Page 22 of 64
ADF7021-N
EVAL-ADF7021-NDBxx should be used for optimum
performance.
voltage must be stabilized. Regulator status (CREG4) can be
monitored using the REGULATOR_READY signal from the
MUXOUT pin.
The free design tool ADI SRD Design Studio™ can also
be used to design loop filters for the ADF7021-N (see the ADI
SRD Design Studio web site for details).
MUXOUT
The MUXOUT pin allows access to various digital points in the
ADF7021-N. The state of MUXOUT is controlled in Register 0
(R0_DB[29:31]).
N Counter
The feedback divider in the ADF7021-N PLL consists of an
8-bit integer counter (R0_DB[19:26]) and a 15-bit, sigma-delta
(Σ-Δ) fractional_N divider (R0_DB[4:18]). The integer counter
is the standard pulse-swallow type that is common in PLLs. This
sets the minimum integer divide value to 23. The fractional divide
value provides very fine resolution at the output, where the output
frequency of the PLL is calculated as
REGULATOR_READY
REGULATOR_READY is the default setting on MUXOUT
after the transceiver is powered up. The power-up time of the
regulator is typically 50 μs. Because the serial interface is powered
from the regulator, the regulator must be at its nominal voltage
before the ADF7021-N can be programmed. The status of the
regulator can be monitored at MUXOUT. When the regulator
ready signal on MUXOUT is high, programming of the
ADF7021-N can begin.
Fractional_ N
⎛
⎞
XTAL
R
⎜
⎟
⎟
fOUT
=
× Integer _ N +
215
⎜
⎝
⎠
When RF_DIVIDE_BY_2 (see the Voltage Controlled
Oscillator (VCO) section) is selected, this formula becomes
DV
DD
REGULATOR_READY (DEFAULT)
FILTER_CAL_COMPLETE
DIGITAL_LOCK_DETECT
RSSI_READY
XTAL
R
Fractional _ N
⎛
⎝
⎞
⎟
⎠
fOUT
=
×0.5× Integer_N +
⎜
215
The combination of Integer_N (maximum = 255) and
Fractional_N (maximum = 32,768/32,768) gives a maximum
N divider of 255 + 1. Therefore, the minimum usable PFD is
MUX
CONTROL
MUXOUT
Tx_Rx
LOGIC_ZERO
TRISTATE
Maximum Required Output Frequency
LOGIC_ONE
PFDMIN Hz =
[ ]
(
255 +1
)
For example, when operating in the European 868 MHz to
870 MHz band, PFDMIN = 3.4 MHz.
REFERENCE IN
DGND
Figure 36. MUXOUT Circuit
FILTER_CAL_COMPLETE
PFD/
4\R
CHARGE
PUMP
VCO
MUXOUT can be set to FILTER_CAL_COMPLETE. This signal
goes low for the duration of both a coarse IF filter calibration
and a fine IF filter calibration. It can be used as an interrupt to
a microcontroller to signal the end of the IF filter calibration.
4\N
THIRD-ORDER
Σ-Δ MODULATOR
DIGITAL_LOCK_DETECT
DIGITAL_LOCK_DETECT indicates when the PLL has locked.
The lock detect circuit is located at the PFD. When the phase
error on five consecutive cycles is less than 15 ns, lock detect is
set high. Lock detect remains high until a 25 ns phase error is
detected at the PFD.
FRACTIONAL_N
INTEGER_N
Figure 35. Fractional_N PLL
Voltage Regulators
The ADF7021-N contains four regulators to supply stable
voltages to the part. The nominal regulator voltage is 2.3 V.
Regulator 1 requires a 3.9 Ω resistor and a 100 nF capacitor in
series between CREG1 and GND, whereas the other regulators
require a 100 nF capacitor connected between CREGx and GND.
When CE is high, the regulators and other associated circuitry
are powered on, drawing a total supply current of 2 mA. Bringing
the CE pin low disables the regulators, reduces the supply current
to less than 1 μA, and erases all values held in the registers.
RSSI_READY
MUXOUT can be set to RSSI_READY. This indicates that the
internal analog RSSI has settled and a digital RSSI readback can
be performed.
Tx_Rx
Tx_Rx signifies whether the ADF7021-N is in transmit or receive
mode. When in transmit mode, this signal is low. When in receive
mode, this signal is high. It can be used to control an external
Tx/Rx switch.
The serial interface operates from a regulator supply. Therefore,
to write to the part, the user must have CE high and the regulator
Rev. 0 | Page 23 of 64
ADF7021-N
A plot of the VCO operating frequency vs. total external
inductance (chip inductor + PCB track) is shown in Figure 38.
VOLTAGE CONTROLLED OSCILLATOR (VCO)
The ADF7021-N contains two VCO cores. The first VCO, the
internal inductor VCO, uses an internal LC tank and supports
842 MHz to 916 MHz and 421 MHz to 458 MHz operating
bands. The second VCO, the external inductor VCO, uses an
external inductor as part of its LC tank and supports the RF
operating band of 80 MHz to 650 MHz.
750
700
650
fMAX (MHz)
600
550
500
450
400
350
To minimize spurious emissions, both VCOs operate at twice
the RF frequency. The VCO signal is then divided by 2 inside
the synthesizer loop, giving the required frequency for the
transmitter and the required local oscillator (LO) frequency for
the receiver. A further divide-by-2 (RF_DIVIDE_BY_2) is
performed outside the synthesizer loop to allow operation in
the 421 MHz to 458 MHz band (internal inductor VCO) and
the 80 MHz to 325 MHz band (external inductor VCO).
fMIN (MHz)
300
250
200
0
5
10
15
20
25
30
TOTAL EXTERNAL INDUCTANCE (nH)
The VCO needs an external 22 nF capacitor between the CVCO
pin and the regulator (CREG1 pin) to reduce internal noise.
Figure 38. Direct RF Output vs. Total External Inductance
The inductance for a PCB track using FR4 material is approxi-
mately 0.57 nH/mm. This should be subtracted from the total
value to determine the correct chip inductor value.
VCO_BIAS
R1_DB(19:22)
Typically, a particular inductor value allows the ADF7021-N to
function over a range of 6% of the RF operating frequency.
When the RF_DIVIDE_BY_2 bit (R1_DB18) is selected, this
range becomes 3%. At 400 MHz, for example, an operating
range of 24 MHz (that is, 376 MHz to 424 MHz) with a single
inductor (VCO range centered at 400 MHz) can be expected.
LOOP FILTER
VCO
MUX
÷2
TO PA
÷2
220µF
CVCO PIN
TO
N DIVIDER
RF_DIVIDE_BY_2
R1_DB18
Figure 37. Voltage Controlled Oscillator (VCO)
The VCO tuning voltage can be checked for a particular RF
output frequency by measuring the voltage on the VCOIN pin
when the part is fully powered up in transmit or receive mode.
Internal Inductor VCO
To select the internal inductor VCO, set R1_DB25 to Logic 0,
which is the default setting.
The VCO tuning range is 0.2 V to 2 V. The external inductor
value should be chosen to ensure that the VCO is operating
as close as possible to the center of this tuning range. This is
particularly important for RF frequencies <200 MHz, where
the VCO gain is reduced and a tuning range of < 6 MHz exists.
VCO bias current can be adjusted using R1_DB[19:22]. To
ensure VCO oscillation, the minimum bias current setting under
all conditions when using the internal inductor VCO is 0x8.
The VCO should be recentered, depending on the required
frequency of operation, by programming the VCO_ADJUST
bits (R1_DB[23:24]). This is detailed in Table 9.
The VCO operating frequency range can be adjusted by
programming the VCO_ADJUST bits (R1_DB[23:24]). This
typically allows the VCO operating range to be shifted up or
down by a maximum of 1% of the RF frequency.
External Inductor VCO
When using the external inductor VCO, the center frequency of the
VCO is set by the internal varactor capacitance and the combined
inductance of the external chip inductor, bond wire, and PCB track.
The external inductor is connected between the L2 and L1 pins.
To select the external inductor VCO, set R1_DB25 to Logic 1.
The VCO_BIAS should be set depending on the frequency of
operation (as indicated in Table 9).
Rev. 0 | Page 24 of 64
ADF7021-N
Table 9. RF Output Frequency Ranges for Internal and External Inductor VCOs and Required Register Settings
Register Settings
RF Frequency
Output (MHz)
VCO to
Be Used
RF Divide
by 2
VCO_INDUCTOR
R1_DB25
RF_DIVIDE_BY_2
R1_DB18
VCO_ADJUST
R1_DB[23:24]
VCO_BIAS
R1_DB[19:22]
870 to 916
842 to 870
440 to 458
421 to 440
450 to 650
200 to 450
80 to 200
Internal L
Internal L
Internal L
Internal L
External L
External L
External L
No
No
Yes
Yes
No
No
Yes
0
0
0
0
1
1
1
0
0
1
1
0
0
1
11
00
11
00
XX
XX
XX
8
8
8
8
4
3
2
These spurs are attenuated by the loop filter. They are more
noticeable on channels close to integer multiples of the reference
where the difference frequency may be inside the loop bandwidth;
thus, the name integer boundary spurs. The occurrence of these
spurs is rare because the integer frequencies are around multiples
of the reference, which is typically >10 MHz. To avoid having
very small or very large values in the fractional register, choose
a suitable reference frequency.
CHOOSING CHANNELS FOR BEST SYSTEM
PERFORMANCE
An interaction between the RF VCO frequency and the
reference frequency can lead to fractional spur creation. When
the synthesizer is in fractional mode (that is, the RF VCO and
reference frequencies are not integer related), spurs can appear
on the VCO output spectrum at an offset frequency that
corresponds to the difference frequency between an integer
multiple of the reference and the VCO frequency.
Rev. 0 | Page 25 of 64
ADF7021-N
TRANSMITTER
RF OUTPUT STAGE
1
2
3
4
...
8
... 16
DATA BITS
The power amplifier (PA) of the ADF7021-N is based on a
single-ended, controlled current, open-drain amplifier that has
been designed to deliver up to 13 dBm into a 50 Ω load at a
maximum frequency of 950 MHz.
PA RAMP 0
(NO RAMP)
PA RAMP 1
(256 CODES PER BIT)
PA RAMP 2
(128 CODES PER BIT)
The PA output current and consequently, the output power, are
programmable over a wide range. The PA configuration is shown
in Figure 39. The output power is set using R2_DB[13:18].
R2_DB(11:12)
PA RAMP 3
(64 CODES PER BIT)
PA RAMP 4
(32 CODES PER BIT)
PA RAMP 5
(16 CODES PER BIT)
2
PA RAMP 6
(8 CODES PER BIT)
6
IDAC
R2_DB(13:18)
PA RAMP 7
(4 CODES PER BIT)
RFOUT
RFGND
Figure 40. PA Ramping Settings
R2_DB7
+
R0_DB27
PA Bias Currents
The PA_BIAS bits (R2_DB[11:12]) facilitate an adjustment of
the PA bias current to further extend the output power control
range, if necessary. If this feature is not required, the default
value of 9 μA is recommended. If output power of greater than
10 dBm is required, a PA bias setting of 11 μA is recommended.
The output stage is powered down by resetting R2_DB7.
FROM VCO
Figure 39. PA Configuration
The PA is equipped with overvoltage protection, which makes it
robust in severe mismatch conditions. Depending on the appli-
cation, users can design a matching network for the PA to exhibit
optimum efficiency at the desired radiated output power level
for a wide range of antennas, such as loop or monopole antennas.
See the LNA/PA Matching section for more information.
MODULATION SCHEMES
The ADF7021-N supports 2FSK, 3FSK, and 4FSK modulation.
The implementation of these modulation schemes is shown in
Figure 41.
PA Ramping
TO
PA STAGE
LOOP FILTER
PFD/
CHARGE
PUMP
REF
When the PA is switched on or off quickly, its changing input
impedance momentarily disturbs the VCO output frequency.
This process is called VCO pulling, and it manifests as spectral
splatter or spurs in the output spectrum around the desired carrier
frequency. Some radio emissions regulations place limits on
these PA transient-induced spurs (for example, the ETSI EN 300 220
regulations). By gradually ramping the PA on and off, PA transient
spurs are minimized.
÷2
VCO
÷N
FRACTIONAL_N
THIRD-ORDER
Σ-Δ MODULATOR
INTEGER_N
Tx_FREQUENCY_
DEVIATION
The ADF7021-N has built-in PA ramping configurability. As
Figure 40 illustrates, there are eight ramp rate settings, defined
as a certain number of PA setting codes per one data bit period.
The PA steps through each of its 64 code levels but at different
speeds for each setting. The ramp rate is set by configuring
R2_DB[8:10].
2FSK
TxDATA
GAUSSIAN
2
3FSK
4FSK
PRE-
CODER
1 – D PR
SHAPING
OR
MUX
RAISED COSINE
FILTERING
4FSK
BIT SYMBOL
MAPPER
If the PA is enabled/disabled by the PA_ENABLE bit (R2_DB7),
it ramps up and down. If it is enabled/disabled by the Tx/Rx bit
(R0_DB27), it ramps up and turns hard off.
Figure 41. Transmit Modulation Implementation
Rev. 0 | Page 26 of 64
ADF7021-N
Setting the Transmit Data Rate
3-Level Frequency Shift Keying (3FSK)
In all modulation modes except oversampled 2FSK mode, an
accurate clock is provided on the TxRxCLK pin to latch the data
from the microcontroller into the transmit section at the required
data rate. The exact frequency of this clock is defined by
In 3-level FSK modulation (also known as modified duobinary
FSK), the binary data (Logic 0 and Logic 1) is mapped onto
three distinct frequencies: the carrier frequency (fC), the carrier
frequency minus a deviation frequency (fC − fDEV), and the
carrier frequency plus the deviation frequency (fC + fDEV).
XTAL
DATA CLK =
DEMOD _ CLK _ DIVIDE×CDR _ CLK _ DIVIDE×32
A Logic 0 is mapped to the carrier frequency while a Logic 1 is
either mapped onto the fC − fDEV frequency or the fC + fDEV
frequency.
where:
XTAL is the crystal or TCXO frequency.
DEMOD_CLK_DIVIDE is the divider that sets the demodulator
clock rate (R3_DB[6:9]).
0
+1
–1
CDR_CLK_DIVIDE is the divider that sets the CDR clock rate
(R3_DB[10:17]).
fC
–
fDEV
fC
fC + fDEV
Refer to the Register 3—Transmit/Receive Clock Register
section for more programming information.
RF FREQUENCY
Figure 42. 3FSK Symbol-to-Frequency Mapping
Setting the FSK Transmit Deviation Frequency
Compared to 2FSK, this bits-to-frequency mapping results in a
reduced transmission bandwidth because some energy is removed
from the RF sidebands and transferred to the carrier frequency.
At low modulation index, 3FSK improves the transmit spectral
efficiency by up to 25% when compared to 2FSK.
In all modulation modes, the deviation from the center
frequency is set using the Tx_FREQUENCY_DEVIATION bits
(R2_DB[19:27]).
The deviation from the center frequency in Hz is as follows:
For direct RF output,
Bit-to-symbol mapping for 3FSK is implemented using a linear
convolutional encoder that also permits Viterbi detection to be
used in the receiver. A block diagram of the transmit hardware
used to realize this system is shown in Figure 43. The convolu-
tional encoder polynomial used to implement the transmit
spectral shaping is
PFD×Tx_ FREQUENCY_ DEVIATION
fDEV [Hz] =
216
For RF_DIVIDE_BY_2 enabled,
PFD×Tx_ FREQUENCY_ DEVIATION
fDEV [Hz] = 0.5×
216
P(D) = 1 − D2
where Tx_FREQUENCY_DEVIATION is a number from 1 to
511 (R2_DB[19:27]).
where:
P is the convolutional encoder polynomial.
D is the unit delay operator.
In 4FSK modulation, the four symbols (00, 01, 11, 10) are
transmitted as 3 ꢁ fDEV and 1 ꢁ fDEV
.
A digital precoder with transfer function 1/P(D) implements an
inverse modulo-2 operation of the 1 − D2 shaping filter in the
transmitter.
Binary Frequency Shift Keying (2FSK)
Two-level frequency shift keying is implemented by setting the
N value for the center frequency and then toggling it with the
TxDATA line. The deviation from the center frequency is set
using the Tx_FREQUENCY_DEVIATION bits, R2_DB[19:27].
Tx DATA
0, 1
0, 1
CONVOLUTIONAL
ENCODER
P(D)
PRECODER
1/P(D)
2FSK is selected by setting the MODULATION_SCHEME bits
(R2_DB[4:6]) to 000.
0, +1, –1
Minimum shift keying (MSK) or Gaussian minimum shift
keying (GMSK) is supported by selecting 2FSK modulation and
using a modulation index of 0.5. A modulation index of 0.5 is
set up by configuring R2_DB[19:27] for an fDEV = 0.25 ꢁ
transmit data rate.
fC
fC
fC
FSK MOD
CONTROL
AND
+
–
fDEV
fDEV
TO
N DIVIDER
DATA FILTERING
Figure 43. 3FSK Encoding
Rev. 0 | Page 27 of 64
ADF7021-N
The signal mapping of the input binary transmit data to the
3-level convolutional output is shown in Table 10. The
convolutional encoder restricts the maximum number of
sequential +1s or −1s to two and delivers an equal number of
+1s and −1s to the FSK modulator, thus ensuring equal spectral
energy in both RF sidebands.
The transmit clock from Pin TxRxCLK is available after writing
to Register 3 in the power-up sequence for receive mode. The
MSB of the first symbol should be clocked into the ADF7021-N
on the first transmit clock pulse from the ADF7021-N after
writing to Register 3. Refer to Figure 6 for more timing
information.
Oversampled 2FSK
Table 10. 3-Level Signal Mapping of the Convolutional Encoder
In oversampled 2FSK, there is no data clock from the TxRxCLK
pin. Instead, the transmit data at the TxRxDATA pin is sampled
at 32 times the programmed rate.
1
0
0
0
1
1
0
0
0
0
1
0
1
0
1
0
0
1
0
1
TxDATA
1
0
1
1
0
Precoder Output
Encoder Output
+1
−1 +1
+1
−1
This is the only modulation mode that can be used with the UART
mode interface for data transmission (refer to the Interfacing to
a Microcontroller/DSP section for more information).
Another property of this encoding scheme is that the transmitted
symbol sequence is dc-free, which facilitates symbol detection
and frequency measurement in the receiver. In addition, there is
no code rate loss associated with this 3-level convolutional encoder;
that is, the transmitted symbol rate is equal to the data rate
presented at the transmit data input.
SPECTRAL SHAPING
Gaussian or raised cosine filtering can be used to improve
transmit spectral efficiency. The ADF7021-N supports Gaussian
filtering (bandwidth time [BT] = 0.5) on 2FSK modulation.
Raised cosine filtering can be used with 2FSK, 3FSK, or 4FSK
modulation. The roll-off factor (alpha) of the raised cosine filter
has programmable options of 0.5 and 0.7. Both the Gaussian
and raised cosine filters are implemented using linear phase
digital filter architectures that deliver precise control over the
BT and alpha filter parameters, and guarantee a transmit spectrum
that is very stable over temperature and supply variation.
3FSK is selected by setting the MODULATION_SCHEME bits
(R2_DB[4:6]) to 010. It can also be used with raised cosine
filtering to further increase the spectral efficiency of the transmit
signal.
4-Level Frequency Shift Keying (4FSK)
In 4FSK modulation, two bits per symbol spectral efficiency is
realized by mapping consecutive input bit-pairs in the Tx data
bit stream to one of four possible symbols (−3, −1, +1, +3). Thus,
the transmitted symbol rate is half of the input bit rate.
Gaussian Frequency Shift Keying (GFSK)
Gaussian frequency shift keying reduces the bandwidth occupied
by the transmitted spectrum by digitally prefiltering the transmit
data. The BT product of the Gaussian filter used is 0.5.
By minimizing the separation between symbol frequencies,
4FSK can have high spectral efficiency. The bit-to-symbol
mapping for 4FSK is gray coded and is shown in Figure 44.
Gaussian filtering can only be used with 2FSK modulation. This
is selected by setting R2_DB[4:6] to 001.
Tx DATA
0
0
0
1
1
0
1
1
Raised Cosine Filtering
f
Raised cosine filtering provides digital prefiltering of the transmit
data by using a raised cosine filter with a roll-off factor (alpha)
of either 0.5 or 0.7. The alpha is set to 0.5 by default, but the
raised cosine filter bandwidth can be increased to provide less
aggressive data filtering by using an alpha of 0.7 (set R2_DB30
to Logic 1). Raised cosine filtering can be used with 2FSK,
3FSK, and 4FSK.
+3fDEV
+fDEV
SYMBOL
FREQUENCIES
–fDEV
–3fDEV
Raised cosine filtering is enabled by setting R2_DB[4:6] as
outlined in Table 11.
t
Figure 44. 4FSK Bit-to-Symbol Mapping
The inner deviation frequencies (+fDEV and −fDEV) are set using
the Tx_FREQUENCY_DEVIATION bits, R2_DB[19:27]. The
outer deviation frequencies are automatically set to three times
the inner deviation frequency.
Rev. 0 | Page 28 of 64
ADF7021-N
Table 12. Bit/Symbol Latency in Transmit Mode for Various
Modulation Schemes
MODULATION AND FILTERING OPTIONS
The various modulation and data filtering options are described
in Table 11.
Modulation
Latency
2FSK
1 bit
Table 11. Modulation and Filtering Options
GFSK
4 bits
Modulation
BINARY FSK
2FSK
Data Filtering R2_DB[4:6]
RC2FSK, Alpha = 0.5
RC2FSK, Alpha = 0.7
3FSK
RC3FSK, Alpha = 0.5
RC3FSK, Alpha = 0.7
4FSK
5 bits
4 bits
None
None
None
000
000
000
1 bit
5 bits
4 bits
MSK1
OQPSK with Half Sine
Baseband Shaping2
GFSK
GMSK3
1 symbol
5 symbols
4 symbols
Gaussian
Gaussian
Raised cosine
None
001
001
101
100
RC4FSK, Alpha = 0.5
RC4FSK, Alpha = 0.7
RC2FSK
Oversampled 2FSK
3-LEVEL FSK
3FSK
TEST PATTERN GENERATOR
The ADF7021-N has a number of built-in test pattern generators
that can be used to facilitate radio link setup or RF measurement.
None
Raised cosine
010
110
RC3FSK
A full list of the supported patterns is shown in Table 13. The
data rate for these test patterns is the programmed data rate set
in Register 3.
4-LEVEL FSK
4FSK
RC4FSK
None
Raised cosine
011
111
The PN9 sequence is suitable for test modulation when carrying
out adjacent channel power (ACP) or occupied bandwidth
measurements.
1 MSK is 2FSK modulation with a modulation index = 0.5.
2 Offset quadrature phase shift keying (OQPSK) with half sine baseband shaping
is spectrally equivalent to MSK.
3 GMSK is GFSK with a modulation index = 0.5.
Table 13. Transmit Test Pattern Generator Options
TRANSMIT LATENCY
Test Pattern
R15_DB[8:10]
Transmit latency is the delay time from the sampling of a
bit/symbol by the TxRxCLK signal to when that bit/symbol
appears at the RF output. The latency without any data filtering
is one bit. The addition of data filtering adds a further latency as
outlined in Table 12.
Normal
Transmit Carrier
000
001
010
011
100
101
110
Transmit + fDEV Tone
Transmit − fDEV Tone
Transmit 1010 Pattern
Transmit PN9 Sequence
Transmit SWD Pattern Repeatedly
It is important that the ADF7021-N be left in transmit mode
after the last data bit is sampled by the data clock to account for
this latency. The ADF7021-N should stay in transmit mode for
a time equal to the number of latency bit periods for the applied
modulation scheme. This ensures that all of the data sampled by
the TxRxCLK signal appears at RF.
The figures for latency in Table 12 assume that the positive
TxRxCLK edge is used to sample data (default). If the TxRxCLK
is inverted by setting R2_DB[28:29], an additional 0.5 bit
latency can be added to all values in Table 12.
Rev. 0 | Page 29 of 64
Error! Unknown document
property name.
ADF7021-N
RECEIVER SECTION
chosen as a compromise between interference rejection and
attenuation of the desired signal.
RF FRONT END
The ADF7021-N is based on a fully integrated, low IF receiver
architecture. The low IF architecture facilitates a very low
external component count and does not suffer from powerline-
induced interference problems.
If the AGC loop is disabled, the gain of the IF filter can be set to one
of three levels by using the FILTER_GAIN bits (R9_DB[22:23]).
The filter gain is adjusted automatically if the AGC loop is
enabled.
Figure 45 shows the structure of the receiver front end. The
many programming options allow users to trade off sensitivity,
linearity, and current consumption to best suit their application.
To achieve a high level of resilience against spurious reception,
the low noise amplifier (LNA) features a differential input.
Switch SW2 shorts the LNA input when transmit mode is
selected (R0_DB27 = 0). This feature facilitates the design of a
combined LNA/PA matching network, avoiding the need for an
external Rx/Tx switch. See the LNA/PA Matching section for
details on the design of the matching network.
IF Filter Bandwidth and Center Frequency Calibration
To compensate for manufacturing tolerances, the IF filter should be
calibrated after power-up to ensure that the bandwidth and
center frequency are correct. Coarse and fine calibration
schemes are provided to offer a choice between fast calibration
(coarse calibration) and high filter centering accuracy (fine
calibration). Coarse calibration is enabled by setting R5_DB4
high. Fine calibration is enabled by setting R6_DB4 high.
For details on when it is necessary to perform a filter
calibration, and in what applications to use either a coarse
calibration or fine calibration, refer to the IF Filter Bandwidth
Calibration section.
I (TO FILTER)
RFIN
Tx/Rx SELECT
SW2 LNA
LO
(R0_DB27)
RFINB
Q (TO FILTER)
RSSI/AGC
LNA_MODE
(R9_DB25)
The RSSI is implemented as a successive compression log amp
following the baseband (BB) channel filtering. The log amp
achieves 3 dB log linearity. It also doubles as a limiter to
convert the signal-to-digital levels for the FSK demodulator.
The offset correction circuit uses the BBOS_CLK_DIVIDE bits
(R3_DB[4:5]), which should be set between 1 MHz and 2 MHz.
The RSSI level is converted for user readback and for digitally
controlled AGC by an 80-level (7-bit) flash ADC. This level can
be converted to input power in dBm. By default, the AGC is on
when powered up in receive mode.
MIXER LINEARITY
(R9_DB28)
LNA_BIAS
(R9_DB[26:27])
LNA_GAIN
(R9_DB[20:21])
LNA/MIXER_ENABLE
(R8_DB6)
Figure 45. RF Front End
The LNA is followed by a quadrature downconversion mixer,
which converts the RF signal to the IF frequency of 100 kHz.
An important consideration is that the output frequency of the
synthesizer must be programmed to a value 100 kHz below the
center frequency of the received channel. The LNA has two
basic operating modes: high gain/low noise mode and low
gain/low power mode. To switch between these two modes, use
the LNA_MODE bit (R9_DB25). The mixer is also configurable
between a low current and an enhanced linearity mode using
the MIXER_LINEARITY bit (R9_DB28).
OFFSET
CORRECTION
FSK
DEMOD
1
A
A
A
LATCH
CLK
IFWR
IFWR
IFWR
IFWR
RSSI
ADC
R
Based on the specific sensitivity and linearity requirements of
the application, it is recommended to adjust the LNA_MODE
bit and MIXER_LINEARITY bit as outlined in Table 15.
Figure 46. RSSI Block Diagram
RSSI Thresholds
The gain of the LNA is configured by the LNA_GAIN bits
(R9_DB[20:21]) and can be set by either the user or the
automatic gain control (AGC) logic.
When the RSSI is above AGC_HIGH_THRESHOLD
(R9_DB[11:17]), the gain is reduced. When the RSSI is
below AGC_LOW_THRESHOLD (R9_DB[4:10]), the gain
is increased. The thresholds default to 30 and 70 on power-up
in receive mode. A delay (set by AGC_CLK_DIVIDE,
R3_DB[26:31]) is programmed to allow for settling of the loop.
A value of 13 is recommended to give an AGC update rate of
7.7 kHz.
IF FILTER
IF Filter Settings
Out-of-band interference is rejected by means of a fifth-order
Butterworth polyphase IF filter centered on a frequency of
100 kHz. The bandwidth of the IF filter can be programmed to
9 kHz, 13.5 kHz, or 18.5 kHz by R4_DB[30:31] and should be
Rev. 0 | Page 30 of 64
ADF7021-N
The user has the option of changing the two threshold values
from the defaults of 30 and 70 (Register 9). The default AGC
setup values should be adequate for most applications. The
threshold values must be more than 30 apart for the AGC to
operate correctly.
By using the recommended setting for AGC_CLK_DIVIDE, the
total AGC settling time is
Number of AGC Gain Changes
AGC Settling Time [sec] =
AGC Update Rate [Hz]
The worst case for AGC settling occurs when the AGC control
loop has to cycle through all five gain settings, which gives a
maximum AGC settling time of 650 μs.
Offset Correction Clock
In Register 3, the user should set the BBOS_CLK_DIVIDE bits
(R3_DB[4:5]) to give a baseband offset clock (BBOS CLK)
frequency between 1 MHz and 2 MHz.
RSSI Formula (Converting to dBm)
The RSSI formula is
BBOS CLK [Hz] = XTAL/(BBOS_CLK_DIVIDE)
where BBOS_CLK_DIVIDE can be set to 4, 8, 16, or 32.
AGC Information and Timing
Input Power [dBm] = −130 dBm + (Readback Code + Gain
Mode Correction) ꢁ 0.5
where:
AGC is selected by default and operates by setting the appropriate
LNA and filter gain settings for the measured RSSI level. It is
possible to disable AGC by writing to Register 9 if the user wants to
enter one of the modes listed in Table 15. The time for the AGC
circuit to settle and, therefore, the time it takes to measure the RSSI
accurately, is typically 390 μs. However, this depends on how many
gain settings the AGC circuit has to cycle through. After each gain
change, the AGC loop waits for a programmed time to allow
transients to settle. This AGC update rate is set according to
Readback Code is given by Bit RV7 to Bit RV1 in the Register 7
readback register (see Figure 58 and the Readback Format
section).
Gain Mode Correction is given by the values in Table 14.
The LNA gain (LG2, LG1) and filter gain (FG2, FG1) values
are also obtained from the readback register, as part of an RSSI
readback.
Table 14. Gain Mode Correction
SEQ _ CLK _ DIVIDE [Hz]
AGC Update Rate [Hz] =
LNA Gain
(LG2, LG1)
Filter Gain
(FG2, FG1)
Gain Mode
Correction
AGC _ CLK _ DIVIDE
H (1, 0)
M (0, 1)
M (0, 1)
M (0, 1)
L (0, 0)
H (1, 0)
H (1, 0)
M (0, 1)
L (0, 0)
L (0, 0)
0
where:
24
38
58
86
AGC_CLK_DIVIDE is set by R3_DB[26:31]. A value of 13 is
recommended.
SEQ_CLK_DIVIDE = 100 kHz (R3_DB[18:25]).
An additional factor should be introduced to account for losses
in the front-end-matching network/antenna.
Table 15. LNA/Mixer Modes
LNA_MODE LNA_GAIN
MIXER_LINEARITY Sensitivity (2FSK, DR =
Rx Current
Consumption (mA)
Input IP3
(dBm)
Receiver Mode
(R9_DB25)
(R9_DB[20:21]) (R9_DB28)
4.8 kbps, fDEV = 4 kHz)
High Sensitivity
Mode (Default)
0
30
30
0
1
−118
24.6
24.6
−24
Enhanced Linearity
High Gain
0
−114.5
−20
Medium Gain
1
1
10
10
0
1
−112
22.1
22.1
−13.5
−9
Enhanced Linearity
Medium Gain
−105.5
Low Gain
1
1
3
3
0
1
−100
−92.3
22.1
22.1
−5
−3
Enhanced Linearity
Low Gain
Rev. 0 | Page 31 of 64
ADF7021-N
Correlator Demodulator
DEMODULATION, DETECTION, AND CDR
The correlator demodulator can be used for 2FSK, 3FSK, and
4FSK demodulation. Figure 48 shows the operation of the
correlator demodulator for 2FSK.
System Overview
An overview of the demodulation, detection, and clock and
data recovery (CDR) of the received signal on the ADF7021-N
is shown in Figure 47.
FREQUENCY CORRELATOR
DISCRIM BW
The quadrature outputs of the IF filter are first limited and
then fed to either the correlator FSK demodulator or to the
linear FSK demodulator. The correlator demodulator is used
to demodulate 2FSK, 3FSK, and 4FSK. The linear demodulator
is used for frequency measurement and is enabled when the
AFC loop is active. The linear demodulator can also be used
to demodulate 2FSK.
I
OUTPUT LEVELS:
2FSK = +1, –1
3FSK = +1, 0, –1
LIMITERS
Q
4FSK = +3, +1, –1, –3
IF
IF – fDEV
IF + fDEV
R4_DB9
Rx_INVERT
R4_DB(10:19)
DISCRIMINATOR_BW
Following the demodulator, a digital post demodulator filter
removes excess noise from the demodulator signal output.
Threshold/slicer detection is used for data recovery of 2FSK and
4FSK. Data recovery of 3FSK can be implemented using either
threshold detection or Viterbi detection.
R4_DB7
DOT_PRODUCT
Figure 48. 2FSK Correlator Demodulator Operation
The quadrature outputs of the IF filter are first limited and then
fed to a digital frequency correlator that performs filtering and
frequency discrimination of the 2FSK/3FSK/4FSK spectrum.
An on-chip CDR PLL is used to resynchronize the received bit
stream to a local clock. It outputs the retimed data and clock on
the TxRxDATA and TxRxCLK pins, respectively.
For 2FSK modulation, data is recovered by comparing the
output levels from two correlators. The performance of this
frequency discriminator approximates that of a matched filter
detector, which is known to provide optimum detection in the
presence of additive white Gaussian noise (AWGN). This
method of FSK demodulation provides approximately 3 dB to
4 dB better sensitivity than a linear demodulator.
LIMITERS
I
FREQUENCY
CORRELATOR
Q
MUX
LINEAR
DEMODULATOR
THRESHOLD
DETECTION
2/3/4FSK
TxRxDATA
TxRxCLK
CLOCK
AND
DATA
MUX
RECOVERY
VITERBI
DETECTION
3FSK
Figure 47. Overview of Demodulation, Detection, and CDR Process
Rev. 0 | Page 32 of 64
ADF7021-N
Linear Demodulator
3FSK and 4FSK Threshold Detection
Figure 49 shows a block diagram of the linear demodulator.
4FSK demodulation is implemented using the correlator
demodulator followed by the post demodulator filter and
threshold detection. The output of the post demodulation
filter is a 4-level signal that represents the transmitted symbols
(−3, −1, +1, +3). Threshold detection of 4FSK requires three
threshold settings, one that is always fixed at 0 and two that
are programmable and are symmetrically placed above and
below zero using the 3FSK/4FSK_SLICER_THRESHOLD bits
(R13_DB[4:10]).
I
LEVEL
IF
+
2FSK RxDATA
LIMITER
Q
SLICER
2FSK
FREQUENCY
LINEAR
DISCRIMINATOR
RxCLK
R4_DB(20:29)
FREQUENCY
READBACK
AND AFC LOOP
Figure 49. Block Diagram of Linear FSK Demodulator
3FSK demodulation is implemented using the correlator demodu-
lator, followed by a post demodulator filter. The output of the
post demodulator filter is a 3-level signal that represents the
transmitted symbols (−1, 0, +1). Data recovery of 3FSK can be
implemented using threshold detection or Viterbi detection.
Threshold detection is implemented using two thresholds that
are programmable and are symmetrically placed above and
below zero using the 3FSK/4FSK_SLICER_THRESHOLD bits
(R13_DB[4:10]).
A digital frequency discriminator provides an output signal that
is linearly proportional to the frequency of the limiter outputs.
The discriminator output is filtered and averaged using a combined
averaging filter and envelope detector. The demodulated 2FSK
data from the post demodulator filter is recovered by slicing against
the output of the envelope detector, as shown in Figure 49. This
method of demodulation corrects for frequency errors between
transmitter and receiver when the received spectrum is close to
or within the IF bandwidth. This envelope detector output is
also used for AFC readback and provides the frequency estimate
for the AFC control loop.
3FSK Viterbi Detection
Viterbi detection of 3FSK operates on a four-state trellis and is
implemented using two interleaved Viterbi detectors operating
at half the symbol rate. The Viterbi detector is enabled by
R13_DB11.
Post Demodulator Filter
A second-order, digital low-pass filter removes excess noise from
the demodulated bit stream at the output of the discriminator.
The bandwidth of this post demodulator filter is programmable
and must be optimized for the user’s data rate and received
modulation type. If the bandwidth is set too narrow, performance
degrades due to intersymbol interference (ISI). If the bandwidth
is set too wide, excess noise degrades the performance of the
receiver. The POST_DEMOD_BW bits (R4_DB[20:29]) set the
bandwidth of this filter.
To facilitate different run length constraints in the transmitted
bit stream, the Viterbi path memory length is programmable
in steps of 4 bits, 6 bits, 8 bits, or 32 bits by setting the
VITERBI_PATH_MEMORY bits (R13_DB[13:14]). This
should be set equal to or longer than the maximum number
of consecutive 0s in the interleaved transmit bit stream.
When used with Viterbi detection, the receiver sensitivity
for 3FSK is typically 3 dB greater than that obtained using
threshold detection. When the Viterbi detector is enabled,
however, the receiver bit latency is increased by twice the
Viterbi path memory length.
2FSK Bit Slicer/Threshold Detection
2FSK demodulation can be implemented using the correlator
FSK demodulator or the linear FSK demodulator. In both cases,
threshold detection is used for data recovery at the output of the
post demodulation filter.
Clock Recovery
An oversampled digital clock and data recovery (CDR) PLL is
used to resynchronize the received bit stream to a local clock
in all modulation modes. The oversampled clock rate of the PLL
(CDR CLK) must be set at 32 times the symbol rate (see the
Register 3—Transmit/Receive Clock Register section). The
maximum data/symbol rate tolerance of the CDR PLL is
determined by the number of zero-crossing symbol transitions
in the transmitted packet. For example, if using 2FSK with a
101010 preamble, a maximum tolerance of 3.0% of the data
rate is achieved. However, this tolerance is reduced during
recovery of the remainder of the packet where symbol transi-
tions may not be guaranteed to occur at regular intervals.
To maximize the data rate tolerance of the CDR, some form
of encoding and/or data scrambling is recommended that
guarantees a number of transitions at regular intervals.
The output signal levels of the correlator demodulator are
always centered about zero. Therefore, the slicer threshold level
can be fixed at zero, and the demodulator performance is
independent of the run-length constraints of the transmit data
bit stream. This results in robust data recovery that does not
suffer from the classic baseline wander problems that exist in
the more traditional FSK demodulators.
When the linear demodulator is used for 2FSK demodulation,
the output of the envelope detector is used as the slicer threshold,
and this output tracks frequency errors that are within the IF
filter bandwidth.
Rev. 0 | Page 33 of 64
ADF7021-N
For example, using 2FSK with Manchester-encoded data
achieves a data rate tolerance of 2.0%.
For 3FSK,
K = Round
3
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
100×10
2× fDEV
The CDR PLL is designed for fast acquisition of the recovered
symbols during preamble and typically achieves bit synchro-
nization within 5-symbol transitions of preamble.
For 4FSK,
K = Round4FSK
where:
Round is rounded to the nearest integer.
Round4FSK is rounded to the nearest of the following integers: 32,
31, 28, 27, 24, 23, 20, 19, 16, 15, 12, 11, 8, 7, 4, 3.
3
In 4FSK modulation, the tolerance using the +3, −3, +3, −3
preamble is 3% of the symbol rate (or 1.5% of the data rate).
However, this tolerance is reduced during recovery of the
remainder of the packet where symbol transitions may not be
guaranteed to occur at regular intervals. To maximize the
symbol/data rate tolerance, the remainder of the 4FSK packet
should be constructed so that the transmitted symbols retain close
to dc-free properties by using data scrambling and/or by inserting
specific dc balancing symbols that are inserted in the transmitted
bit stream at regular intervals such as after every 8 or 16 symbols.
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
100×10
4× fDEV
f
DEV is the transmit frequency deviation in Hz. For 4FSK, fDEV is
the frequency deviation used for the 1 symbols (that is, the
inner frequency deviations).
To optimize the coefficients of the correlator, R4_DB7 and
R4_DB[8:9] must also be assigned. The value of these bits
depends on whether K is odd or even. These bits are assigned
according to Table 17 and Table 18.
In 3FSK modulation, the linear convolutional encoder scheme
guarantees that the transmitted symbol sequence is dc-free,
facilitating symbol detection. However, Tx data scrambling is
recommended to limit the run length of zero symbols in the
transmit bit stream. Using 3FSK, the CDR data rate tolerance is
typically 0.5%.
Table 17. Assignment of Correlator K Value for 2FSK and 3FSK
K
K/2
(K + 1)/2
R4_DB7
R4_DB[8:9]
RECEIVER SETUP
Correlator Demodulator Setup
Even
Even
Odd
Odd
Even
Odd
N/A
N/A
N/A
N/A
Even
Odd
0
0
1
1
00
10
00
10
To enable the correlator for various modulation modes, refer to
Table 16.
Table 18. Assignment of Correlator K Value for 4FSK
Table 16. Enabling the Correlator Demodulator
K
R4_DB7
R4_DB[8:9]
Received Modulation
DEMOD_SCHEME (R4_DB[4:6])
Even
Odd
0
1
00
00
2FSK
3FSK
4FSK
001
010
011
Linear Demodulator Setup
To optimize receiver sensitivity, the correlator bandwidth must be
optimized for the specific deviation frequency and modulation
used by the transmitter. The discriminator bandwidth is
controlled by R4_DB[10:19] and is defined as
The linear demodulator can be used for 2FSK demodulation. To
enable the linear demodulator, set the DEMOD_SCHEME bits
(R4_DB[4:6]) to 000.
Post Demodulator Filter Setup
DEMOD CLK ×K
The 3 dB bandwidth of the post demodulator filter should be
set according to the received modulation type and data rate.
The bandwidth is controlled by R4_DB[20:29] and is given by
DISCRIMINATOR _ BW =
where:
400×103
2
11 ×π× fCUTOFF
DEMOD CLK is as defined in the Register 3—Transmit/Receive
POST _ DEMOD _ BW =
Clock Register section.
DEMOD CLK
K is set for each modulation mode according to the following:
where fCUTOFF is the target 3 dB bandwidth in Hz of the post
demodulator filter.
For 2FSK,
3
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
100×10
K = Round
Table 19. Post Demodulator Filter Bandwidth Settings for
2FSK/3FSK/4FSK Modulation Schemes
fDEV
Received
Modulation
Post Demodulator Filter Bandwidth,
fCUTOFF (Hz)
2FSK
3FSK
4FSK
0.75 × data rate
1 × data rate
1.6 × symbol rate (= 0.8 × data rate)
Rev. 0 | Page 34 of 64
ADF7021-N
3FSK Viterbi Detector Setup
3FSK Threshold Detector Setup
The Viterbi detector can be used for 3FSK data detection. This
is activated by setting R13_DB11 to Logic 1.
To activate threshold detection of 3FSK, R13_DB11 should be
set to Logic 0. The 3FSK/4FSK_SLICER_THRESHOLD bits
(R13_DB[4:10]) should be set as outlined in the 3FSK Viterbi
Detector Setup section.
The Viterbi path memory length is programmable in steps of 4,
6, 8, or 32 bits (VITERBI_PATH_MEMORY, R13_DB[13:14]).
3FSK CDR Setup
The path memory length should be set equal to or greater than
the maximum number of consecutive 0s in the interleaved
transmit bit stream.
In 3FSK, a transmit preamble of at least 40 bits of continuous
1s is recommended to ensure a maximum number of symbol
transitions for the CDR to acquire lock.
The Viterbi detector also uses threshold levels to implement the
maximum likelihood detection algorithm. These thresholds are
programmable via the 3FSK/4FSK_SLICER_THRESHOLD bits
(R13_DB[4:10]).
The clock and data recovery for 3FSK requires a number of
parameters in Register 13 to be set (see Table 20).
4FSK Threshold Detector Setup
These bits are assigned as follows:
The threshold for the 4FSK detector is set using the
3FSK/4FSK_SLICER_THRESHOLD bits (R13_DB[4:10]).
The threshold should be set according to
3FSK/4FSK_SLICER_THRESHOLD =
Transmit Frequency Deviation× K
100×103
⎛
⎞
57×⎜
⎟
⎟
⎜
3FSK/4FSK_SLICER_THRESHOLD =
⎝
⎠
4FSK Outer Tx Deviation× K
100×103
⎛
⎞
where K is the value calculated for correlator discriminator
bandwidth.
⎜
⎜
⎟
⎟
78×
⎝
⎠
where K is the value calculated for correlator discriminator
bandwidth.
Table 20. 3FSK CDR Settings
Parameter
Recommended Setting
Purpose
PHASE_CORRECTION (R13_DB12)
3FSK_CDR_THRESHOLD (R13_DB[15:21])
1
Phase correction is on
Sets CDR decision threshold levels
Transmit Frequency Deviation× K
⎛
⎜
⎞
⎟
62×
100×103
⎝
⎠
where K is the value calculated for correlator
discriminator bandwidth.
3FSK_PREAMBLE_TIME_VALIDATE (R13_DB [22:25]) 15
Preamble detector time qualifier
Rev. 0 | Page 35 of 64
ADF7021-N
The linear demodulator (AFC disabled) tracks frequency errors
in the receive signal when the receive signal is within the IF
filter bandwidth. For example, for a receive signal with an
occupied bandwith = 9 kHz, using the 18.5 kHz IF filter
bandwidth allows the linear demodulator to track the signal at
an error of 4.75 kHz with no increase in bit errors or loss in
sensitivity.
DEMODULATOR CONSIDERATIONS
2FSK Preamble
The recommended preamble bit pattern for 2FSK is a dc-free
pattern (such as a 10101010… pattern). Preamble patterns with
longer run-length constraints (such as 11001100…) can also be
used but result in a longer synchronization time of the received
bit stream in the receiver. The preamble needs to allow enough
bits for AGC settling of the receiver and CDR acquisition. A
minimum of 16 preamble bits is recommended when using the
correlator demodulator and 48 bits when using the linear demod-
ulator. When the receiver uses the internal AFC, the minimum
recommended number of preamble bits is 64.
Correlator Demodulator and Low Modulation Indices
The modulation index in 2FSK is defined as
2× fDEV
Modulation Index =
Data Rate
The receiver sensitivity performance and receiver frequency
tolerance can be maximized at low modulation index by
increasing the discriminator bandwidth of the correlator
demodulator. For modulation indices of less than 0.4, it is
recommended to double the correlator bandwidth by
calculating K as follows:
The remaining fields that follow the preamble header do not
have to use dc-free coding. For these fields, the ADF7021-N can
accommodate coding schemes with a run length of greater than
eight bits without any performance degradation. Refer to
Application Note AN-915 for more information.
4FSK Preamble and Data Coding
1003
⎛
⎞
The recommended preamble bit pattern for 4FSK is a repeating
00100010… bit sequence. This 2-level sequence of repeating
−3, +3, −3, +3 symbols is dc-free and maximizes the symbol
timing performance and data recovery of the 4FSK preamble in
the receiver. The minimum recommended length of the
preamble is 32 bits (16 symbols).
⎜
⎜
⎟
⎟
K = Round
2× fDEV
⎝
⎠
The DISCRIMINATOR_BW in Register 4 should be recalculated
using the new K value. Figure 27 highlights the improved
sensitivity that can be achieved for 2FSK modulation, at low
modulation indices, by doubling the correlator bandwidth.
The remainder of the 4FSK packet should be constructed so
that the transmitted symbols retain close to a dc-free balance by
using data scrambling and/or by inserting specific dc balancing
symbols in the transmitted bit stream at regular intervals, such
as after every 8 or 16 symbols.
AFC OPERATION
The ADF7021-N also supports a real-time AFC loop that is
used to remove frequency errors due to mismatches between
the transmit and receive crystals/TCXOs. The AFC loop uses
the linear frequency discriminator block to estimate frequency
errors. The linear FSK discriminator output is filtered and
averaged to remove the FSK frequency modulation using a
combined averaging filter and envelope detector. In receive
mode, the output of the envelope detector provides an estimate
of the average IF frequency.
Demodulator Tolerance to Frequency Errors
Without AFC
The ADF7021-N has a number of options to combat frequency
errors that exist due to mismatches between the transmit and
receive crystals/TCXOs.
With AFC disabled, the correlator demodulator is tolerant to
frequency errors over a 0.3 ꢁ fDEV range, where fDEV is the FSK
frequency deviation. For larger frequency errors, the frequency
tolerance can be increased by adjusting the value of K and thus
doubling the correlator bandwidth.
Two methods of AFC supported on the ADF7021-N are
external AFC and internal AFC.
External AFC
Here, the user reads back the frequency information through
the ADF7021-N serial port and applies a frequency correction
value to the fractional-N synthesizer-N divider.
K should then be calculated as
3
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
100×10
2× fDEV
The frequency information is obtained by reading the 16-bit
signed AFC readback, as described in the Readback Format
section, and by applying the following formula:
K = Round
The DISCRIMINATOR_BW setting in Register 4 should also be
recalculated using the new K value. Doubling the correlator
bandwidth to improve frequency error tolerance in this manner
typically results in a 1 dB to 2 dB loss in receiver sensitivity.
Frequency Readback [Hz] = (AFC READBACK ꢁ DEMOD
CLK)/218
Although the AFC READBACK value is a signed number, under
normal operating conditions, it is positive. In the absence of
frequency errors, the frequency readback value is equal to the
IF frequency of 100 kHz.
Rev. 0 | Page 36 of 64
ADF7021-N
Internal AFC
When AFC errors are removed using either the internal or
external AFC, further improvement in receiver sensitivity can
be obtained by reducing the IF filter bandwidth using the
IF_FILTER_BW bits (R4_DB[30:31]).
The ADF7021-N supports a real-time, internal, automatic
frequency control loop. In this mode, an internal control loop
automatically monitors the frequency error and adjusts the
synthesizer-N divider using an internal proportional integral
(PI) control loop.
AUTOMATIC SYNC WORD DETECTION (SWD)
The ADF7021-N also supports automatic detection of the sync
or ID fields. To activate this mode, the sync (or ID) word must
be preprogrammed into the ADF7021-N. In receive mode, this
preprogrammed word is compared to the received bit stream.
When a valid match is identified, the external SWD pin is
asserted by the ADF7021-N on the next Rx clock pulse.
The internal AFC control loop parameters are controlled in
Register 10. The internal AFC loop is activated by setting
R10_DB4 to 1. A scaling coefficient must also be entered, based
on the crystal frequency in use. This is set up in R10_DB[5:16]
and should be calculated using
24
This feature can be used to alert the microprocessor that a
valid channel has been detected. It relaxes the computational
requirements of the microprocessor and reduces the overall
power consumption.
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
2
×500
AFC _ SCALING _ FACTOR = Round
XTAL
Maximum AFC Range
The maximum frequency correction range of the AFC loop is
programmable on the ADF7021-N. This is set by R10_DB[24:31].
The maximum AFC correction range is the difference in
frequency between the upper and lower limits of the AFC
tuning range. For example, if the maximum AFC correction
range is set to 10 kHz, the AFC can adjust the receiver LO
within the fLO 5 kHz range.
The SWD signal can also be used to frame the received packet
by staying high for a preprogrammed number of bytes. The data
packet length can be set in R12_DB[8:15].
The SWD pin status can be configured by setting R12_DB[6:7].
R11_DB[4:5] are used to set the length of the sync/ID word, which
can be 12, 16, 20, or 24 bits long. A value of 24 bits is recommended
to minimize false sync word detection in the receiver that can
occur during recovery of the remainder of the packet or when a
noise/no signal is present at the receiver input. The transmitter
must transmit the sync byte MSB first and the LSB last to ensure
proper alignment in the receiver sync-byte-detection hardware.
However, when RF_DIVIDE_BY_2 (R1_DB18) is enabled, the
programmed range is halved. The user should account for this
halving by doubling the programmed maximum AFC range.
The recommended maximum AFC correction range should be
≤1.5 ꢁ IF filter bandwidth. If the maximum frequency correction
range is set to be >1.5 ꢁ IF filter bandwidth, the attenuation of
the IF filter can degrade the AFC loop sensitivity.
An error tolerance parameter can also be programmed that
accepts a valid match when up to three bits of the word are
incorrect. The error tolerance value is assigned in R11_DB[6:7].
The adjacent channel rejection (ACR) performance of the
receivers can be degraded when AFC is enabled and the AFC
correction range is close to the IF filter bandwidth. However,
because the AFC correction range is programmable, the user
can trade off correction range and ACR performance.
Rev. 0 | Page 37 of 64
ADF7021-N
APPLICATIONS INFORMATION
IF FILTER BANDWIDTH CALIBRATION
Lower Tone Frequency (kHz)
XTAL
The IF filter should be calibrated on every power-up in receive
mode to correct for errors in the bandwidth and filter center
frequency due to process variations. The automatic calibration
requires no external intervention once it is initiated by a write
to Register 5. Depending on numerous factors, such as IF filter
bandwidth, received signal bandwidth, and temperature variation,
the user must determine whether to carry out a coarse
calibration or a fine calibration.
IF_CAL_LOWER_TONE_DIVIDE × 2
Upper Tone Frequency (kHz)
XTAL
IF_CAL_UPPER_TONE_DIVIDE × 2
It is recommended to place the lower tone and upper tone as
outlined in Table 22.
The performance of both calibration methods is outlined in
Table 21.
Table 22. IF Filter Fine Calibration Tone Frequencies
IF Filter
Bandwidth
Lower Tone
Frequency
Upper Tone
Frequency
Table 21. IF Filter Calibration Specifications
Filter Calibration
Method
Center Frequency
Calibration
Time (Typ)
9 kHz
13.5 kHz
18.5 kHz
78.1 kHz
79.4 kHz
78.1 kHz
116.3 kHz
116.3 kHz
119 kHz
Accuracy1
Coarse Calibration
Fine Calibration
100 kHz 2.5 kHz
100 kHz 0.6 kHz
200 μs
8.2 ms
Because the filter attenuation is slightly asymmetrical, it is
necessary to have a small offset in the filter center frequency to
give near equal rejection at the upper and lower adjacent
channels. The calibration tones given in Table 22 give this small
positive offset in the IF filter center frequency.
1 After calibration.
Calibration Setup
IF Filter calibration is initiated by writing to Register 5 and
setting the IF_CAL_COARSE bit (R5_DB4). This initiates a
coarse filter calibration. If the IF_FINE_CAL bit (R6_DB4) has
already been configured high, the coarse calibration is followed
by a fine calibration, otherwise the calibration ends.
In some applications, an offset may not be required, and the
user may wish to center the IF filter exactly at 100 kHz. In this
case, the user can alter the tone frequencies from those given in
Table 22 to adjust the fine calibration result.
Once initiated by writing to the part, the calibration is performed
automatically without any user intervention. Calibration time is
200 μs for coarse calibration and a few milliseconds for fine
calibration, during which time the ADF7021-N should not be
accessed. The IF filter calibration logic requires that the
IF_FILTER_DIVIDER bits (R5_DB[5:13]) be set such that
The calibration algorithm adjusts the filter center frequency
and measures the RSSI 10 times during the calibration. The
time for an adjustment plus RSSI measurement is given by
IF_CAL_DWELL_TIME
IF Tone Calibration Time =
SEQ CLK
XTAL[Hz]
= 50 kHz
It is recommended that the IF tone calibration time be at least
800 μs. The total time for the IF filter fine calibration is given by
IF _ FILTER _ DIVIDER
The fine calibration uses two internally generated tones at
certain offsets around the IF filter. The two tones are attenuated
by the IF filter, and the level of this attenuation is measured
using the RSSI. The filter center frequency is adjusted to allow
equal attenuation of both tones. The attenuation of the two test
tones is then remeasured. This continues for a maximum of
10 RSSI measurements, at which stage the calibration algorithm
sets the IF filter center frequency to within 0.6 kHz of 100 kHz.
IF Filter Fine Calibration Time = IF Tone Calibration Time × 10
When to Use Coarse Calibration
It is recommended to perform a coarse calibration on every
receive mode power-up. This calibration typically takes 200 μs.
The FILTER_CAL_COMPLETE signal from MUXOUT can be
used to monitor the filter calibration duration or to signal the
end of calibration. The ADF7021-N should not be accessed
during calibration.
The frequency of these tones is set by the IF_CAL_LOWER_
TONE_DIVIDE (R6_DB[5:12]) and IF_CAL_UPPER_TONE_
DIVIDE (R6_DB[13:20]) bits, outlined in the following equations:
Rev. 0 | Page 38 of 64
ADF7021-N
When to Use a Fine Calibration
LNA/PA MATCHING
In cases where the receive signal bandwidth is very close to the
bandwidth of the IF filter, it is recommended to perform a fine
filter calibration every time the unit powers up in receive mode.
The ADF7021-N exhibits optimum performance in terms of
sensitivity, transmit power, and current consumption, only if its
RF input and output ports are properly matched to the antenna
impedance. For cost-sensitive applications, the ADF7021-N is
equipped with an internal Rx/Tx switch that facilitates the use
of a simple, combined passive PA/LNA matching network.
Alternatively, an external Rx/Tx switch such as the ADG919 can
be used, which yields a slightly improved receiver sensitivity
and lower transmitter power consumption.
A fine calibration should be performed if
OBW + Coarse Calibration Variation > IF_FILTER_BW
where:
OBW is the 99% occupied bandwidth of the transmit signal.
Coarse Calibration Variation is 2.5 kHz.
IF_FILTER_BW is set by R4_DB[30:31].
Internal Rx/Tx Switch
The FILTER_CAL_COMPLETE signal from MUXOUT (set by
R0_DB[29:31]) can be used to monitor the filter calibration
duration or to signal the end of calibration. A coarse filter
calibration is automatically performed prior to a fine filter
calibration.
Figure 50 shows the ADF7021-N in a configuration where
the internal Rx/Tx switch is used with a combined LNA/PA
matching network. This is the configuration used on the EVAL-
ADF7021-NDBxx evaluation board. For most applications, the
slight performance degradation of 1 dB to 2 dB caused by the
internal Rx/Tx switch is acceptable, allowing the user to take
advantage of the cost saving potential of this solution. The
design of the combined matching network must compensate for
the reactance presented by the networks in the Tx and the Rx
paths, taking the state of the Rx/Tx switch into consideration.
When to Use Single Fine Calibration
In applications where the receiver powers up numerous times in
a short period, it is only necessary to perform a one-time fine
calibration on the initial receiver power-up.
After the initial coarse calibration and fine calibration, the result of
the fine calibration can be read back through the serial interface
using the FILTER_CAL_READBACK result (refer to the Filter
Bandwidth Calibration Readback section). On subsequent
power-ups in receive mode, the filter is manually adjusted using
the previous fine filter calibration result. This manual adjust is
performed using the IF_FILTER_ADJUST bits (R5_DB[14:19]).
V
BAT
L1
C1
PA_OUT
PA
ANTENNA
Z _PA
OPT
OPTIONAL
BPF OR LPF
Z
_RFIN
IN
C
A
RFIN
This method should only be used if the successive power-ups in
receive mode are over a short duration, during which time there
is little variation in temperature (<15ꢀC).
L
LNA
A
RFINB
Z
_RFIN
IN
IF Filter Variation with Temperature
C
B
ADF7021-N
When calibrated, the filter center frequency can vary with changes
in temperature. If the ADF7021-N is used in an application where
it remains in receive mode for a considerable length of time, the
user must consider this variation of filter center frequency with
temperature. This variation is typically 1 kHz per 20ꢀC, which
means that if a coarse filter calibration and fine filter calibration
are performed at 25ꢀC, the initial maximum error is 0.5 kHz,
and the maximum possible change in the filter center frequency
over temperature (−40ꢀC to +85ꢀC) is 3.25 kHz. This gives a
total error of 3.75 kHz.
Figure 50. ADF7021-N with Internal Rx/Tx Switch
The procedure typically requires several iterations until an
acceptable compromise has been reached. The successful imple-
mentation of a combined LNA/PA matching network for the
ADF7021-N is critically dependent on the availability of an
accurate electrical model for the PCB. In this context, the use of a
suitable CAD package is strongly recommended. To avoid this
effort, a small form-factor reference design for the ADF7021-N is
provided, including matching and harmonic filter components.
The design is on a 2-layer PCB to minimize cost. Gerber files
are available at www.analog.com.
If the receive signal occupied bandwidth is considerably less
than the IF filter bandwidth, the variation of filter center
frequency over the operating temperature range may not be
an issue. Alternatively, if the IF filter bandwidth is not wide
enough to tolerate the variation with temperature, a periodic
filter calibration can be performed or, alternatively, the on-chip
temperature sensor can be used to determine when a filter cali-
bration is necessary by monitoring for changes in temperature.
Rev. 0 | Page 39 of 64
ADF7021-N
External Rx/Tx Switch
Depending on the antenna configuration, the user may need a
harmonic filter at the PA output to satisfy the spurious emission
requirement of the applicable government regulations. The
harmonic filter can be implemented in various ways, for example, a
discrete LC pi or T-stage filter. The immunity of the ADF7021-N
to strong out-of-band interference can be improved by adding a
band-pass filter in the Rx path. Alternatively, the ADF7021-N
blocking performance can be improved by selecting one of the
enhanced linearity modes, as described in Table 15.
Figure 51 shows a configuration using an external Rx/Tx switch.
This configuration allows an independent optimization of the
matching and filter network in the transmit and receive path.
Therefore, it is more flexible and less difficult to design than the
configuration using the internal Rx/Tx switch. The PA is biased
through Inductor L1, while C1 blocks dc current. Together, L1
and C1 form the matching network that transforms the source
impedance into the optimum PA load impedance, ZOPT_PA.
V
IMAGE REJECTION CALIBRATION
BAT
The image channel in the ADF7021-N is 200 kHz below the
desired signal. The polyphase filter rejects this image with an
asymmetric frequency response. The image rejection performance
of the receiver is dependent on how well matched the I and Q
signals are in amplitude and how well matched the quadrature
is between them (that is, how close to 90ꢀ apart they are). The
uncalibrated image rejection performance is approximately
29 dB (at 450 MHz). However, it is possible to improve on this
performance by as much as 20 dB by finding the optimum I/Q
gain and phase adjust settings.
L1
C1
PA_OUT
OPTIONAL
LPF
PA
ANTENNA
Z
_PA
OPT
Z
_RFIN
IN
C
A
OPTIONAL
BPF
(SAW)
RFIN
L
LNA
A
RFINB
ADG919
Rx/Tx – SELECT
Z
_RFIN
IN
C
B
ADF7021-N
Calibration Using Internal RF Source
Figure 51. ADF7021-N with External Rx/Tx Switch
With the LNA powered off, an on-chip generated, low level RF
tone is applied to the mixer inputs. The LO is adjusted to make
the tone fall at the image frequency where it is attenuated by the
image rejection of the IF filter. The power level of this tone is then
measured using the RSSI readback. The I/Q gain and phase adjust
DACs (R5_DB[20:31]) are adjusted and the RSSI is remeasured.
This process is repeated until the optimum values for the gain
and phase adjust are found that provide the lowest RSSI readback
level, thereby maximizing the image rejection performance of
the receiver.
ZOPT_PA depends on various factors, such as the required
output power, the frequency range, the supply voltage range,
and the temperature range. Selecting an appropriate ZOPT_PA
helps to minimize the Tx current consumption in the application.
Application Note AN-764 and Application Note AN-859 contain a
number of ZOPT_PA values for representative conditions. Under
certain conditions, however, it is recommended to obtain a suitable
ZOPT_PA value by means of a load-pull measurement.
Due to the differential LNA input, the LNA matching network
must be designed to provide both a single-ended-to-differential
conversion and a complex, conjugate impedance match. The
network with the lowest component count that can satisfy these
requirements is the configuration shown in Figure 51, consisting
of two capacitors and one inductor.
Rev. 0 | Page 40 of 64
ADF7021-N
ADF7021-N
RFIN
LNA
RFINB
POLYPHASE
IF FILTER
RSSI/
LOG AMP
MUX
INTERNAL
SIGNAL
SOURCE
7-BIT ADC
PHASE ADJUST
Q
I
FROM LO
SERIAL
INTERFACE
4
PHASE ADJUST
REGISTER 5
RSSI READBACK
4
GAIN ADJUST
REGISTER 5
MICROCONTROLLER
I/Q GAIN/PHASE ADJUST AND
RSSI MEASUREMENT
ALGORITHM
Figure 52. Image Rejection Calibration Using the Internal Calibration Source and a Microcontroller
IR_GAIN_ADJUST_I/Q bit (R5_DB30), whereas the
IR_GAIN_ADJUST_UP/DN bit (R5_DB31) sets whether
the gain adjustment defines a gain or an attenuation adjust.
Using the internal RF source, the RF frequencies that can be
used for image calibration are programmable and are odd
multiples of the reference frequency.
The calibration results are valid over changes in the ADF7021-N
supply voltage. However, there is some variation with temperature.
A typical plot of variation in image rejection over temperature
after initial calibrations at −40ꢀC, +25ꢀC, and +85ꢀC is shown in
Figure 53. The internal temperature sensor on the ADF7021-N
can be used to determine if a new IR calibration is required.
Calibration Using External RF Source
IR calibration can also be implemented using an external RF
source. The IR calibration procedure is the same as that used for
the internal RF source, except that an RF tone is applied to the
LNA input.
Calibration Procedure and Setup
60
CAL AT +25°C
The IR calibration algorithm available from Analog Devices, Inc., is
based on a low complexity, 2D optimization algorithm that can
be implemented in an external microprocessor or microcontroller.
50
40
30
20
10
0
CAL AT +85°C
CAL AT –40°C
To enable the internal RF source, the IR_CAL_SOURCE_
DRIVE_LEVEL bits (R6_DB[28:29]) should be set to the
maximum level. The LNA should be set to its minimum gain
setting, and the AGC should be disabled if the internal source is
being used. Alternatively, an external RF source can be used.
V
= 3.0V
IF BW = 25kHz
DD
INTERFERER SIGNAL:
WANTED SIGNAL:
RF FREQ = 430MHz
MODULATION = 2FSK
DATA RATE = 9.6kbps,
PRBS9
fDEV = 4kHz
LEVEL= –100dBm
RF FREQ = 429.8MHz
MODULATION = 2FSK
DATA RATE = 9.6kbps,
PRBS11
The magnitude of the phase adjust is set by using the IR_PHASE_
ADJUST_MAG bits (R5_DB[20:23]). This correction can be
applied to either the I channel or Q channel, depending on the
value of the IR_PHASE_ADJUST_DIRECTION bit (R5_DB24).
fDEV = 4kHz
–60
–40
–20
0
20
40
60
80
100
TEMPERATURE (°C)
The magnitude of the I/Q gain is adjusted by the IR_GAIN_
ADJUST_MAG bits (R5_DB[25:29]). This correction can be
applied to either the I or Q channel, depending on the value of
Figure 53. Image Rejection Variation with Temperature After Initial
Calibrations at −40°C, +25°C, and +85°C
Rev. 0 | Page 41 of 64
ADF7021-N
to a particular application, such as setting up sync byte
detection or enabling AFC. When going from Tx to Rx or vice
versa, the user needs to toggle the Tx/Rx bit and write only to
Register 0 to alter the LO by 100 kHz.
PACKET STRUCTURE AND CODING
The suggested packet structure to use with the ADF7021-N is
shown in Figure 54.
SYNC
WORD
ID
FIELD
Table 23. Minimum Register Writes Required for Tx/Rx Setup
PREAMBLE
DATA FIELD
CRC
Mode
Registers
Reg 1 Reg 3 Reg 0 Reg 2
Reg 1 Reg 3 Reg 0 Reg 5 Reg 4
Reg 0
Figure 54. Typical Format of a Transmit Protocol
Tx
Rx
Refer to the Receiver Setup section for information on the
required preamble structure and length for the various modulation
schemes.
Tx to Rx and Rx to Tx
The recommended programming sequences for transmit and
receive are shown in Figure 55 and Figure 56, respectively. The
difference in the power-up routine for a TCXO and XTAL
reference is shown in these figures.
PROGRAMMING AFTER INITIAL POWER-UP
Table 23 lists the minimum number of writes needed to set up
the ADF7021-N in either Tx or Rx mode after CE is brought
high. Additional registers can also be written to tailor the part
Rev. 0 | Page 42 of 64
ADF7021-N
TCXO
REFERENCE
XTAL
REFERENCE
POWER-DOWN
CE LOW
CE HIGH
WAIT 10µs + 1ms
(REGULATOR POWER-UP + TYPICAL XTAL SETTLING)
CE HIGH
WAIT 10µs (REGULATOR POWER-UP)
WRITE TO REGISTER 1 (TURNS ON VCO)
WAIT 0.7ms (TYPICAL VCO SETTLING)
WRITE TO REGISTER 3 (TURNS ON Tx/Rx CLOCKS)
WRITE TO REGISTER 0 (TURNS ON PLL)
WAIT 40µs (TYPICAL PLL SETTLING)
WRITE TO REGISTER 2 (TURNS ON PA)
WAIT FOR PA TO RAMP UP (ONLY IF PA RAMP ENABLED)
Tx MODE
WAIT FOR Tx LATENCY NUMBER OF BITS
(REFER TO TABLE 12)
WRITE TO REGISTER 2 (TURNS OFF PA)
WAIT FOR PA TO RAMP DOWN
CE LOW
POWER-DOWN
OPTIONAL. ONLY NECESSARY IF PA
RAMP DOWN IS REQUIRED.
Figure 55. Power-Up Sequence for Transmit Mode
Rev. 0 | Page 43 of 64
ADF7021-N
TCXO
REFERENCE
XTAL
REFERENCE
POWER-DOWN
CE LOW
CE HIGH
CE HIGH
WAIT 10µs (REGULATOR POWER-UP)
WAIT 10µs + 1ms
(REGULATOR POWER-UP + TYPICAL XTAL SETTLING)
WRITE TO REGISTER 1 (TURNS ON VCO)
WAIT 0.7ms (TYPICAL VCO SETTLING)
WRITE TO REGISTER 3 (TURNS ON Tx/Rx CLOCKS)
OPTIONAL:
ONLY NECESSARY IF
IF FILTER FINE CAL IS REQUIRED.
WRITE TO REGISTER 6 (SETS UP IF FILTER CALIBRATION)
WRITE TO REGISTER 5 (STARTS IF FILTER CALIBRATION)
WAIT 0.2ms (COARSE CAL) OR WAIT 8.2ms
(COARSE CALIBRATION + FINE CALIBRATION)
WRITE TO REGISTER 11 (SET UP SWD)
WRITE TO REGISTER 12 (ENABLE SWD)
OPTIONAL:
ONLY NECESSARY IF
SWD IS REQUIRED.
WRITE TO REGISTER 0 (TURNS ON PLL)
WAIT 40µs (TYPICAL PLL SETTLING)
WRITE TO REGISTER 4 (TURNS ON DEMOD)
WRITE TO REGISTER 10 (TURNS ON AFC)
Rx MODE
OPTIONAL:
ONLY NECESSARY IF
AFC IS REQUIRED.
CE LOW
POWER-DOWN
OPTIONAL.
Figure 56. Power-Up Sequence for Receive Mode
Rev. 0 | Page 44 of 64
ADF7021-N
APPLICATIONS CIRCUIT
The ADF7021-N requires very few external components for
operation. Figure 57 shows the recommended application
circuit. Note that the power supply decoupling and regulator
capacitors are omitted for clarity.
For recommended component values, refer to the ADF7021-N
evaluation board data sheet and AN-859 application note
accessible from the ADF7021-N product page. Follow the
reference design schematic closely to ensure optimum
performance in narrow-band applications.
LOOP FILTER
VDD
TCXO
EXT VCO L*
CVCO
CAP
REFERENCE
VDD
VDD
1
36
35
CLKOUT
TxRxCLK
TxRxDATA
SWD
VCOIN
CREG1
VDD1
2
3
4
5
6
MATCHING
TO
MICROCONTROLLER
Tx/Rx SIGNAL
INTERFACE
34
33
32
VDD
VDD
T-STAGE LC
FILTER
RFOUT
RFGND
RFIN
ANTENNA
CONNECTION
VDD2
VDD
31
CREG2
ADCIN
GND2
ADF7021-N
30
29
7
8
RFINB
R
LNA
28
27
26
25
9
VDD4
RSET
CREG4
GND4
SCLK
TO
10
11
12
SREAD
SDATA
SLE
MICROCONTROLLER
CONFIGURATION
INTERFACE
RLNA
RESISTOR
CHIP ENABLE
TO MICROCONTROLLER
RSET
RESISTOR
*PIN 44 AND PIN 46 CAN BE LEFT FLOATING IF EXTERNAL INDUCTOR VCO IS NOT USED.
NOTES
1. PINS [13:18], PINS [20:21], AND PIN 23 ARE TEST PINS AND ARE NOT USED IN NORMAL OPERATION.
Figure 57. Typical Application Circuit (Regulator Capacitors and Power Supply Decoupling Not Shown)
Rev. 0 | Page 45 of 64
ADF7021-N
SERIAL INTERFACE
The serial interface allows the user to program the 16-/32-bit
registers using a 3-wire interface (SCLK, SDATA, and SLE).
It consists of a level shifter, 32-bit shift register, and 16 latches.
Signals should be CMOS compatible. The serial interface is
powered by the regulator, and, therefore, is inactive when CE is low.
AFC Readback
The AFC readback is valid only during the reception of FSK
signals with either the linear or correlator demodulator active.
The AFC readback value is formatted as a signed 16-bit integer
comprising Bit RV1 to Bit RV16 and is scaled according to the
following formula:
Data is clocked into the register, MSB first, on the rising edge of
each clock (SCLK). Data is transferred to one of 16 latches on the
rising edge of SLE. The destination latch is determined by the
value of the four control bits (C4 to C1); these are the bottom
4 LSBs, DB3 to DB0, as shown in Figure 2. Data can also be read
back on the SREAD pin.
FREQ RB [Hz] = (AFC_READBACK ꢁ DEMOD CLK)/218
In the absence of frequency errors, FREQ RB is equal to the IF
frequency of 100 kHz. Note that, for the AFC readback to yield
a valid result, the downconverted input signal must not fall outside
the bandwidth of the analog IF filter. At low input signal levels,
the variation in the readback value can be improved by averaging.
READBACK FORMAT
The readback operation is initiated by writing a valid control
word to the readback register and enabling the READBACK bit
(R7_DB8 = 1). The readback can begin after the control word
has been latched with the SLE signal. SLE must be kept high
while the data is being read out. Each active edge at the SCLK
pin successively clocks the readback word out at the SREAD
pin, as shown in Figure 58, starting with the MSB first. The data
appearing at the first clock cycle following the latch operation
must be ignored. An extra clock cycle is needed after the 16th
readback bit to return the SREAD pin to tristate. Therefore, 18
total clock cycles are needed for each read back. After the 18th
clock cycle, SLE should be brought low.
RSSI Readback
The format of the readback word is shown in Figure 58. It
comprises the RSSI-level information (Bit RV1 to Bit RV7), the
current filter gain (FG1, FG2), and the current LNA gain (LG1,
LG2) setting. The filter and LNA gain are coded in accordance
with the definitions in the Register 9—AGC Register section. For
signal levels below −100 dBm, averaging the measured RSSI values
improves accuracy. The input power can be calculated from the
RSSI readback value as outlined in the RSSI/AGC section.
READBACK MODE
READBACK VALUE
DB8
DB7
RV8
FG1
DB6
RV7
RV7
DB5
RV6
RV6
DB4
RV5
RV5
DB3
RV4
RV4
DB2
RV3
RV3
DB1
RV2
RV2
DB0
RV1
RV1
DB15 DB14 DB13 DB12 DB11 DB10 DB9
AFC READBACK
RSSI READBACK
RV16 RV15 RV14 RV13 RV12 RV11 RV10 RV9
X
X
X
X
X
X
X
X
X
X
LG2
X
LG1
X
FG2
X
BATTERY VOLTAGE/ADCIN/
TEMP. SENSOR READBACK
X
RV7
RV7
RV7
RV6
RV6
RV6
RV5
RV5
RV5
RV4
RV4
RV4
RV3
RV3
RV3
RV2
RV2
RV2
RV1
RV1
RV1
SILICON REVISION
RV16 RV15 RV14 RV13 RV12 RV11 RV10 RV9
RV8
RV8
FILTER CAL READBACK
0
0
0
0
0
0
0
0
Figure 58. Readback Value Table
Rev. 0 | Page 46 of 64
ADF7021-N
Battery Voltage/ADCIN/Temperature Sensor Readback
Filter Bandwidth Calibration Readback
The battery voltage is measured at Pin VDD4. The readback
information is contained in Bit RV1 to Bit RV7. This also
applies to the readback of the voltage at the ADCIN pin and the
temperature sensor. From the readback information, the battery
or ADCIN voltage can be determined using
The filter calibration readback word is contained in Bit RV1 to
Bit RV8 (see Figure 58). This readback can be used for manual
filter adjust, thereby avoiding the need to do an IF filter
calibration in some instances. The manual adjust value is
programmed by R5_DB[14:19]. To calculate the manual adjust
based on a filter calibration readback, use the following formula:
V
V
BATTERY = (BATTERY VOLTAGE READBACK)/21.1
ADCIN = (ADCIN VOLTAGE READBACK)/42.1
IF_FILTER_ADJUST = FILTER_CAL_READBACK − 128
The result should be programmed into R5_DB[14:19] as outlined
in the Register 5—IF Filter Setup Register section.
The temperature can be calculated using
Temp [ꢀC] = −40 + (68.4 − TEMP READBACK) ꢁ 9.32
Silicon Revision Readback
The silicon revision readback word is valid without setting any
other registers. The silicon revision word is coded with four
quartets in BCD format. The product code (PC) is coded with
three quartets extending from Bit RV5 to Bit RV16. The revision
code (RC) is coded with one quartet extending from Bit RV1 to
Bit RV4. The product code for the ADF7021-N should read
back as PC = 0x211. The current revision code should read as
RC = 0x1.
Rev. 0 | Page 47 of 64
ADF7021-N
SPI Mode
INTERFACING TO A MICROCONTROLLER/DSP
In SPI mode, the TxRxCLK pin is configured to input transmit
data in transmit mode. In receive mode, the receive data is available
on the TxRxDATA pin. The data clock in both transmit and receive
modes is available on the CLKOUT pin. In transmit mode, data is
clocked into the ADF7021-N on the positive edge of CLKOUT. In
receive mode, the TxRxDATA data pin should be sampled by
the microcontroller on the positive edge of the CLKOUT.
Standard Transmit/Receive Data Interface
The standard transmit/receive signal and configuration interface
to a microcontroller is shown in Figure 59. In transmit mode,
the ADF7021-N provides the data clock on the TxRxCLK pin,
and the TxRxDATA pin is used as the data input. The transmit
data is clocked into the ADF7021-N on the rising edge of
TxRxCLK.
MICROCONTROLLER
ADuC84x
ADF7021-N
ADF7021-N
TxRxCLK
MISO
MISO
TxRxDATA
TxRxDATA
CLKOUT
MOSI
SCLK
SPI
MOSI
SCLOCK
SS
TxRxCLK
CE
P3.7
CE
SWD
SREAD
SLE
P3.2/INT0
P2.4
SWD
SREAD
SLE
GPIO
P2.5
SDATA
SCLK
GPIO
P2.6
P2.7
SDATA
SCLK
Figure 61. ADF7021-N (SPI Mode) to Microcontroller Interface
Figure 59. ADuC84x to ADF7021-N Connection Diagram
To enable SPI interface mode, set R0_DB28 high and set
R15_DB[17:19] to 0x7. Figure 8 and Figure 9 show the relevant
timing diagrams for SPI mode, while Figure 61 shows the
recommended interface to a microcontroller using the SPI
mode of the ADF7021-N.
In receive mode, the ADF7021-N provides the synchronized
data clock on the TxRxCLK pin. The receive data is available on
the TxRxDATA pin. The rising edge of TxRxCLK should be
used to clock the receive data into the microcontroller. Refer to
Figure 4 and Figure 5 for the relevant timing diagrams.
ADSP-BF533 interface
In 4FSK transmit mode, the MSB of the transmit symbol is
clocked into the ADF7021-N on the first rising edge of the data
clock from the TxRxCLK pin. In 4FSK receive mode, the MSB
of the first payload symbol is clocked out on the first negative
edge of the data clock after the SWD and should be clocked into
the microcontroller on the following rising edge. Refer to Figure 6
and Figure 7 for the relevant timing diagrams.
The suggested method of interfacing to the Blackfin® ADSP-
BF533 is given in Figure 62.
ADSP-BF533
ADF7021-N
SCLK
SCK
MOSI
MISO
SDATA
SREAD
SLE
PF5
RSCLK1
DT1PRI
DR1PRI
RFS1
TxRxCLK
TxRxDATA
UART Mode
In UART mode, the TxRxCLK pin is configured to input transmit
data in transmit mode. In receive mode, the receive data is available
on the TxRxDATA pin, thus providing an asynchronous data
interface. The UART mode can only be used with oversampled
2FSK. Figure 60 shows a possible interface to a microcontroller
using the UART mode of the ADF7021-N. To enable this UART
interface mode, set R0_DB28 high. Figure 8 and Figure 9 show
the relevant timing diagrams for UART mode.
SWD
CE
PF6
Figure 62. ADSP-BF533 to ADF7021-N Connection Diagram
MICROCONTROLLER
ADF7021-N
TxRxCLK
TxDATA
RxDATA
UART
TxRxDATA
CE
SWD
SREAD
SLE
GPIO
SDATA
SCLK
Figure 60. ADF7021-N (UART Mode) to
Asynchronous Microcontroller Interface
Rev. 0 | Page 48 of 64
ADF7021-N
REGISTER 0—N REGISTER
ADDRESS
MUXOUT
INTEGER_N
FRACTIONAL_N
BITS
FRACTIONAL_N
DIVIDE RATIO
TR1
Tx/Rx
M15 M14 M13 ...
M3
M2
M1
0
1
TRANSMIT
RECEIVE
0
0
0
.
.
.
1
1
1
1
0
0
0
.
.
.
1
1
1
1
0
0
0
.
.
.
1
1
1
1
...
...
...
...
...
...
...
...
...
...
0
0
0
.
.
.
1
1
1
1
0
0
1
.
.
.
0
0
1
1
0
1
0
.
.
.
0
1
0
1
0
1
2
.
.
.
U1
UART_MODE
0
1
DISABLED
ENABLED
32764
32765
32766
32767
M3
M2
M1
MUXOUT
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
REGULATOR_READY (DEFAULT)
FILTER_CAL_COMPLETE
DIGITAL_LOCK_DETECT
RSSI_READY
Tx_Rx
LOGIC_ZERO
TRISTATE
LOGIC_ONE
INTEGER_N
N8
N7
N6
N5
N4
N3
N2
N1
DIVIDE RATIO
0
0
.
0
0
.
0
0
.
1
1
.
0
1
.
1
0
.
1
0
.
1
0
.
23
24
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
1
1
1
1
1
1
0
1
253
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
1
254
255
Figure 63. Register 0—N Register Map
•
The RF output frequency is calculated by the following:
For the direct output
•
FILTER_CAL_ COMPLETE in the MUXOUT map in
Figure 63 indicates when a coarse or coarse plus fine
IF filter calibration has finished. DIGITAL_
LOCK_DETECT indicates when the PLL has locked.
RSSI_READY indicates that the RSSI signal has settled
and an RSSI readback can be performed.
Fractional _ N
⎛
⎝
⎞
⎟
⎠
RFOUT = PFD× Integer _ N +
⎜
215
For the RF_DIVIDE_BY_2 (R1_DB18) selected
•
Tx_Rx gives the status of DB27 in this register, which
can be used to control an external Tx/Rx switch.
Fractional _ N
⎛
⎝
⎞
⎟
⎠
RFOUT = PFD×0.5× Integer _ N +
⎜
215
•
In UART/SPI mode, the TxRxCLK pin is used to input the
Tx data. The Rx Data is available on the TxRxDATA pin.
Rev. 0 | Page 49 of 64
ADF7021-N
REGISTER 1—VCO/OSCILLATOR REGISTER
XTAL_
BIAS
ADDRESS
BITS
CLKOUT_
DIVIDE
R_COUNTER
VCO_BIAS
RF R_COUNTER
R3 R2 R1 DIVIDE RATIO
VCO CENTER
FREQ ADJUST
RF_DIVIDE_BY_2
RFD1
0
0
.
0
1
.
1
0
.
1
2
.
VA2
VA1
0
1
OFF
ON
0
0
1
1
0
1
0
1
NOMINAL
VCO ADJUST UP 1
VCO ADJUST UP 2
VCO ADJUST UP 3
.
.
.
.
.
.
.
.
1
1
1
7
VCO_BIAS
CURRENT
0.25mA
VB4
VB3 VB2 VB1
CLKOUT_
DIVIDE RATIO
OFF
0
0
.
0
0
.
0
1
.
1
0
.
CL4
CL3
CL2
CL1
0.5mA
0
0
0
.
0
0
0
.
0
0
1
.
0
1
0
.
2
4
.
1
1
1
1
3.75mA
LOOP
CONDITION
.
.
.
.
.
VE1
.
.
.
.
.
0
1
VCO OFF
VCO ON
30
1
1
1
1
VCL1
VCO_INDUCTOR
XTAL_
DOUBLER
0
1
INTERNAL L VCO
EXTERNAL L VCO
D1
0
1
DISABLE
ENABLED
I
(mA)
CP
XOSC_ENABLE
X1
0
1
CP1
RSET
CP2
3.6kΩ
0.3
OFF
ON
0
0
1
1
0
1
0
1
0.9
XTAL_
BIAS
1.5
XB2 XB1
2.1
0
0
1
1
0
1
0
1
20µA
25µA
30µA
35µA
Figure 64. Register 1—VCO/Oscillator Register Map
•
The R_COUNTER and XTAL_DOUBLER relationship is
as follows:
•
•
The VCO_BIAS bits should be set according to Table 9.
The VCO_ADJUST bits adjust the center of the VCO
operating band. Each bit typically adjusts the VCO band
up by 1% of the RF operating frequency (0.5% if
RF_DIVIDE_BY_2 is enabled).
XTAL
If XTAL_DOUBLER = 0,
PFD =
R _COUNTER
XTAL × 2
R _COUNTER
If XTAL_DOUBLER =1,
PFD =
•
Setting VCO_INDUCTOR to external allows the use of the
external inductor VCO, which gives RF operating
frequencies of 80 MHz to 650 MHz. If the internal
inductor VCO is being used for operation, set this bit low.
•
•
CLOCKOUT_DIVIDE is a divided-down and inverted
version of the XTAL and is available on Pin 36 (CLKOUT).
Set XOSC_ENABLE high when using an external crystal.
If using an external oscillator (such as TCXO) with CMOS-
level outputs into Pin OSC2, set XOSC_ENABLE low. If
using an external oscillator with a 0.8 V p-p clipped sine
wave output into Pin OSC1, set XOSC_ENABLE high.
Rev. 0 | Page 50 of 64
ADF7021-N
REGISTER 2—TRANSMIT MODULATION REGISTER
TxDATA_
INVERT
MODULATION_
SCHEME
ADDRESS
BITS
Tx_FREQUENCY_DEVIATION
POWER_AMPLIFIER
PA_BIAS PA_RAMP
PE1 PA_ENABLED
OFF
ON
PA2 PA1 PA_BIAS
0
1
0
0
1
1
0
1
0
1
5µA
7µA
9µA
11µA
DI2 DI1 TxDATA_INVERT
PR3 PR2 PR1 PA_RAMP RATE
0
0
1
1
0
1
0
1
NORMAL
INVERT CLK
INVERT DATA
INV CLK AND DATA
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
NO RAMP
256 CODES/BIT
128 CODES/BIT
64 CODES/BIT
32 CODES/BIT
16 CODES/BIT
8 CODES/BIT
4 CODES/BIT
fDEV
TFD9 ... TFD3 TFD2 TFD1
0
0
0
0
.
...
...
...
...
...
...
0
0
0
0
.
0
0
1
1
.
0
1
0
1
.
0
1
2
3
S3
S2
S1
MODULATION_SCHEME
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
2FSK
GAUSSIAN 2FSK
3FSK
.
1
1
1
1
511
4FSK
OVERSAMPLED 2FSK
RAISED COSINE 2FSK
RAISED COSINE
RAISED COSINE
3FSK
4FSK
POWER_
NRC1 R-COSINE_ALPHA
P6
.
.
AMPLIFIER
P2
P1
0.5 (Default)
0.7
0
1
0
0
0
0
.
.
.
.
.
.
.
.
.
.
.
.
.
.
0
0
1
1
.
0
1
0
1
.
0 (PA OFF)
1 (–16.0 dBm)
2
3
.
.
.
.
1
.
1
1
1
63 (13 dBm)
Figure 65. Register 2—Transmit Modulation Register Map
•
The 2FSK/3FSK/4FSK frequency deviation is expressed by
the following:
•
•
In the case of 4FSK, there are tones at 3 ꢁ the frequency
deviation and at 1 ꢁ the deviation.
Direct output
The power amplifier (PA) ramps at the programmed rate
(R2_DB[8:10]) until it reaches its programmed level
(R2_DB[13:18]). If the PA is enabled/disabled by the
PA_ENABLE bit (R2_DB7), it ramps up and down. If it is
enabled/disabled by the Tx/Rx bit (R0_DB27), it ramps up
and turns hard off.
Frequency Deviation [Hz] =
Tx_FREQUENCY_DEVIATION ×PFD
216
With RF_DIVIDE_BY_2 (R1_DB18) enabled
Frequency Deviation [Hz] =
•
R-COSINE_ALPHA sets the roll-off factor (alpha) of the
raised cosine data filter to either 0.5 or 0.7. The alpha is set
to 0.5 by default, but the raised cosine filter bandwidth can
be increased to provide less aggressive data filtering by
using an alpha of 0.7.
Tx_FREQUENCY_DEVIATION ×PFD
0.5×
216
where Tx_FREQUENCY_DEVIATION is set by
R2_DB[19:27] and PFD is the PFD frequency.
Rev. 0 | Page 51 of 64
ADF7021-N
REGISTER 3—TRANSMIT/RECEIVE CLOCK REGISTER
DEMOD_CLK_
DIVIDE
ADDRESS
BITS
AGC_CLK_DIVIDE
SEQ_CLK_DIVIDE
CDR_CLK_DIVIDE
SK8 SK7 ...
...
SK3 SK2 SK1 SEQ_CLK_DIVIDE
BK2 BK1 BBOS_CLK_DIVIDE
0
0
.
0
0
.
0
0
.
0
1
.
1
0
.
1
2
.
0
0
1
0
1
4
...
...
...
...
0
1
1
8
16
32
1
1
1
1
1
1
1
1
0
1
254
255
OK4 OK3 OK2 OK1 DEMOD_CLK_DIVIDE
GD6
GD5
GD4
GD3
GD2
GD1 AGC_CLK_DIVIDE
0
0
0
0
INVALID
0
...
1
0
...
1
0
...
1
1
...
1
1
...
15
0
0
...
1
0
0
...
1
0
0
...
1
0
0
...
1
0
0
...
1
0
1
...
1
INVALID
1
...
63
FS8
FS7
...
FS3
FS2
FS1 CDR_CLK_ DIVIDE
0
0
.
1
1
0
0
.
1
1
...
...
...
...
...
0
0
.
1
1
0
1
.
1
1
1
0
.
0
1
1
2
.
254
255
Figure 66. Register 3—Transmit/Receive Clock Register Map
•
•
•
Baseband offset clock frequency (BBOS CLK) must be
greater than 1 MHz and less than 2 MHz, where
•
The sequencer clock (SEQ CLK) supplies the clock to the
digital receive block. It should be as close to 100 kHz as
possible.
XTAL
BBOS _CLK _ DIVIDE
BBOS CLK =
XTAL
SEQ _ CLK _ DIVIDE
SEQ CLK =
Set the demodulator clock (DEMOD CLK) such that
2 MHz ≤ DEMOD CLK ≤ 15 MHz, where
•
The time allowed for each AGC step to settle is determined
by the AGC update rate. It should be set close to 8 kHz.
XTAL
DEMOD CLK =
SEQ CLK
AGC Update Rate [Hz] =
DEMOD _CLK _ DIVIDE
AGC _ CLK _ DIVIDE
For 2FSK/3FSK, the data/clock recovery frequency (CDR
CLK) needs to be within 2% of (32 ꢁ data rate). For 4FSK,
the CDR CLK needs to be within 2% of (32 ꢁ symbol rate).
DEMOD CLK
CDR CLK =
CDR _ CLK _ DIVIDE
Rev. 0 | Page 52 of 64
ADF7021-N
REGISTER 4—DEMODULATOR SETUP REGISTER
Rx_
INVERT
DEMOD_
SCHEME
ADDRESS
BITS
POST_DEMOD_BW
DISCRIMINATOR_BW
IF_FILTER _
BW
DP1
DOT_PRODUCT
IFB2 IFB1
0
1
CROSS_PRODUCT
DOT_PRODUCTD
0
0
1
1
0
1
0
1
9 kHz
13.5 kHz
18.5 kHz
INVALID
RI2 RI1
Rx_INVERT
0
1
0
1
0
0
1
1
NORMAL
INVERT CLK
INVERT DATA
INVERT CLK/DATA
DEMOD_SCHEME
DS3 DS2 DS1
POST_DEMOD_
BW
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
2FSK LINEAR DEMODULATOR
2FSK CORRELATOR DEMODULATOR
3FSK DEMOD
4FSK DEMOD
RESERVED
RESERVED
RESERVED
RESERVED
DW10 .
DW6 DW5 DW4 DW3 DW2 DW1
0
0
.
.
.
.
.
.
.
.
.
.
0
0
.
.
.
0
0
.
.
.
0
0
.
.
.
0
0
.
.
.
0
1
.
.
.
1
0
.
.
.
1
2
.
.
.
.
.
1
.
1
.
1
.
1
.
1
.
1
.
1
1023
TD10 .
TD6 TD5 TD4 TD3 TD2 TD1
DISCRIMINATOR_BW
0
0
.
.
.
.
.
.
.
.
.
.
0
0
.
.
.
0
0
.
.
.
0
0
.
.
.
0
0
.
.
.
0
1
.
.
.
1
0
.
.
.
1
2
.
.
.
.
.
1
.
0
.
1
.
0
.
1
.
0
.
0
660
Figure 67. Register 4—Demodulator Setup Register Map
where:
•
To solve for DISCRIMINATOR_BW, use the following
equation:
Round is rounded to the nearest integer.
Round4FSK is rounded to the nearest of the following integers:
32, 31, 28, 27, 24, 23, 20, 19, 16, 15, 12, 11, 8, 7, 4, 3.
DEMOD CLK × K
DISCRIMINATOR_BW =
400 ×103
f
f
DEV is the transmit frequency deviation in Hz. For 4FSK,
DEV is the frequency deviation used for the 1 symbols
where the maximum value = 660.
(that is, the inner frequency deviations).
•
•
•
For 2FSK,
•
Rx_INVERT (R4_DB[8:9]) and DOT_PRODUCT
(R4_DB7) need to be set as outlined in Table 17 and
Table 18.
3
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
100 ×10
K = Round
For 3FSK,
K = Round
For 4FSK,
K = Round4FSK
fDEV
211 ×π × fCUTOFF
POST_DEMOD_BW =
3
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
100×10
2× fDEV
DEMOD CLK
where the cutoff frequency (fCUTOFF) of the post demod-
ulator filter should typically be 0.75 ꢁ the data rate in
2FSK. In 3FSK, it should be set equal to the data rate, while
in 4FSK, it should be set equal to 1.6 ꢁ symbol rate.
3
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
100×10
4× fDEV
Rev. 0 | Page 53 of 64
ADF7021-N
REGISTER 5—IF FILTER SETUP REGISTER
IR_GAIN_
ADJUST_MAG
IR_PHASE_
ADJUST_MAG
ADDRESS
BITS
IF_FILTER_ADJUST
IF_FILTER_DIVIDER
CC1 IF_CAL_COARSE
0
1
NO CAL
DO CAL
IR PHASE
ADJUST
PM3 PM2 PM1 PM1
0
0
0
.
0
1
2
...
15
0
0
0
.
0
0
1
.
0
1
0
.
IF_FILTER_
IFD6 IFD5 IFD4 IFD3 IFD2 IFD1
DIVIDER
IFD9
.
0
0
.
.
.
.
.
.
.
.
.
.
0
0
.
0
0
.
0
0
.
0
0
.
0
1
.
1
0
.
1
2
.
1
1
1
1
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
1
.
.
.
.
.
.
.
PD1
IR_PHASE_ADJUST_DIRECTION
1
1
1
1
1
1
511
0
1
ADJUST I CH
ADJUST Q CH
IR_GAIN_
ADJUST_MAG
GM5 GM4 GM3 GM2 GM1
0
0
0
.
0
0
0
.
0
0
0
.
0
0
1
.
0
1
0
.
0
1
2
...
31
IFA6 IFA5
IFA2 IFA1 IF_FILTER_ADJUST
...
...
...
...
...
...
...
...
...
...
...
0
0
0
..
0
1
1
1
1
1
0
0
0
..
1
0
0
0
.
0
0
1
..
1
0
0
1
.
0
1
0
..
1
0
1
0
.
0
+1
+2
...
+31
0
–1
–2
...
1
1
1
1
1
GQ1 IR_GAIN_ADJUST_I/Q
0
1
ADJUST I CH
ADJUST Q CH
GA1 IR_GAIN_ADJUST_UP/DN
1
1
–31
1
0
1
GAIN
ATTENUATE
Figure 68. Register 5—IF Filter Setup Register Map
•
•
A coarse IF filter calibration is performed when the
IF_CAL_COARSE bit (R5_DB4) is set. If the IF_FINE_
CAL bit (R6_DB4) has been previously set, a fine IF filter
calibration is automatically performed after the coarse
calibration.
•
IF_FILTER_ADJUST allows the IF fine filter calibration
result to be programmed directly on subsequent receiver
power-ups, thereby saving on the need to redo a fine filter
calibration in some instances. Refer to the Filter Bandwidth
Calibration Readback section for information about using
the IF_FILTER_ ADJUST bits.
Set IF_FILTER_DIVIDER such that
•
R5_DB[20:31] are used for image rejection calibration. Refer
to the Image Rejection Calibration section for details on how
to program these parameters.
XTAL
= 50 kHz
IF _ FILTER _ DIVIDER
Rev. 0 | Page 54 of 64
ADF7021-N
REGISTER 6—IF FINE CAL SETUP REGISTER
ADDRESS
BITS
IF_CAL_DWELL_TIME
IF_CAL_UPPER_TONE_DIVIDE
IF_CAL_LOWER_TONE_DIVIDE
IRD1 IR_CAL_SOURCE ÷2
0
1
SOURCE ÷2 OFF
SOURCE ÷2 ON
IF_CAL_UPPER_
TONE_DIVIDE
FC1 IF_FINE_CAL
... UT3 UT2 UT1
UT8
UT7
0
1
DISABLED
ENABLED
0
1
1
.
1
0
1
.
0
0
0
.
0
0
0
.
...
...
...
...
...
0
0
0
.
1
2
3
.
IR_CAL_SOURCE_
IRC2 IRC1 DRIVE_LEVEL
0
0
1
1
0
1
0
1
OFF
.
.
.
.
.
.
LOW
MED
HIGH
0
1
...
1
1
1
127
IF_CAL_LOWER_
TONE_DIVIDE
LT7
...
LT3 LT2 LT1
LT8
0
0
0
.
0
1
1
.
1
0
1
.
0
0
0
.
0
0
0
.
...
...
...
...
...
1
2
3
.
IF_CAL_
DWELL_TIME
CD3 CD2 CD1
CD7
...
0
0
0
.
0
1
1
.
1
0
1
.
0
0
0
.
...
...
...
...
...
1
2
3
.
.
.
.
.
.
.
1
1
1
1
1
...
255
.
.
.
.
.
1
1
1
1
...
127
Figure 69. Register 6—IF Fine Cal Setup Register Map
•
A fine IF filter calibration is set by enabling the
IF_FINE_CAL Bit (R6_DB4). A fine calibration is then
carried out only when Register 5 is written to and R5_DB4
is set.
•
The IF tone calibration time is the amount of time that is
spent at an IF calibration tone. It is dependent on the
sequencer clock. For best practice, is recommended to have
the IF tone calibration time be at least 500 μs.
IF _ CAL _ DWELL _TIME
IF Tone Calibration Time =
Lower Tone Frequency (kHz) =
SEQ CLK
The total time for a fine IF filter calibration is
IF Tone Calibration Time ꢁ 10
XTAL
IF_CAL_LOWER_TONE_DIVIDE × 2
Upper Tone Frequency (kHz) =
•
R6_DB[28:30] control the internal source for the image
rejection (IR) calibration. The IR_CAL_SOURCE_
DRIVE_LEVEL bits (R6_DB[28:29]) set the drive strength
of the source, whereas the IR_CAL_SOURCE_÷2 bit
(R6_DB30) allows the frequency of the internal signal
source to be divided by 2.
XTAL
IF_CAL_UPPER_TONE_DIVIDE × 2
It is recommended to place the lower tone and upper tone
as outlined in Table 24.
Table 24. IF Filter Fine Calibration Tone Frequencies
IF Filter
Bandwidth
Lower Tone
Frequency
Upper Tone
Frequency
9 kHz
13.5 kHz
18.5 kHz
78.1 kHz
79.4 kHz
78.1 kHz
116.3 kHz
116.3 kHz
119 kHz
Rev. 0 | Page 55 of 64
ADF7021-N
REGISTER 7—READBACK SETUP REGISTER
READBACK_
SELECT
ADC_
MODE
CONTROL
BITS
DB1
DB0
DB7
RB2
DB8
RB3
DB6
RB1
DB5
AD2
DB4
AD1
DB3
DB2
C4 (0) C3 (1) C2 (1) C1 (1)
RB3 READBACK_SELECT
AD2 AD1 ADC_MODE
0
1
DISABLED
ENABLED
0
0
1
1
0
1
0
1
MEASURE RSSI
BATTERY VOLTAGE
TEMP SENSOR
TO EXTERNAL PIN
RB2 RB1 READBACK MODE
0
0
1
1
0
1
0
1
AFC WORD
ADC OUTPUT
FILTER CAL
SILICON REV
Figure 70. Register 7—Readback Setup Register Map
•
•
Readback of the measured RSSI value is valid only in Rx
mode. Readback of the battery voltage, temperature sensor, or
voltage at the external pin is not valid in Rx mode.
•
For AFC readback, use the following equations (see the
Readback Format section):
FREQ RB [Hz] = (AFC READBACK ꢁ DEMOD CLK)/218
To read back the battery voltage, the temperature sensor, or
the voltage at the external pin in Tx mode, users should
first power up the ADC using R8_DB8 because it is turned
off by default in Tx mode to save power.
V
V
BATTERY = BATTERY VOLTAGE READBACK/21.1
ADCIN = ADCIN VOLTAGE READBACK/42.1
Temperature [ꢀC] = −40 + (68.4 − TEMP READBACK) ꢁ 9.32
Rev. 0 | Page 56 of 64
ADF7021-N
REGISTER 8—POWER-DOWN TEST REGISTER
CONTROL
BITS
Rx_RESET
DB10 DB9
LE1 PD6
DB1
DB2
DB0
DB7
PD4
DB15 DB14 DB13 DB12 DB11
DB8
PD5
DB6
PD3
DB5
DB4
PD1
DB3
PD7
SW1
C4 (1) C3 (0) C2 (0) C1 (0)
CR1
PD1 SYNTH_ENABLE
CR1 COUNTER_RESET
0
1
SYNTH OFF
SYNTH ON
0
1
NORMAL
RESET
CDR
RESET
DEMOD
RESET
PD3 LNA/MIXER_ENABLE
PD7 PA (Rx MODE)
0
1
LNA/MIXER OFF
LNA/MIXER ON
0
1
PA OFF
PA ON
PD4 FILTER_ENABLE
SW1 Tx/Rx SWITCH
0
1
FILTER OFF
FILTER ON
0
1
DEFAULT (ON)
OFF
PD5 ADC_ENABLE
LE1 LOG_AMP_ENABLE
0
1
ADC OFF
ADC ON
0
1
LOG AMP OFF
LOG AMP ON
PD6 DEMOD_ENABLE
0
1
DEMOD OFF
DEMOD ON
Figure 71. Register 8—Power-Down Test Register Map
•
It is not necessary to write to this register under normal
operating conditions.
•
For a combined LNA/PA matching network, R8_DB11
should always be set to 0, which enables the internal Tx/Rx
switch. This is the power-up default condition.
Rev. 0 | Page 57 of 64
ADF7021-N
REGISTER 9—AGC REGISTER
FILTER_
GAIN
LNA_
GAIN
AGC_
MODE
ADDRESS
BITS
AGC_HIGH_THRESHOLD
AGC_LOW_THRESHOLD
ML1 MIXER_LINEARITY
AGC_LOW_
THRESHOLD
AGC_MODE
GL7 GL6 GL5 GL4 GL3 GL2 GL1
0
1
DEFAULT
HIGH
0
1
2
3
AUTO AGC
0
0
0
0
.
0
0
0
0
.
0
0
0
0
.
0
0
0
0
.
0
0
0
1
.
0
1
1
0
.
1
0
1
0
.
1
2
3
4
.
.
MANUAL AGC
FREEZE AGC
RESERVED
LI2 LI1 LNA_BIAS
0
0
800µA (DEFAULT)
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
1
1
1
0
1
61
62
63
LG1 LNA_MODE
0
1
DEFAULT
REDUCED GAIN
AGC_HIGH_
THRESHOLD
GH7 GH6 GH5 GH4 GH3 GH2 GH1
FI1 FILTER_CURRENT
0
0
0
0
.
.
.
1
1
1
0
0
0
0
.
.
.
0
0
0
0
0
0
0
.
.
.
0
0
1
0
0
0
0
.
.
.
1
1
0
0
0
0
1
.
.
.
1
1
0
0
1
1
0
.
.
.
1
1
0
1
0
1
0
.
.
.
0
1
0
1
2
3
4
.
.
.
78
79
80
0
1
LOW
HIGH
FG2 FG1 FILTER_GAIN
0
0
1
1
0
1
0
1
8
24
72
INVALID
LG2 LG1 LNA_GAIN
0
0
1
1
0
1
0
1
3
10
30
INVALID
Figure 72. Register 9—AGC Register Map
•
•
It is necessary to program this register only if AGC
settings, other than the defaults, are required.
•
•
AGC high and low settings must be more than 30 apart to
ensure correct operation.
In receive mode, AGC is set to automatic AGC by default
on power-up. The default thresholds are AGC_ LOW_
THRESHOLD = 30 and AGC_HIGH_ THRESHOLD = 70.
See the RSSI/AGC section for details.
An LNA gain of 30 is available only if LNA_MODE
(R9_DB25) is set to 0.
Rev. 0 | Page 58 of 64
ADF7021-N
REGISTER 10—AFC REGISTER
ADDRESS
BITS
MAX_AFC_RANGE
KP
KI
AFC_SCALING_FACTOR
AE1 AFC_EN
KP3 KP2 KP1 KP
KI4 KI3 KI2 KI1 KI
0
0
.
0
0
.
0
0
.
0
0
.
0
1
.
2^0
2^1
...
0
0
.
0
1
.
2^0
2^1
...
0
1
OFF
AFC ON
1
1
1
1
1
2^7
1
1
2^15
MAX_AFC_
MA3 MA2 MA1
RANGE
AFC_SCALING_
FACTOR
...
MA8
...
M3
M2
M1
M12
0
0
0
0
.
.
.
1
1
1
...
...
...
...
...
...
...
...
...
...
0
1
1
0
.
.
.
0
1
1
0
0
0
1
.
.
.
1
1
1
1
0
1
0
.
.
.
1
0
1
1
2
3
4
.
.
.
253
254
255
0
0
0
0
.
.
.
1
1
1
...
...
...
...
...
...
...
...
...
...
0
0
0
1
.
.
.
1
1
1
0
1
1
0
.
.
.
0
1
1
1
0
1
0
.
.
.
1
0
1
1
2
3
4
.
.
.
4093
4094
4095
Figure 73. Register 10—AFC Register Map
•
The AFC_SCALING_FACTOR can be expressed as
•
When the RF_DIVIDE_BY_2 (R1_DB18) is enabled, the
programmed AFC correction range is halved. The user
accounts for this halving by doubling the programmed
MAX_AFC_RANGE value.
24
⎛
⎞
2
×500
⎜
⎜
⎟
⎟
AFC _ SCALING_ FACTOR = Round
XTAL
⎝
⎠
•
•
The settings for KI and KP affect the AFC settling time and
AFC accuracy. The allowable range of each parameter is
KI > 6 and KP < 7.
•
Signals that are within the AFC pull-in range but outside
the IF filter bandwidth are attenuated by the IF filter. As a
result, the signal can be below the sensitivity point of the
receiver and, therefore, not detectable by the AFC.
The recommended settings to give optimal AFC
performance are KI = 11 and KP = 4. To trade off between
AFC settling time and AFC accuracy, the KI and KP
parameters can be adjusted from the recommended settings
(staying within the allowable range) such that
AFC Correction Range = MAX_AFC_RANGE ꢁ 500 Hz
Rev. 0 | Page 59 of 64
ADF7021-N
REGISTER 11—SYNC WORD DETECT REGISTER
CONTROL
BITS
SYNC_BYTE_SEQUENCE
SYNC_BYTE_
PL2 PL1 LENGTH
0
0
1
1
0
1
0
1
12 BITS
16 BITS
20 BITS
24 BITS
MATCHING_
MT2 MT1 TOLERANCE
0
0
1
1
0
1
0
1
ACCEPT 0 ERRORS
ACCEPT 1 ERROR
ACCEPT 2 ERRORS
ACCEPT 3 ERRORS
Figure 74. Register 11—Sync Word Detect Register Map
REGISTER 12—SWD/THRESHOLD SETUP REGISTER
CONTROL
BITS
DATA_PACKET_LENGTH
DATA_PACKET_LENGTH
0
1
...
INVALID
1 BYTE
...
255 255 BYTES
SWD_MODE
0
1
2
SWD PIN LOW
SWD PIN HIGH AFTER NEXT SYNCWORD
SWD PIN HIGH AFTER NEXT SYNCWORD
FOR DATA PACKET LENGTH NUMBER OF BYTES
INTERRUPT PIN HIGH
3
LOCK_THRESHOLD_MODE
0
1
2
THRESHOLD FREE RUNNING
LOCK THRESHOLD AFTER NEXT SYNCWORD
LOCK THRESHOLD AFTER NEXT SYNCWORD
FOR DATA PACKET LENGTH NUMBER OF BYTES
LOCK THRESHOLD
3
Figure 75. Register 12—SWD/Threshold Setup Register Map
Lock threshold locks the threshold of the envelope detector. This has the effect of locking the slicer in linear demodulation and locking
the AFC and AGC loops when using linear or correlator demodulation.
Rev. 0 | Page 60 of 64
ADF7021-N
REGISTER 13—3FSK/4FSK DEMOD REGISTER
Refer to the Receiver Setup section for information about programming these settings.
3FSK_PREAMBLE_
TIME_VALIDATE
3FSK/4FSK_
SLICER_THRESHOLD
CONTROL
BITS
3FSK_CDR_THRESHOLD
3FSK_VITERBI_
DETECTOR
VD1
3FSK_CDR_
VT3 VT2 VT1
VT7
...
THRESHOLD
0
1
DISABLED
ENABLED
0
0
0
0
.
0
0
1
1
.
0
0
0
0
.
...
...
...
...
...
...
0
1
0
1
.
OFF
1
2
3
.
PHASE_
CORRECTION
DISABLED
ENABLED
PC1
0
1
.
.
.
.
.
1
1
1
1
...
127
SLICER
ST3 ST2 ST1
ST7
...
THRESHOLD
VITERBI_PATH _
MEMORY
0
0
0
0
.
0
0
1
1
.
0
0
0
0
.
...
...
...
...
...
...
0
1
0
1
.
OFF
VM2 VM1
1
2
3
.
0
0
1
1
0
1
0
1
4 BITS
6 BITS
8 BITS
32 BITS
.
.
.
.
.
1
1
1
1
...
127
3FSK_PREMABLE_
TIME_VALIDATE
PTV4 PTV3 PTV2 PTV1
0
0
0
0
.
0
0
0
0
.
0
0
1
1
.
0
1
0
1
.
0
1
2
3
.
.
.
.
.
.
1
1
1
1
15
Figure 76. Register 13—3FSK/4FSK Demod Register Map
Rev. 0 | Page 61 of 64
ADF7021-N
REGISTER 14—TEST DAC REGISTER
ADDRESS
BITS
TEST_DAC_GAIN
TEST_DAC_OFFSET
ED_LEAK_FACTOR
LEAKAGE =
PULSE_EXTENSION
TEST_DAC_GAIN
0
1
2
3
NO PULSE EXTENSION
0
NO GAIN
0
1
2
3
4
5
6
7
2^–8
2^–9
EXTENDED BY 1
EXTENDED BY 2
EXTENDED BY 3
1
...
15
× 2^1
...
× 2^15
2^–10
2^–11
2^–12
2^–13
2^–14
2^–15
ED_PEAK_RESPONSE
0
FULL RESPONSE TO PEAK
0.5 RESPONSE TO PEAK
0.25 RESPONSE TO PEAK
0.125 RESPONSE TO PEAK
1
2
3
Figure 77. Register 14—Test DAC Register Map
The demodulator tuning parameters, PULSE_EXTENSION,
ED_LEAK_FACTOR, and ED_PEAK_RESPONSE, can be
enabled only by setting R15_DB[4:7] to 0x9.
While the correlators and filters are clocked by DEMOD CLK,
CDR CLK clocks the test DAC. Note that although the test
DAC functions in regular user mode, the best performance is
achieved when the CDR CLK is increased to or above the
frequency of DEMOD CLK. The CDR block does not function
when this condition exists.
Using the Test DAC to Implement Analog FM DEMOD
and Measuring SNR
For detailed information about using the test DAC, see
Application Note AN-852.
Programming Register 14 enables the test DAC. Both the
linear and correlator/demodulator outputs can be multiplexed
into the DAC.
The test DAC allows the post demodulator filter out for both
linear and correlator demodulators to be viewed externally. The
test DAC also takes the 16-bit filter output and converts it to a
high frequency, single-bit output using a second-order, error
feedback Σ-Δ converter. The output can be viewed on the SWD
pin. This signal, when filtered appropriately, can then be used to
do the following:
Register 14 allows a fixed offset term to be removed from the
signal (to remove the IF component in the ddt case). It also has
a signal gain term to allow the usage of the maximum dynamic
range of the DAC.
•
Monitor the signals at the FSK post demodulator filter
output. This allows the demodulator output SNR to be
measured. Eye diagrams of the received bit stream can also
be constructed to measure the received signal quality.
•
Provide analog FM demodulation.
Rev. 0 | Page 62 of 64
ADF7021-N
REGISTER 15—TEST MODE REGISTER
ANALOG_TEST_
MODES
PLL_TEST_
MODES
Σ-Δ_TEST_
MODES
Tx_TEST_
MODES
Rx_TEST_
MODES
ADDRESS
BITS
CLK_MUX
CAL_OVERRIDE
0
1
2
3
AUTO CAL
OVERRIDE GAIN
OVERRIDE BW
PFD/CP_TEST_MODES
0
1
2
3
4
5
6
7
DEFAULT, NO BLEED
(+VE) CONSTANT BLEED
(–VE) CONSTANT BLEED
(–VE) PULSED BLEED
(–VE) PULSE BLD, DELAY UP?
CP PUMP UP
OVERRIDE BW AND GAIN
REG1_PD
0
1
NORMAL
PWR DWN
CP TRI-STATE
CP PUMP DN
FORCE_LD_HIGH
Σ-Δ_TEST_MODES
0
1
NORMAL
FORCE
0
1
2
3
4
5
6
7
DEFAULT, 3RD ORDER SD, NO DITHER
1ST ORDER SD
2ND ORDER SD
DITHER TO FIRST STAGE
DITHER TO SECOND STAGE
DITHER TO THIRD STAGE
DITHER × 8
ANALOG_TEST_MODES
0
1
2
3
4
5
6
7
8
9
BAND GAP VOLTGE
40µA CURRENT FROM REG4
FILTER I CHANNEL: STAGE 1
FILTER I CHANNEL: STAGE 2
FILTER I CHANNEL: STAGE 1
FILTER Q CHANNEL: STAGE 1
FILTER Q CHANNEL: STAGE 2
FILTER Q CHANNEL: STAGE 1
ADC REFERENCE VOLTAGE
BIAS CURRENT FROM RSSI 5µA
DITHER × 32
Tx_TEST_MODES
0
1
2
3
4
5
6
NORMAL OPERATION
Tx CARRIER ONLY
Tx +VE TONE ONLY
Tx –VE TONE ONLY
Tx "1010" PATTERN
Tx PN9 DATA, AT PROGRAMED RATE
Tx SYNC BYTE REPEATEDLY
10 FILTER COARSE CAL OSCILLATOR O/P
11 ANALOG RSSI I CHANNEL
12 OSET LOOP +VE FBACK V (I CH)
13 SUMMED O/P OF RSSI RECTIFIER+
14 SUMMED O/P OF RSSI RECTIFIER–
15 BIAS CURRENT FROM BB FILTER
Rx_TEST_MODES
0
1
2
3
4
5
6
7
8
9
NORMAL
SCLK, SDATA -> I, Q
REVERSE I,Q
I,Q TO TxRxCLK, TxRxDATA
3FSK SLICER ON TxRxDATA
CORRELATOR SLICER ON TxRxDATA
LINEAR SLICER ON RXDATA
SDATA TO CDR
ADDITIONAL FILTERING ON I, Q
ENABLE REG 14 DEMOD PARAMETERS
PLL_TEST_MODES
0
1
2
3
4
5
6
7
8
9
NORMAL OPERATION
R DIV
N DIV
RCNTR/2 ON MUXOUT
NCNTR/2 ON MUXOUT
ACNTR TO MUXOUT
PFD PUMP UP TO MUXOUT
PFD PUMP DN TO MUXOUT
SDATA TO MUXOUT (OR SREAD?)
10 POWER DOWN DDT AND ED IN T/4 MODE
11 ENVELOPE DETECTOR WATCHDOG DISABLED
12 RESERVED
13 PROHIBIT CALACTIVE
14 FORCE CALACTIVE
ANALOG LOCK DETECT ON MUXOUT
10 END OF COARSE CAL ON MUXOUT
11 END OF FINE CAL ON MUXOUT
12
FORCE NEW PRESCALER CONFIG.
FOR ALL N
15 ENABLE DEMOD DURING CAL
13 TEST MUX SELECTS DATA
14 LOCK DETECT PERCISION
15 RESERVED
CLK MUXES ON CLKOUT PIN
0
1
2
3
4
5
6
7
NORMAL, NO OUTPUT
DEMOD CLK
CDR CLK
SEQ CLK
BB OFFSET CLK
SIGMA DELTA CLK
ADC CLK
TxRxCLK
Figure 78. Register 15—Test Mode Register Map
•
•
Analog RSSI can be viewed on the Test_A pin by setting
ANALOG_TEST_MODES to 11.
Tx_TEST_MODES can be used to enable test modulation.
•
The CDR block can be bypassed by setting Rx_TEST_
MODES to 4, 5, or 6, depending on the demodulator used.
Rev. 0 | Page 63 of 64
ADF7021-N
OUTLINE DIMENSIONS
0.30
0.23
0.18
7.00
BSC SQ
0.60 MAX
0.60 MAX
PIN 1
INDICATOR
37
36
48
1
PIN 1
INDICATOR
EXPOSED
PAD
(BOTTOM VIEW)
4.25
4.10 SQ
3.95
TOP
VIEW
6.75
BSC SQ
0.50
0.40
0.30
25
24
12
13
0.25 MIN
5.50
REF
0.80 MAX
0.65 TYP
1.00
0.85
0.80
12° MAX
0.05 MAX
0.02 NOM
COPLANARITY
0.08
0.50 BSC
0.20 REF
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-220-VKKD-2
Figure 79. 48-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
7 mm × 7 mm Body, Very Thin Quad
(CP-48-3)
Dimensions shown in millimeters
ORDERING GUIDE
Model
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
Package Option
CP-48-3
CP-48-3
ADF7021-NBCPZ1
48-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
48-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
48-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
Die on Film
ADF7021-NBCPZ-RL1
ADF7021-NBCPZ-RL71
ADF7021-NDF
CP-48-3
EVAL-ADF70XXMBZ21
EVAL-ADF7021-NDBIZ1
EVAL-ADF7021-NDBEZ1
EVAL-ADF7021-NDBZ21
EVAL-ADF7021-NDBZ51
Evaluation Platform Mother Board
426 MHz to 429 MHz Daughter Board
426 MHz to 429 MHz Daughter Board
860 MHz to 870 MHz Daughter Board
Matching Unpopulated Daughter Board
1 Z = RoHS Compliant Part.
©2008 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D07246-0-2/08(0)
Rev. 0 | Page 64 of 64
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