ADP3192A [ADI]
8-Bit Programmable 2- to 4-Phase Synchronous Buck Controller; 8位可编程2至4相同步降压控制器![ADP3192A](http://pdffile.icpdf.com/pdf1/p00116/img/icpdf/ADP3192A_636862_icpdf.jpg)
型号: | ADP3192A |
厂家: | ![]() |
描述: | 8-Bit Programmable 2- to 4-Phase Synchronous Buck Controller |
文件: | 总32页 (文件大小:1033K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
8-Bit Programmable 2- to 4-Phase
Synchronous Buck Controller
ADP3192A
FUNCTIONAL BLOCK DIAGRAM
FEATURES
VCC
31
RT RAMPADJ
12 13
Selectable 2-, 3-, or 4-phase operation at up to
1 MHz per phase
7.7 mV worst-case differential sensing error over
temperature
Logic-level PWM outputs for interface to external high
power drivers
Fast enhanced PWM (FEPWM) flex mode for excellent load
transient performance
Active current balancing between all output phases
Built-in power-good/crowbar blanking supports on-the-fly
VID code changes
Digitally programmable 0.5 V to 1.6 V output supports both
VR10.x and VR11 specifications
SHUNT
REGULATOR
OSCILLATOR
19
OD
UVLO
SHUTDOWN
SET EN
RESET
+
18
1
GND
EN
30
29
28
CMP
PWM1
PWM2
PWM3
–
–
+
850mV
+
RESET
CMP
–
DAC
+ 150mV
+
–
+
RESET
CMP
–
2/3/4-PHASE
CSREF
DRIVER LOGIC
RESET
27 PWM4
+
–
+
CMP
–
DAC
– 350mV
CURRENT
LIMIT
DELAY
2
PWRGD
CROWBAR
25
24
23
22
SW1
SW2
SW3
Programmable short-circuit protection with programmable
latch-off delay
10
9
TTSENSE
VRHOT
THERMAL
THROTTLING
CONTROL
VRFAN
8
SW4
17
15
CSCOMP
CSREF
11
7
ILIMIT
CURRENT
+
–
APPLICATIONS
MEASUREMENT
AND LIMIT
DELAY
16
21
CSSUM
IMON
Desktop PC power supplies for next generation
Intel® processors
VRM modules
20
5
IREF
–
+
4
FB
COMP
+
PRECISION
14
LLSET
REFERENCE
–
BOOT
VOLTAGE
AND
SOFT START
CONTROL
3
FBRTN
VIDSEL
6
SS
VID DAC
40
ADP3192A
32
33
34
35
36
37
38
39
VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0
Figure 1.
GENERAL DESCRIPTION
The ADP3192A1 is a highly efficient, multiphase, synchronous
buck-switching regulator controller optimized for converting a
12 V main supply into the core supply voltage required by high
performance Intel processors. It uses an internal 8-bit DAC to read
a voltage identification (VID) code directly from the processor,
which is used to set the output voltage between 0.5 V and 1.6 V.
The ADP3192A also includes programmable no load offset and
slope functions to adjust the output voltage as a function of the
load current, optimally positioning it for a system transient. In
addition, the ADP3192A provides accurate and reliable short-
circuit protection, adjustable current limiting, and a delayed
power-good output that accommodates on-the-fly output
voltage changes requested by the CPU.
This device uses a multimode PWM architecture to drive the
logic-level outputs at a programmable switching frequency that
can be optimized for VR size and efficiency. The phase relation-
ship of the output signals can be programmed to provide 2-, 3-,
or 4-phase operation, allowing for the construction of up to
four complementary buck-switching stages.
The ADP3192A has a built-in shunt regulator that allows the part
to be connected to the 12 V system supply through a series resistor.
The ADP3192A is specified over the extended commercial
temperature range of 0°C to 85°C and is available in a
40-lead LFCSP.
1 Protected by U.S. Patent Number 6,683,441; other patents pending.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registeredtrademarks arethe property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.461.3113
www.analog.com
©2007 Analog Devices, Inc. All rights reserved.
ADP3192A
TABLE OF CONTENTS
Features .............................................................................................. 1
Power-Good Monitoring........................................................... 14
Output Crowbar ......................................................................... 14
Output Enable and UVLO ........................................................ 14
Thermal Monitoring.................................................................. 14
Application Information................................................................ 19
Setting the Clock Frequency..................................................... 19
Soft Start Delay Time................................................................. 19
Current-Limit Latch-Off Delay Times .................................... 19
Inductor Selection...................................................................... 19
Current Sense Amplifier............................................................ 20
Inductor DCR Temperature Correction ................................. 21
Output Offset.............................................................................. 22
COUT Selection ............................................................................. 22
Power MOSFETs......................................................................... 23
Ramp Resistor Selection............................................................ 24
COMP Pin Ramp ....................................................................... 25
Current-Limit Setpoint.............................................................. 25
Feedback Loop Compensation Design.................................... 25
CIN Selection and Input Current di/dt Reduction.................. 27
Thermal Monitor Design .......................................................... 27
Shunt Resistor Design................................................................ 28
Tuning the ADP3192A .............................................................. 28
Layout and Component Placement ......................................... 30
Outline Dimensions....................................................................... 31
Ordering Guide .......................................................................... 31
Applications....................................................................................... 1
Functional Block Diagram .............................................................. 1
General Description......................................................................... 1
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Test Circuits....................................................................................... 5
Absolute Maximum Ratings............................................................ 6
ESD Caution.................................................................................. 6
Pin Configuration and Function Descriptions............................. 7
Typical Performance Characteristics ............................................. 9
Theory of Operation ...................................................................... 10
Start-Up Sequence...................................................................... 10
Phase Detection Sequence......................................................... 10
Master Clock Frequency............................................................ 11
Output Voltage Differential Sensing........................................ 11
Output Current Sensing ............................................................ 11
Active Impedance Control Mode............................................. 11
Current Control Mode and Thermal Balance ........................ 11
Voltage Control Mode................................................................ 12
Current Reference ...................................................................... 12
Fast Enhanced PWM Mode...................................................... 12
Delay Timer................................................................................. 12
Soft Start ...................................................................................... 12
Current-Limit, Short-Circuit, and Latch-Off Protection...... 13
Dynamic VID.............................................................................. 13
REVISION HISTORY
5/07—Revision 0: Initial Version
Rev. 0 | Page 2 of 32
ADP3192A
SPECIFICATIONS
VCC = 5 V, FBRTN = GND, TA = 0°C to 85°C, unless otherwise noted.1
Table 1.
Parameter
Symbol
Conditions
Min
Typ
Max
Unit
REFERENCE CURRENT
Reference Bias Voltage
Reference Bias Current
ERROR AMPLIFIER
Output Voltage Range2
Accuracy
VIREF
IIREF
1.5
14.25 15
V
RIREF = 100 kΩ
15.75 μA
VCOMP
VFB
0
−7.7
4.4
+7.7
V
mV
Relative to nominal DAC output, referenced
to FBRTN, LLSET = CSREF (see Figure 2)
VFB(BOOT)
In startup
CSREF − LLSET = 80 mV
1.092 1.1
1.108
−82
+1
16.5
200
V
Load Line Positioning Accuracy
Differential Nonlinearity
Input Bias Current
FBRTN Current
Output Current
−78
−1
−80
mV
LSB
μA
μA
μA
IFB
IFB = IIREF
13.5
15
65
IFBRTN
ICOMP
GBW(ERR)
FB forced to VOUT – 3%
COMP = FB
COMP = FB
500
20
25
Gain Bandwidth Product
Slew Rate
MHz
V/μs
LLSET Input Voltage Range
LLSET Input Bias Current
Boot Voltage Hold Time
VID INPUTS
VLLSET
ILLSET
tBOOT
Relative to CSREF
−250
−10
+250 mV
+10
nA
ms
CDELAY = 10 nF
2
Input Low Voltage
Input High Voltage
Input Current
VID Transition Delay Time2
No CPU Detection Turn-Off Delay Time2
OSCILLATOR
Frequency Range2
Frequency Variation
VIL(VID)
VIH(VID)
IIN(VID)
VID(X), VIDSEL
VID(X), VIDSEL
0.4
V
V
μA
ns
μs
0.8
−1
VID code change to FB change
VID code change to PWM going low
400
5
fOSC
fPHASE
0.25
180
4
220
MHz
kHz
kHz
kHz
V
TA = 25°C, RT = 205 kΩ, 4-phase
TA = 25°C, RT = 118 kΩ, 4-phase
TA = 25°C, RT = 55 kΩ, 4-phase
RT = 205 kΩ to GND
200
400
800
2.0
Output Voltage
VRT
VRAMPADJ
IRAMPADJ
1.9
−50
1
2.1
+50
50
RAMPADJ Output Voltage
RAMPADJ Input Current Range
CURRENT SENSE AMPLIFIER
Offset Voltage
Input Bias Current
Gain Bandwidth Product
Slew Rate
Input Common-Mode Range
Output Voltage Range
Output Current
Current Limit Latch-Off Delay Time
IMON Output
RAMPADJ − FB
mV
μA
VOS(CSA)
IBIAS(CSSUM)
GBW(CSA)
CSSUM − CSREF (see Figure 3)
−1.0
−10
+1.0
+10
mV
nA
MHz
V/μs
V
CSSUM = CSCOMP
CCSCOMP = 10 pF
CSSUM and CSREF
10
10
0
0.05
3.5
3.5
V
ICSCOMP
tOC(DELAY)
IMON
500
8
μA
ms
%
CDELAY = 10 nF
10 × (CSREF − CSCOMP) > 50 mV
−6
+6
CURRENT BALANCE AMPLIFIER
Common-Mode Range
Input Resistance
Input Current
Input Current Matching
VSW(X)CM
RSW(X)
ISW(X)
−600
10
8
+200 mV
SW(X) = 0 V
SW(X) = 0 V
SW(X) = 0 V
17
12
26
20
+4
kΩ
μA
%
−4
ΔISW(X)
CURRENT-LIMIT COMPARATOR
ILIMIT Bias Current
IILIMIT
IILIMIT = 2/3 × IIREF
9
10
11
μA
Rev. 0 | Page 3 of 32
ADP3192A
Parameter
Symbol
Conditions
Min
1.09
3
Typ
Max
Unit
V
V
mV
mV/V
ILIMIT Voltage
VILIMIT
RILIMIT = 121 kΩ (VILIMIT = (IILIMIT × RILIMIT))
1.21
1.33
Maximum Output Voltage
Current-Limit Threshold Voltage
Current-Limit Setting Ratio
DELAY TIMER
VCL
VCSREF − VCSCOMP, RILIMIT = 121 kΩ
VCL/VILIMIT
80
100
82.6
125
Normal Mode Output Current
Output Current in Current Limit
Threshold Voltage
SOFT START
IDELAY
IDELAY(CL)
VDELAY(TH)
IDELAY = IIREF
IDELAY(CL) = 0.25 × IIREF
12
3.0
1.6
15
3.75
1.7
18
4.5
1.8
μA
μA
V
Output Current
ISS
During startup, ISS = IIREF
12
15
18
μA
ENABLE INPUT
Threshold Voltage
Hysteresis
Input Current
VTH(EN)
VHYS(EN)
IIN(EN)
800
80
850
100
−1
2
900
125
mV
mV
μA
Delay Time
tDELAY(EN)
EN > 950 mV, CDELAY = 10 nF
ms
OD OUTPUT
Output Low Voltage
VOL(
160
5
500
mV
V
)
OD
Output High Voltage
OD Pull-Down Resistor
VOH(
4
0
)
OD
60
kΩ
THERMAL THROTTLING CONTROL
TTSENSE Voltage Range
TTSENSE Bias Current
TTSENSE VRFAN Threshold Voltage
TTSENSE VRHOT Threshold Voltage
TTSENSE Hysteresis
VRFAN Output Low Voltage
VRHOT Output Low Voltage
POWER-GOOD COMPARATOR
Undervoltage Threshold
Overvoltage Threshold
Output Low Voltage
Power-Good Delay Time
During Soft Start2
VID Code Changing
VID Code Static
Crowbar Trip Point
Crowbar Reset Point
Crowbar Delay Time
VID Code Changing
VID Code Static
Internally limited
5
V
−133 −123 −113 μA
1.06
765
1.105 1.15
810
50
150
150
V
855
mV
mV
mV
mV
VOL(VRFAN)
VOL(VRHOT)
IVRFAN(SINK) = −4 mA
IVRHOT(SINK) = −4 mA
300
300
VPWRGD(UV) Relative to nominal DAC output
VPWRGD(OV) Relative to nominal DAC output
VOL(PWRGD)
−400 −350 −300 mV
100
150
150
200
300
mV
mV
IPWRGD(SINK) = −4 mA
CDELAY = 10 nF
2
ms
μs
ns
mV
mV
100
250
200
150
375
VCROWBAR
tCROWBAR
Relative to nominal DAC output
Relative to FBRTN
Overvoltage to PWM going low
100
320
200
430
100
250
400
μs
ns
PWM OUTPUTS
Output Low Voltage
Output High Voltage
SUPPLY
VOL(PWM)
VOH(PWM)
IPWM(SINK) = −400 μA
IPWM(SOURCE) = 400 μA
160
5
500
mV
V
4.0
VSYSTEM = 12 V, RSHUNT = 340 Ω (see Figure 2)
VCC2
VCC
IVCC
4.65
5
5.55
25
11
V
DC Supply Current
VSYSTEM = 13.2 V, RSHUNT = 340 Ω
mA
mA
V
UVLO Turn-On Current
UVLO Threshold Voltage
UVLO Turn-Off Voltage
6.5
4.1
VUVLO
VCC rising
VCC falling
9
V
1 All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC).
2 Guaranteed by design or bench characterization, not tested in production.
Rev. 0 | Page 4 of 32
ADP3192A
TEST CIRCUITS
12V
12V
ADP3192A
680Ω
680Ω
680Ω
680Ω
8-BIT CODE
VCC
31
5
+
1µF
100nF
COMP
40
10kΩ
FB
1
1.25V
4
EN
PWM1
PWM2
PWM3
PWM4
NC
SW1
SW2
SW3
SW4
NC
PWRGD
FBRTN
FB
COMP
SS
DELAY
VRFAN
VRHOT
TTSENSE
LLSET
CSREF
GND
–
+
1kΩ
ADP3192A
14
15
18
10nF
ΔV
10nF
VID
DAC
1V
100kΩ
250kΩ
20kΩ
ΔV = FB = 80mV – FB = 0mV
FB ΔV ΔV
NC = NO CONNECT
Figure 4. Positioning Voltage
100nF
Figure 2. Closed-Loop Output Voltage Accuracy
12V
ADP3192A
680Ω
680Ω
VCC
31
17
CSCOMP
CSSUM
CSREF
GND
100nF
39kΩ
1kΩ
1V
16
15
18
CSCOMP – 1V
V
=
OS
40
Figure 3. Current Sense Amplifier VOS
Rev. 0 | Page 5 of 32
ADP3192A
ABSOLUTE MAXIMUM RATINGS
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Table 2.
Parameter
Rating
VCC
FBRTN
−0.3 V to +6 V
−0.3 V to +0.3 V
−0.3 V to VCC + 0.3 V
−5 V to +25 V
−10 V to +25 V
−0.3 V to VCC + 0.3 V
−65°C to +150°C
0°C to 85°C
PWM3 to PWM4, RAMPADJ
SW1 to SW4
<200 ns
All Other Inputs and Outputs
Storage Temperature Range
Operating Ambient Temperature Range
Operating Junction Temperature
Thermal Impedance (θJA)
Lead Temperature
Soldering (10 sec)
Absolute maximum ratings apply individually only, not in
combination. Unless otherwise specified, all other voltages
referenced to GND.
125°C
39°C/W
ESD CAUTION
300°C
260°C
Infrared (15 sec)
Rev. 0 | Page 6 of 32
ADP3192A
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
EN
PWRGD
FBRTN
FB
COMP
SS
DELAY
VRFAN
VRHOT
1
2
3
4
5
6
7
8
9
30 PWM1
29 PWM2
28 PWM3
27 PWM4
26 NC
25 SW1
24 SW2
23 SW3
22 SW4
21 IMON
INDICATOR
ADP3192A
TOP VIEW
(Not to Scale)
TTSENSE 10
NOTES
1. NC = NO CONNECT.
2. THE EXPOSED EPAD ON BOTTOM SIDE OF PACKAGE IS AN
ELECTRICAL CONNECTION AND SHOULD BE SOLDERED TO GROUND.
Figure 5. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. Mnemonic Description
1
2
EN
PWRGD
Power Supply Enable Input. Pulling this pin to GND disables the PWM outputs and pulls the PWRGD output low.
Power-Good Output. Open-drain output that signals when the output voltage is outside of the proper
operating range.
3
4
FBRTN
FB
Feedback Return. VID DAC and error amplifier reference for remote sensing of the output voltage.
Feedback Input. Error amplifier input for remote sensing of the output voltage. An external resistor between
this pin and the output voltage sets the no load offset point.
5
6
COMP
SS
Error Amplifier Output and Compensation Point.
Soft Start Delay Setting Input. An external capacitor connected between this pin and GND sets the soft start
ramp-up time.
7
DELAY
Delay Timer Setting Input. An external capacitor connected between this pin and GND sets the overcurrent
latch-off delay time, boot voltage hold time, EN delay time, and PWRGD delay time.
8
VRFAN
VRHOT
TTSENSE
VR Fan Activation Output. Open-drain output that signals when the temperature at the monitoring point
connected to TTSENSE exceeds the programmed VRFAN temperature threshold.
VR Hot Output. Open-drain output that signals when the temperature at the monitoring point connected to
TTSENSE exceeds the programmed VRHOT temperature threshold.
VR Hot Thermal Throttling Sense Input. An NTC thermistor between this pin and GND is used to remotely sense
the temperature at the desired thermal monitoring point.
9
10
11
12
ILIMIT
RT
Current-Limit Setpoint. An external resistor from this pin to GND sets the current-limit threshold of the converter.
Frequency Setting Resistor Input. An external resistor connected between this pin and GND sets the oscillator
frequency of the device.
13
14
15
RAMPADJ
LLSET
PWM Ramp Current Input. An external resistor from the converter input voltage to this pin sets the internal
PWM ramp.
Output Load Line Programming Input. This pin can be directly connected to CSCOMP, or it can be connected to
the center point of a resistor divider between CSCOMP and CSREF. Connecting LLSET to CSREF disables positioning.
Current Sense Reference Voltage Input. The voltage on this pin is used as the reference for the current sense
amplifier and the power-good and crowbar functions. This pin should be connected to the common point of
the output inductors.
CSREF
16
17
18
CSSUM
CSCOMP
GND
Current Sense Summing Node. External resistors from each switch node to this pin sum the average inductor
currents together to measure the total output current.
Current Sense Compensation Point. A resistor and capacitor from this pin to CSSUM determines the gain of the
current sense amplifier and the positioning loop response time.
Ground. All internal biasing and the logic output signals of the device are referenced to this ground.
Rev. 0 | Page 7 of 32
ADP3192A
Pin No. Mnemonic
Description
19
20
21
OD
Output Disable Logic Output. This pin is actively pulled low when the EN input is low or when VCC is below its
UVLO threshold to signal to the driver IC that the driver high-side and low-side outputs should go low.
Current Reference Input. An external resistor from this pin to ground sets the reference current for IFB, IDELAY, ISS,
IREF
I
ILIMIT, and ITTSENSE.
IMON
Analog Output. Represents the total load current.
22 to 25 SW4 to SW1
Current Balance Inputs. Inputs for measuring the current level in each phase. The SW pins of unused phases
should be left open.
26 NC
No Connection.
27 to 30 PWM4 to PWM1 Logic-Level PWM Outputs. Each output is connected to the input of an external MOSFET driver such as the
ADP3120A. Connecting the PWM4 and PWM3 outputs to VCC causes that phase to turn off, allowing the
ADP3192A to operate as a 2-, 3-, or 4-phase controller.
31
32 to 39 VID7 to VID0
40 VIDSEL
VCC
Supply Voltage for the Device. A 340 Ω resistor should be placed between the 12 V system supply and the VCC
pin. The internal shunt regulator maintains VCC = 5 V.
Voltage Identification DAC Inputs. These eight pins are pulled down to GND, providing a Logic 0 if left open. When
in normal operation mode, the DAC output programs the FB regulation voltage from 0.5 V to 1.6 V (see Table 4).
VID DAC Selection Pin. The logic state of this pin determines whether the internal VID DAC decodes VID0 to
VID7 as extended VR10 or VR11 inputs.
Rev. 0 | Page 8 of 32
ADP3192A
TYPICAL PERFORMANCE CHARACTERISTICS
7000
6000
5000
4000
MASTER CLOCK
3000
2000
PHASE 1
IN 4 PHASE DESIGN
1000
0
13
20 30 43 68 75 82 130 180 270 395 430 680 850
RT (kΩ)
Figure 6. Master Clock Frequency vs. RT
Rev. 0 | Page 9 of 32
ADP3192A
THEORY OF OPERATION
The ADP3192A combines a multimode, fixed frequency,
PWM control with multiphase logic outputs for use in 2-, 3-,
and 4-phase synchronous buck CPU core supply power
converters. The internal VID DAC is designed to interface
with the Intel 8-bit VRD/VRM 11-compatible CPU and 7-bit
VRD/VRM 10×-compatible CPU. Multiphase operation is
important for producing the high currents and low voltages
demanded by today’s microprocessors. Handling the high
currents in a single-phase converter places high thermal
demands on the components in the system, such as the
inductors and MOSFETs.
5V
UVLO
THRESHOLD
SUPPLY
VTT I/O
(ADP3192A EN)
0.85V
V
DELAY(TH)
(1.7V)
DELAY
V
BOOT
V
V
VID
VID
(1.1V)
1V
SS
TD3
V
BOOT
The multimode control of the ADP3192A ensures a stable,
high performance topology for the following:
(1.1V)
TD1
VCC_CORE
TD4
TD2
•
•
Balancing currents and thermals between phases
High speed response at the lowest possible switching
frequency and output decoupling
VR READY
(ADP3192A PWRGD)
TD5
50µs
•
Minimizing thermal switching losses by using lower
frequency operation
Tight load line regulation and accuracy
High current output due to 4-phase operation
Reduced output ripple due to multiphase cancellation
PC board layout noise immunity
Ease of use and design due to independent component
selection
Flexibility in operation for tailoring design to low cost or
high performance
CPU
VID INVALID
VID VALID
VID INPUTS
Figure 7. System Start-Up Sequence
•
•
•
•
•
PHASE DETECTION SEQUENCE
During startup, the number of operational phases and their
phase relationship is determined by the internal circuitry that
monitors the PWM outputs. Normally, the ADP3192A operates
as a 4-phase PWM controller. Connecting the PWM4 pin to
VCC programs 3-phase operation and connecting the PWM4
and PWM3 pins to VCC programs 2-phase operation.
•
Prior to soft start, while EN is low, the PWM3 and PWM4 pins
sink approximately 100 μA. An internal comparator checks each
pin’s voltage vs. a threshold of 3 V. If the pin is tied to VCC, it is
above the threshold. Otherwise, an internal current sink pulls
the pin to GND, which is below the threshold. PWM1 and
PWM2 are low during the phase detection interval that occurs
during the first four clock cycles of TD2. After this time, if the
remaining PWM outputs are not pulled to VCC, the 100 μA
current sink is removed, and they function as normal PWM
outputs. If they are pulled to VCC, the 100 μA current source is
removed, and the outputs are put into a high impedance state.
START-UP SEQUENCE
The ADP3192A follows the VR11 start-up sequence shown in
Figure 7. After both the EN and UVLO conditions are met,
the DELAY pin goes through one cycle (TD1). The first four
clock cycles of TD2 are blanked from the PWM outputs and
used for phase detection as explained in the Phase Detection
Sequence section. Then, the soft start ramp is enabled (TD2),
and the output comes up to the boot voltage of 1.1 V. The boot
hold time is determined by the DELAY pin as it goes through a
second cycle (TD3). During TD3, the processor VID pins settle
to the required VID code. When TD3 is over, the ADP3192A
soft starts either up or down to the final VID voltage (TD4).
After TD4 is complete and the PWRGD masking time (equal to
VID on-the-fly masking) is complete, a third ramp on the
DELAY pin sets the PWRGD blanking (TD5).
The PWM outputs are logic-level devices intended for driving
external gate drivers such as the ADP3120A. Because each
phase is monitored independently, operation approaching 100%
duty cycle is possible. In addition, more than one output can be
on at the same time to allow overlapping phases.
Rev. 0 | Page 10 of 32
ADP3192A
An additional resistor divider connected between CSREF and
CSCOMP (with the midpoint connected to LLSET) can be used
to set the load line required by the microprocessor. The current
information is then given as CSREF − LLSET. This difference
signal is used internally to offset the VID DAC for voltage
positioning. The difference between CSREF and CSCOMP is
then used as a differential input for the current-limit comparator.
This allows the load line to be set independently of the current-
limit threshold. In the event that the current-limit threshold
and load line are not independent, the resistor divider between
CSREF and CSCOMP can be removed, and the CSCOMP pin
can be directly connected to LLSET. To disable voltage position-
ing entirely (that is, no load line), connect LLSET to CSREF.
MASTER CLOCK FREQUENCY
The clock frequency of the ADP3192A is set with an external
resistor connected from the RT pin to GND. The frequency
follows the graph in Figure 6. To determine the frequency per
phase, the clock is divided by the number of phases in use. If all
phases are in use, divide by 4. If PWM4 is tied to VCC, divide
the master clock by 3 for the frequency of the remaining phases.
If PWM3 and PWM4 are tied to VCC, divide by 2.
OUTPUT VOLTAGE DIFFERENTIAL SENSING
The ADP3192A combines differential sensing with a high
accuracy VID DAC and reference, and a low offset error ampli-
fier. This maintains a worst-case specification of 7.7 mV
differential sensing error over its full operating output voltage
and temperature range. The output voltage is sensed between
the FB pin and FBRTN pin. FB should be connected through
a resistor to the regulation point, usually the remote sense pin
of the microprocessor. FBRTN should be connected directly
to the remote sense ground point. The internal VID DAC
and precision reference are referenced to FBRTN, which has a
minimal current of 65 μA to allow accurate remote sensing. The
internal error amplifier compares the output of the DAC to the
FB pin to regulate the output voltage.
To provide the best accuracy for sensing current, the CSA is
designed to have a low offset input voltage. Also, the sensing gain
is determined by external resistors to make it extremely accurate.
ACTIVE IMPEDANCE CONTROL MODE
For controlling the dynamic output voltage droop as a function
of output current, a signal proportional to the total output current
at the LLSET pin can be scaled to equal the regulator droop
impedance multiplied by the output current. This droop voltage
is then used to set the input control voltage to the system. The
droop voltage is subtracted from the DAC reference input
voltage to tell the error amplifier where the output voltage
should be. This allows enhanced feed-forward response.
OUTPUT CURRENT SENSING
The ADP3192A provides a dedicated current-sense amplifier
(CSA) to monitor the total output current for proper voltage
positioning vs. load current and for current-limit detection.
Sensing the load current at the output gives the total average
current being delivered to the load. This is an inherently more
accurate method than peak current detection or sampling the
current across a sense element such as the low-side MOSFET.
This amplifier can be configured several ways, depending on
the objectives of the system, as follows:
CURRENT CONTROL MODE AND THERMAL
BALANCE
The ADP3192A has individual inputs (SW1 to SW4) for each
phase that are used for monitoring the current of each phase.
This information is combined with an internal ramp to create
a current balancing feedback system that has been optimized for
initial current balance accuracy and dynamic thermal balancing
during operation. This current balance information is independent
of the average output current information used for positioning
as described in the Output Current Sensing section.
•
•
•
Output inductor DCR sensing without a thermistor for
lowest cost
Output inductor DCR sensing with a thermistor for
improved accuracy with tracking of inductor temperature
Sense resistors for highest accuracy measurements
The magnitude of the internal ramp can be set to optimize the
transient response of the system. It also monitors the supply
voltage for feed-forward control for changes in the supply. A
resistor connected from the power input voltage to the RAMPADJ
pin determines the slope of the internal PWM ramp. External
resistors can be placed in series with individual phases to create
an intentional current imbalance if desired, such as when one
phase has better cooling and can support higher currents.
Resistor RSW1 through Resistor RSW4 (see Figure 10) can be used
for adjusting thermal balance in this 4-phase example. It is best
to have the ability to add these resistors during the initial design;
therefore, ensure that placeholders are provided in the layout.
The positive input of the CSA is connected to the CSREF pin,
which is connected to the output voltage. The inputs to the
amplifier are summed together through resistors from the sensing
element, such as the switch node side of the output inductors,
to the inverting input CSSUM. The feedback resistor between
CSCOMP and CSSUM sets the gain of the amplifier and a filter
capacitor is placed in parallel with this resistor. The gain of the
amplifier is programmable by adjusting the feedback resistor.
Rev. 0 | Page 11 of 32
ADP3192A
To increase the current in any given phase, enlarge RSW for that
phase (make RSW = 0 for the hottest phase and do not change it
during balancing). Increasing RSW to only 500 Ω makes a substan-
tial increase in phase current. Increase each RSW value by small
amounts to achieve balance, starting with the coolest phase first.
FAST ENHANCED PWM MODE
Fast enhanced PWM mode (FEPWM) is intended to improve
the transient response of the ADP3192A to a load setup. In
previous generations of controllers, when a load step-up
occurred, the controller had to wait until the next turn-on of
the PWM signal to respond to the load change. Enhanced
PWM mode allows the controller to immediately respond when
a load step-up occurs. This allows the phases to respond more
quickly when a load increase takes place.
VOLTAGE CONTROL MODE
A high gain, high bandwidth, voltage mode error amplifier is
used for the voltage mode control loop. The control input voltage
to the positive input is set via the VID logic according to the
voltages listed in Table 4.
DELAY TIMER
This voltage is also offset by the droop voltage for active
positioning of the output voltage as a function of current,
commonly known as active voltage positioning. The output
of the amplifier is the COMP pin, which sets the termination
voltage for the internal PWM ramps.
The delay times for the start-up timing sequence are set with a
capacitor from the DELAY pin to GND. In UVLO, or when EN is
logic low, the DELAY pin is held at GND. After the UVLO and
EN signals are asserted, the first delay time (TD1 in Figure 7) is
initiated. A current flows out of the DELAY pin to charge CDLY
.
This current is equal to IREF, which is typically 15 μA. A compara-
tor monitors the DELAY voltage with a threshold of 1.7 V. The
delay time is therefore set by the IREF current charging a capacitor
from 0 V to 1.7 V. This DELAY pin is used for multiple delay
timings (TD1, TD3, and TD5) during the start-up sequence. In
addition, DELAY is used for timing the current-limit latch off,
as explained in the Current-Limit, Short-Circuit, and Latch-Off
Protection section.
The negative input (FB) is tied to the output sense location with
Resistor RB and is used for sensing and controlling the output
voltage at this point. A current source (equal to IREF) from the
FB pin flowing through RB is used for setting the no load offset
voltage from the VID voltage. The no load voltage is negative
with respect to the VID DAC. The main loop compensation is
incorporated into the feedback network between FB and COMP.
CURRENT REFERENCE
SOFT START
The IREF pin is used to set an internal current reference. This
reference current sets IFB, IDELAY, ISS, ILIMIT, and ITTSENSE. A resistor
to ground programs the current based on the 1.5 V output.
The soft start times for the output voltage are set with a
capacitor from the SS pin to GND. After TD1 and the phase
detection cycle are complete, the SS time (TD2 in Figure 7)
starts. The SS pin is disconnected from GND, and the capacitor
is charged up to the 1.1 V boot voltage by the SS amplifier,
which has an output current equal to IREF (typically 15 μA).
The voltage at the FB pin follows the ramping voltage on the
SS pin, limiting the inrush current during startup. The soft start
time depends on the value of the boot voltage and CSS.
1.5 V
IREF =
RIREF
Typically, RIREF is set to 100 kΩ to program IREF = 15 μA. The
following currents are then equal to
IFB = IREF = 15 μA
IDELAY = IREF = 15 μA
ISS = IREF = 15 μA
Once the SS voltage is within 100 mV of the boot voltage, the
boot voltage delay time (TD3 in Figure 7) is started. The end of
the boot voltage delay time signals the beginning of the second
soft start time (TD4 in Figure 7). The SS voltage now changes
from the boot voltage to the programmed VID DAC voltage
(either higher or lower) using the SS amplifier with the output
current equal to IREF. The voltage of the FB pin follows the
ramping voltage of the SS pin, limiting the inrush current
during the transition from the boot voltage to the final DAC
voltage. The second soft start time depends on the boot voltage,
the programmed VID DAC voltage, and CSS.
ILIMIT = 2/3 (IREF) = 10 μA
Rev. 0 | Page 12 of 32
ADP3192A
If EN is taken low or if VCC drops below UVLO, DELAY and
SS are reset to ground to be ready for another soft start cycle.
Figure 8 shows typical start-up waveforms for the ADP3192A.
The latch-off function can be reset by either removing and
reapplying the supply voltage to the ADP3192A or by toggling
the EN pin low for a short time. To disable the short-circuit
latch-off function, an external resistor should be placed in
parallel with CDLY. This prevents the DELAY capacitor from
charging up to the 1.7 V threshold. The addition of this resistor
causes a slight increase in the delay times.
1
2
During startup, when the output voltage is below 200 mV,
a secondary current limit is active. This is necessary because
the voltage swing of CSCOMP cannot go below GND. This
secondary current limit controls the internal COMP voltage
to the PWM comparators to 1.5 V. This limits the voltage drop
across the low-side MOSFETs through the current balance
circuitry. An inherent per-phase current limit protects
individual phases if one or more phases stop functioning
because of a faulty component. This limit is based on the
maximum normal mode COMP voltage. Typical overcurrent
latch-off waveforms are shown in Figure 9.
3
4
CH1 1V
CH3 1V
CH2 1V
CH4 10V
M
T
1ms
40.4%
A
CH1
700mV
Figure 8. Typical Start-Up Waveforms
(Channel 1: CSREF, Channel 2: DELAY,
Channel 3: SS, Channel 4: Phase 1 Switch Node)
CURRENT-LIMIT, SHORT-CIRCUIT, AND LATCH-
OFF PROTECTION
1
2
The ADP3192A compares a programmable current-limit
setpoint to the voltage from the output of the current-sense
amplifier. The level of current limit is set with the resistor
from the ILIMIT pin to GND. During operation, the current
from ILIMIT is equal to 2/3 of IREF, giving 10 μA typically.
This current through the external resistor sets the ILIMIT
voltage, which is internally scaled to give a current-limit
threshold of 82.6 mV/V. If the difference in voltage between
CSREF and CSCOMP rises above the current-limit threshold,
the internal current-limit amplifier controls the internal COMP
voltage to maintain the average output current at the limit.
3
4
CH1 1V
CH3 2V
CH2 1V
CH4 10V
M 2ms
T 61.8%
A CH1
680mV
Figure 9. Overcurrent Latch-Off Waveforms
(Channel 1: CSREF, Channel 2: DELAY,
Channel 3: COMP, Channel 4: Phase 1 Switch Node)
If the limit is reached and TD5 in Figure 7 is complete, a latch-
off delay time starts, and the controller shuts down if the fault is
not removed. The current-limit delay time shares the
DELAY pin timing capacitor with the start-up sequence timing.
However, during current limit, the DELAY pin current is
reduced to IREF/4. A comparator monitors the DELAY voltage
and shuts off the controller when the voltage reaches 1.7 V.
Therefore, the current-limit latch-off delay time is set by the
current of IREF/4 charging the delay capacitor from 0 V to 1.7 V.
This delay is four times longer than the delay time during the
start-up sequence.
DYNAMIC VID
The ADP3192A has the ability to dynamically change the VID
inputs while the controller is running. This allows the output
voltage to change while the supply is running and supplying
current to the load, which is commonly referred to as VID on-
the-fly (OTF). A VID OTF can occur under light or heavy load
conditions. The processor signals the controller by changing the
VID inputs in multiple steps from the start code to the finish
code. This change can be positive or negative.
The current-limit delay time starts only after the TD5 is
complete. If there is a current limit during startup, the
ADP3192A goes through TD1 to TD5, and then starts the
latch-off time. Because the controller continues to cycle the
phases during the latch-off delay time, the controller returns to
normal operation and the DELAY capacitor is reset to GND if
the short is removed before the 1.7 V threshold is reached.
When a VID input changes state, the ADP3192A detects the
change and ignores the DAC inputs for a minimum of 400 ns.
This time prevents a false code due to logic skew while the eight
VID inputs are changing. Additionally, the first VID change
initiates the PWRGD and crowbar blanking functions for a
minimum of 100 μs to prevent a false PWRGD or crowbar
event. Each VID change resets the internal timer.
Rev. 0 | Page 13 of 32
ADP3192A
POWER-GOOD MONITORING
OUTPUT ENABLE AND UVLO
The power-good comparator monitors the output voltage via
the CSREF pin. The PWRGD pin is an open-drain output whose
high level, when connected to a pull-up resistor, indicates that
the output voltage is within the specified nominal limits based
on the VID voltage setting. PWRGD goes low if the output
voltage is outside of this specified range, if the VID DAC inputs
are in no CPU mode, or if the EN pin is pulled low. PWRGD is
blanked during a VID OTF event for a period of 200 μs to
prevent false signals during the time the output is changing.
For the ADP3192A to begin switching, the input supply (VCC)
to the controller must be higher than the UVLO threshold and
the EN pin must be higher than its 0.85 V threshold. This
initiates a system start-up sequence. If either UVLO or EN is
less than their respective thresholds, the ADP3192A is disabled.
This holds the PWM outputs at ground, shorts the DELAY
OD
capacitor to ground, and forces PWRGD and
signals low.
OD
In the application circuit (see Figure 10), the
pin should be
OD
connected to the
inputs of the ADP3120A drivers.
The PWRGD circuitry also incorporates an initial turn-on
delay time (TD5) based on the DELAY timer. Prior to the
SS voltage reaching the programmed VID DAC voltage and the
PWRGD masking-time finishing, the PWRGD pin is held low.
Once the SS pin is within 100 mV of the programmed DAC
voltage, the capacitor on the DELAY pin begins to charge.
A comparator monitors the DELAY voltage and enables
PWRGD when the voltage reaches 1.7 V. The PWRGD delay
time is set, therefore, by a current of IREF, charging a capacitor
from 0 V to 1.7 V.
OD
Grounding
disables the drivers such that both DRVH and
DRVL are grounded. This feature is important in preventing the
discharge of the output capacitors when the controller is shut
off. If the driver outputs are not disabled, a negative voltage can
be generated during output due to the high current discharge of
the output capacitors through the inductors.
THERMAL MONITORING
The ADP3192A includes a thermal monitoring circuit to detect
when a point on the VR has exceeded two different user-defined
temperatures. The thermal monitoring circuit requires an NTC
thermistor to be placed between TTSENSE and GND.
OUTPUT CROWBAR
To protect the load and output components of the supply, the
PWM outputs are driven low, which turns on the low-side
MOSFETs when the output voltage exceeds the upper crowbar
threshold. This crowbar action stops once the output voltage
falls below the release threshold of approximately 375 mV.
A fixed current of 8 × IREF (typically giving 123 μA) is sourced
out of the TTSENSE pin and into the thermistor. The current
source is internally limited to 5 V. An internal circuit compares
the TTSENSE voltage to a 1.105 V and a 0.81 V threshold, and
outputs an open-drain signal at the VRFAN and VRHOT
outputs, respectively. Once the voltage on the TTSENSE pin
drops below its respective threshold, the open-drain outputs
assert high to signal the system that an overtemperature event
has occurred. Because the TTSENSE voltage changes slowly
with respect to time, 50 mV of hysteresis is built into these com-
parators. The thermal monitoring circuitry does not depend on
EN and is active when UVLO is above its threshold. When UVLO
is below its threshold, VRFAN and VRHOT are forced low.
Turning on the low-side MOSFETs pulls down the output as
the reverse current builds up in the inductors. If the output
overvoltage is due to a short in the high-side MOSFET, this
action current limits the input supply or blows its fuse,
protecting the microprocessor from being destroyed.
Table 4.VR11 and VR10.x VID Codes for the ADP3192A
VR11 DAC CODES: VIDSEL = HIGH
VR10.x DAC CODES: VIDSEL = LOW
OUTPUT
OFF
OFF
VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VID4 VID3 VID2 VID1 VID0 VID5 VID6
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
N/A
N/A
1
1.60000
1.59375
1.58750
1.58125
1.57500
1.56875
1.56250
1.55625
1.55000
1.54375
1.53750
1.53125
1.52500
1.51875
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
1
1
1
1
0
0
0
0
1
1
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
1
1
1
1
0
0
0
0
0
0
0
0
Rev. 0 | Page 14 of 32
ADP3192A
VR11 DAC CODES: VIDSEL = HIGH
VR10.x DAC CODES: VIDSEL = LOW
OUTPUT
1.51250
1.50625
1.50000
1.49375
1.48750
1.48125
1.47500
1.46875
1.46250
1.45625
1.45000
1.44375
1.43750
1.43125
1.42500
1.41875
1.41250
1.40625
1.40000
1.39375
1.38750
1.38125
1.37500
1.36875
1.36250
1.35625
1.35000
1.34375
1.33750
1.33125
1.32500
1.31875
1.31250
1.30625
1.30000
1.29375
1.28750
1.28125
1.27500
1.26875
1.26250
1.25625
1.25000
1.24375
1.23750
1.23125
1.22500
1.21875
1.21250
1.20625
1.20000
1.19375
1.18750
1.18125
1.17500
1.16875
1.16250
1.15625
1.15000
VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VID4 VID3 VID2 VID1 VID0 VID5 VID6
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
Rev. 0 | Page 15 of 32
ADP3192A
VR11 DAC CODES: VIDSEL = HIGH
VR10.x DAC CODES: VIDSEL = LOW
OUTPUT
1.14375
1.13750
1.13125
1.12500
1.11875
1.11250
1.10625
1.10000
1.09375
OFF
VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VID4 VID3 VID2 VID1 VID0 VID5 VID6
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
0
0
0
0
0
1
1
1
1
0
0
0
0
1
0
0
1
1
0
0
1
1
1
0
1
0
1
0
1
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
N/A
OFF
OFF
OFF
N/A
N/A
N/A
1.08750
1.08125
1.07500
1.06875
1.06250
1.05625
1.05000
1.04375
1.03750
1.03125
1.02500
1.01875
1.01250
1.00625
1.00000
0.99375
0.98750
0.98125
0.97500
0.96875
0.96250
0.95625
0.95000
0.94375
0.93750
0.93125
0.92500
0.91875
0.91250
0.90625
0.90000
0.89375
0.88750
0.88125
0.87500
0.86875
0.86250
0.85625
0.85000
0.84375
0.83750
0.83125
0.82500
0.81875
0.81250
0.80625
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
N/A
N/A
N/A
N/A
Rev. 0 | Page 16 of 32
ADP3192A
VR11 DAC CODES: VIDSEL = HIGH
VR10.x DAC CODES: VIDSEL = LOW
OUTPUT
0.80000
0.79375
0.78750
0.78125
0.77500
0.76875
0.76250
0.75625
0.75000
0.74375
0.73750
0.73125
0.72500
0.71875
0.71250
0.70625
0.70000
0.69375
0.68750
0.68125
0.67500
0.66875
0.66250
0.65625
0.65000
0.64375
0.63750
0.63125
0.62500
0.61875
0.61250
0.60625
0.60000
0.59375
0.58750
0.58125
0.57500
0.56875
0.56250
0.55625
0.55000
0.54375
0.53750
0.53125
0.52500
0.51875
0.51250
0.50625
0.50000
OFF
VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VID4 VID3 VID2 VID1 VID0 VID5 VID6
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
1
1
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
0
1
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
1
1
1
1
1
1
1
1
1
1
1
1
0
1
OFF
Rev. 0 | Page 17 of 32
ADP3192A
0 0 9 6 - 7 8 0 6
V C C
D 7 V I
F E I R
O D
D 6 V I
D 5 V I
D 4 V I
D 3 V I
D 2 V I
D 1 V I
D 0 V I
D
G N
P M
C S C O
C S S U M
C S R E F
E T L L S
A P D J R A M
R T
D S V E I L
I T M I I L
Figure 10. Typical 4-Phase Application Circuit
Rev. 0 | Page 18 of 32
ADP3192A
APPLICATION INFORMATION
The design parameters for a typical Intel VRD 11-compliant
CPU application are as follows:
CURRENT-LIMIT LATCH-OFF DELAY TIMES
The start-up and current-limit delay times are determined by
the capacitor connected to the DELAY pin. The first step is to
set CDLY for the TD1, TD3, and TD5 delay times (see Figure 7).
The DELAY ramp (IDELAY)is generated using a 15 μA internal
current source. The value for CDLY can be approximated using
•
•
•
•
•
•
Input voltage (VIN) = 12 V
VID setting voltage (VVID) = 1.300 V
Duty cycle (D) = 0.108
Nominal output voltage at no load (VONL) = 1.285 V
Nominal output voltage at 115 A load (VOFL) = 1.170 V
Static output voltage drop based on a 1.0 mΩ load line (RO)
TD(x)
VDELAY(TH)
CDLY = IDELAY
×
(3)
from no load to full load (VD) = VONL − VOFL
1.285 V − 1.170 V = 115 mV
=
where:
TD(x) is the desired delay time for TD1, TD3, and TD5.
VDELAY(TH), the DELAY threshold voltage, is given as 1.7 V.
•
•
•
•
•
Maximum output current (IO) = 130 A
Maximum output current step (ΔIO) = 100 A
Maximum output current slew rate (SR) = 200 A/μs
Number of phases (n) = 4
In this example, 2 ms is chosen for all three delay times, which
meets Intel specifications. Solving for CDLY gives a value of
17.6 nF. The closest standard value for CDLY is 18 nF.
Switching frequency per phase (fSW) = 330 kHz
When the ADP3192A enters current limit, the internal current
source changes from 15 μA to 3.75 μA. This makes the latch-off
delay time four times longer than the start-up delay time.
Longer latch-off delay times can be achieved by placing a
SETTING THE CLOCK FREQUENCY
The ADP3192A uses a fixed frequency control architecture. The
frequency is set by an external timing resistor (RT). The clock
frequency and the number of phases determine the switching
frequency per phase, which relates directly to switching losses
as well as the sizes of the inductors, the input capacitors, and
output capacitors. With n = 4 for four phases, a clock frequency
of 1.32 MHz sets the switching frequency (fSW) of each phase to
330 kHz, which represents a practical trade-off between the
switching losses and the sizes of the output filter components.
Figure 6 shows that to achieve a 1.32 MHz oscillator frequency,
the correct value for RT is 130 kΩ. Alternatively, the value for RT
can be calculated using
resistor in parallel with CDLY
.
INDUCTOR SELECTION
The choice of inductance for the inductor determines the ripple
current in the inductor. Less inductance leads to more ripple
current, which increases the output ripple voltage and conduction
losses in the MOSFETs. However, using smaller inductors
allows the converter to meet a specified peak-to-peak transient
deviation with less total output capacitance. Conversely, a higher
inductance means lower ripple current and reduced conduction
losses, but more output capacitance is required to meet the
same peak-to-peak transient deviation.
1
(1)
RT
=
n × fSW × 6 pF
In any multiphase converter, a practical value for the peak-to-
peak inductor ripple current is less than 50% of the maximum
dc current in the same inductor. Equation 4 shows the
relationship between the inductance, oscillator frequency, and
peak-to-peak ripple current in the inductor.
where 6 pF is the internal IC component values. For good initial
accuracy and frequency stability, a 1% resistor is recommended.
SOFT START DELAY TIME
The value of CSS sets the soft start time. The ramp is generated
with a 15 μA internal current source. The value for CSS can be
found using
VVID
×
(
1− D
)
(4)
IR
=
fSW × L
TD2
VBOOT
(2)
CSS =15μA×
Equation 5 can be used to determine the minimum inductance
based on a given output ripple voltage.
where:
VVID × RO
×
(
1−
fSW ×VRIPPLE
Solving Equation 5 for an 8 mV p-p output ripple voltage yields
1.3 V ×1.0 mꢀ × 1−0.432
(n×D)
)
(5)
L ≥
TD2 is the desired soft start time.
VBOOT is internally set to 1.1 V.
Assuming a desired TD2 time of 3 ms, CSS is 41 nF. The closest
standard value for CSS is 39 nF. Although CSS also controls the
time delay for TD4 (determined by the final VID voltage), the
minimum specification for TD4 is 0 ns. This means that as long as
the TD2 time requirement is met, TD4 is within the specification.
(
)
L ≥
= 280 nH
330 kHz × 8 mV
Rev. 0 | Page 19 of 32
ADP3192A
If the resulting ripple voltage is less than what is designed for,
the inductor can be made smaller until the ripple value is met.
This allows optimal transient response and minimum output
decoupling.
Selecting a Standard Inductor
The following power inductor manufacturers can provide design
consultation and deliver power inductors optimized for high
power applications upon request:
The smallest possible inductor should be used to minimize
the number of output capacitors. For this example, choosing a
320 nH inductor is a good starting point and gives a calculated
ripple current of 11 A. The inductor should not saturate at the
peak current of 35.5 A and should be able to handle the sum of
the power dissipation caused by the average current of 30 A in
the winding and core loss.
•
•
•
Coilcraft®
Coiltronics®
Sumida Corporation®
CURRENT SENSE AMPLIFIER
Most designs require the regulator output voltage, measured at
the CPU pins, to drop when the output current increases. The
specified voltage drop corresponds to a dc output resistance (RO),
also referred to as a load line. The ADP3192A has the flexibility
of adjusting RO, independent of current-limit or compensation
components, and it can also support CPUs that do not require
a load line.
Another important factor in the inductor design is the dc
resistance (DCR), which is used for measuring the phase currents.
A large DCR can cause excessive power loss, though too small a
value can lead to increased measurement error. A good rule is
to have the DCR (RL) be about 1 to 1½ times the droop resistance
(RO). This example uses an inductor with a DCR of 1.4 mΩ.
For designs requiring a load line, the impedance gain of the
CS amplifier (RCSA) must be to be greater than or equal to the load
line. All designs, whether they have a load line or not, should
keep RCSA ≥ 1 mΩ.
Designing an Inductor
Once the inductance and DCR are known, the next step is to
either design an inductor or find a standard inductor that comes
as close as possible to meeting the overall design goals. It is also
important to have the inductance and DCR tolerance specified
to control the accuracy of the system. Reasonable tolerances
most manufacturers can meet are 15% inductance and 7% DCR
at room temperature. The first decision in designing the inductor
is choosing the core material. Several possibilities for providing
low core loss at high frequencies include the powder cores (from
Micrometals, Inc., for example, or Kool Mu® from MAGNETICS®)
and the gapped soft ferrite cores (for example, 3F3 or 3F4 from
Philips®). Low frequency powdered iron cores should be avoided
due to their high core loss, especially when the inductor value is
relatively low, and the ripple current is high.
The output current is measured by summing the voltage across
each inductor and passing the signal through a low-pass filter.
This summer filter is the CS amplifier configured with resistors
R
PH(X) (summers), and RCS and CCS (filter). The impedance gain
of the regulator is set by the following equations, where RL is the
DCR of the output inductors:
RCS
RPH
RCSA
=
× RL
(6)
(7)
(X )
L
CCS
=
RL × RCS
The user has the flexibility to choose either RCS or RPH(X). However,
it is best to select RCS equal to 100 kΩ, and then solve for RPH(X)
by rearranging Equation 6. Here, RCSA = RO = 1 mΩ because this
is equal to the design load line.
The best choice for a core geometry is a closed-loop type, such
as a potentiometer core (PQ, U, or E core) or toroid. A good
compromise between price and performance is a core with
a toroidal shape.
RL
RCSA
RPH
=
× RCS
Many useful magnetics design references are available for
quickly designing a power inductor, such as
(
X
)
•
•
Intusoft Magnetics Designer Software
1.4 mꢀ
1.0 mꢀ
RPH
=
×100 kꢀ =140 kꢀ
(
X
)
Designing Magnetic Components for High Frequency dc-dc
Converters, by Colonel Wm. T. McLyman,
Kg Magnetics, Inc., ISBN 1883107008
Next, use Equation 7 to solve for CCS.
320 nH
CCS
=
= 2.28 nF
1.4 mꢀ ×100 kꢀ
Rev. 0 | Page 20 of 32
ADP3192A
It is best to have a dual location for CCS in the layout so that
standard values can be used in parallel to get as close to the
desired value. For best accuracy, CCS should be a 5% or 10%
NPO capacitor. This example uses a 5% combination for CCS
of two 1 nF capacitors in parallel. Recalculating RCS and RPH(X)
using this capacitor combination yields 114 kΩ and 160 kΩ,
respectively. The closest standard 1% value for RPH(X) is 158 kΩ.
4. Compute the relative values for RCS1, RCS2, and RTH using
(
A − B
)
× r1 × r2 − A ×
A × 1− B × r1 − B ×
1− A
(
1− B
)
× r2 + B ×
× r2 −
(
1− A
)
× r1
)
rCS2
=
=
(8)
(9)
(
)
(
1− A
)
(
A − B
(
)
rCS1
1
A
−
1−rCS2 r1 −rCS2
INDUCTOR DCR TEMPERATURE CORRECTION
1
rTH
=
(10)
1
1
When the inductor DCR is used as the sense element and copper
wire is used as the source of the DCR, the user needs to compen-
sate for temperature changes of the inductor’s winding. Fortunately,
copper has a well-known temperature coefficient (TC) of 0.39%/°C.
−
1−rCS2 rCS1
5. Calculate RTH = rTH × RCS, then select the closest value of
thermistor available. Also, compute a scaling factor (k)
based on the ratio of the actual thermistor value used
relative to the computed one.
If RCS is designed to have an opposite and equal percentage
change in resistance to that of the wire, it cancels the temperature
variation of the inductor DCR. Due to the nonlinear nature of
NTC thermistors, Resistor RCS1 and Resistor RCS2 are needed.
Refer to Figure 11 to linearize the NTC and produce the desired
temperature tracking.
RTH
(
ACTUAL)
k =
(11)
RTH
(
CALCULATED
)
6. Calculate values for RCS1 and RCS2 using Equation 12 and
Equation 13.
PLACE AS CLOSE AS POSSIBLE
TO NEAREST INDUCTOR
TO
SWITCH
NODES
TO
OR LOW-SIDE MOSFET
VOUT
SENSE
RCS1 = RCS × k × rCS1
RCS2 = RCS 1−k
(12)
(13)
R
TH
×
)
+
k × rCS2
R
R
R
PH3
PH1
PH2
ADP3192A
In this example, RCS is calculated to be 114 kΩ. Look for an
available 100 kΩ thermistor, 0603 size. One such thermistor
is the Vishay NTHS0603N01N1003JR NTC thermistor with
A = 0.3602 and B = 0.09174. From these values, rCS1 = 0.3795,
rCS2 = 0.7195, and rTH = 1.075.
R
R
CS2
CS1
CSCOMP
17
C
C
CS2
KEEP THIS PATH
CS1
AS SHORT AS POSSIBLE
AND WELL AWAY FROM
SWITCH NODE LINES
CSSUM
CSREF
16
Solving for RTH yields 122.55 kΩ, so 100 kΩ is chosen, making
k = 0.816. Next, find RCS1 and RCS2 to be 35.3 kΩ and 87.9 kΩ.
Finally, choose the closest 1% resistor value, which yields a
choice of 35.7 kΩ and 88.7 kΩ.
15
Figure 11. Temperature Compensation Circuit Values
The following procedure and equations yield values to use for RCS1,
RCS2, and RTH (the thermistor value at 25°C) for a given RCS value:
Load Line Setting
For load line values greater than 1 mΩ, RCSA can be set equal
to RO, and the LLSET pin can be directly connected to the
CSCOMP pin. When the load line value needs to be less than
1 mΩ, two additional resistors are required. Figure 12 shows
the placement of these resistors.
1. Select an NTC based on type and value. Because the value
is unknown, use a thermistor with a value close to RCS. The
NTC should also have an initial tolerance of better than 5%.
2. Based on the type of NTC, find its relative resistance value at
two temperatures. Temperatures that work well are 50°C and
90°C. These resistance values are called A (RTH(50°C))/RTH(25°C)
and B (RTH(90°C))/RTH(25°C)). The relative value of the NTC is
always 1 at 25°C.
)
ADP3192A
CSCOMP
17
3. Find the relative value of RCS required for each of these
temperatures. This is based on the percentage change
needed, which in this example is initially 0.39%/°C. These
temperatures are called r1 (1/(1 + TC × (T1 − 25°C))) and
r2 (1/(1 + TC × (T2 − 25°C))), where TC = 0.0039 for copper,
T1 = 50°C, and T2 = 90°C. From this, r1 = 0.9112 and
r2 = 0.7978.
CSSUM
16
CSREF
15
R
R
LL2
LL1
OPTIONAL LOAD LINE
SELECT SWITCH
LLSET
14
Q
LL
Figure 12. Load Line Setting Resistors
Rev. 0 | Page 21 of 32
ADP3192A
The two resistors RLL1 and RLL2 set up a divider between the
CSCOMP pin and CSREF pin. This resistor divider is input into
the LLSET pin to set the load line slope RO of the VR according
to the following equation:
OUTPUT OFFSET
The Intel specification requires that at no load, the nominal output
voltage of the regulator be offset to a value lower than the nominal
voltage corresponding to the VID code. The offset is set by a
constant current source flowing out of the FB pin (IFB) and flowing
through RB. The value of RB can be found using Equation 19.
RLL2
RO
=
×RCSA
(14)
R
LL1 + RLL2
The resistor values for RLL1 and RLL2 are limited by two factors.
VVID −VONL
RB
RB
=
=
IFB
•
The minimum value is based upon the loading of the
CSCOMP pin. This pin’s drive capability is 500 ꢁA, and the
majority of this should be allocated to the CSA feedback. If
the current through RLL1 and RLL2 is limited to 10% of this
drive capability (50 ꢁA), the following limit can be placed
1.3 V−1.285 V
=1.00 kꢀ
(19)
15 ꢁA
The closest standard 1% resistor value is 1.00 kΩ.
on the minimum value for RLL1 and RLL2
:
COUT SELECTION
I
LIM ×RCSA
50×10−6
The required output decoupling for the regulator is typically
recommended by Intel for various processors and platforms.
R
LL1 + RLL2
≥
(15)
Use some simple design guidelines to determine the require-
ments. These guidelines are based on having both bulk
capacitors and ceramic capacitors in the system.
Here, ILIM is the current-limit current, which is the
maximum signal level that the CSA responds to.
•
The maximum value is based upon minimizing induced
dc offset errors based on the bias current of the LLSET pin.
To keep the induced dc error less than 1 mV, which makes
this error statistically negligible, place the following limit
First, select the total amount of ceramic capacitance. This is
based on the number and type of capacitor to be used. The best
location for ceramic capacitors is inside the socket with 12 to 18,
1206 size being the physical limit. Other capacitors can be
placed along the outer edge of the socket as well.
on the parallel combination of RLL1 and RLL2
:
R
R
LL1 × RLL2
1×10−3
To determine the minimum amount of ceramic capacitance
required, start with a worst-case load step occurring right after
a switching cycle stops. The ceramic capacitance then delivers
the charge to the load while the load is ramping up and until the
VR has responded with the next switching cycle.
= 8.33 kΩ
(16)
≤
LL1 + RLL2 120 ×10−9
Select minimum value resistors to reduce the noise and parasitic
susceptibility of the feedback path.
By combining Equation 16 with Equation 14 and selecting
minimum values for the resistors, the following equations result:
Equation 20 gives the designer a rough approximation for
determining the minimum ceramic capacitance. Due to the
complexity of the PCB parasitics and bulk capacitors, the actual
amount of ceramic capacitance required can vary.
ILIM × RO
50μA
(17)
RLL2
=
⎡
⎤
Δ IO
2 SR
⎛
⎜
⎜
⎝
⎞
RCSA
RO
1
1
1
⎛
⎜
⎞
⎠
(20)
⎟
CZ
≥
×
×
− D −
RLL1
=
−1 ×RLL2
(18)
⎟
(
MIN
)
⎢
⎣
⎥
⎦
⎟
RO
fSW
n
⎝
⎠
The typical ceramic capacitors consist of multiple 10 μF or 22 μF
capacitors. For this example, Equation 20 yields 180.8 μF, therefore,
18, 10 ꢁF ceramic capacitors suffice.
Therefore, both RLL1 and RLL2 need to be in parallel and less than
8.33 kꢀ.
Another useful feature for some VR applications is the ability to
select different load lines. Figure 12 shows an optional MOSFET
switch that allows this feature. Here, design for RCSA = RO(MAX)
(selected with QLL on), and then use Equation 14 to set RO = RO(MIN)
(selected with QLL off).
Next, an upper limit is imposed on the total amount of bulk
capacitance (CX) when the user considers the VID on-the-fly
voltage stepping of the output (voltage step VV in time tV with
error of VERR).
A lower limit is based on meeting the capacitance for load
release for a given maximum load step (ꢂIO) and a maximum
allowable overshoot. The total amount of load release voltage
is given as ΔVO = ΔIO × RO + ΔVrl, where ΔVrl is the maximum
allowable overshoot voltage.
For this design, RCSA = RO = 1 mΩ. As a result, connect LLSET
directly to CSCOMP; the RLL1 and RLL2 resistors are not needed.
Rev. 0 | Page 22 of 32
ADP3192A
This is tested using
LX ≤ CZ × RO 2 × Q2
⎛
⎜
⎞
⎟
⎜
⎜
⎜
⎟
⎟
⎟
L × ꢂIO
ΔVrl
CX
CX
≥
−CZ
(21)
(
(
MIN )
(23)
4
3
⎛
⎞
⎟
⎟
⎠
2
LX ≤ 180 ꢁF ×
(
1 mꢀ
)
×
= 240 pH
⎜
n × RO +
×VVID
⎜
⎜
⎝
⎟
⎠
ΔIO
⎝
where Q2 is limited to 4/3 to ensure a critically damped system.
≤
×
MAX
)
In this example, LX is approximately 240 pH for the 10, Al-Poly
capacitors, which satisfies this limitation. If the LX of the chosen
bulk capacitor bank is too large, the number of ceramic capacitors
needs to be increased, or lower ESL bulks need to be used if
there is excessive undershoot during a load transient.
2
⎛
⎞
⎛
⎞
⎟
⎟
⎠
VV
VVID nKRO
L
⎜
⎟
⎜
×
1+ tV
×
− 1 −CZ (22)
nK 2RO2 VVID
VV
L
⎜
⎜
⎝
⎟
⎜
⎝
⎟
⎠
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
VERR
VV
where K = −1n
.
For this multimode control technique, all ceramic designs can
be used providing the conditions of Equation 20 through
Equation 23 are satisfied.
To meet the conditions of these equations and transient response,
the ESR of the bulk capacitor bank (RX) should be less than two
times the droop resistance (RO). If the CX(MIN) is larger than
POWER MOSFETS
For this example, the N-channel power MOSFETs have been
selected for one high-side switch and two low-side switches per
phase. The main selection parameters for the power MOSFETs
are VGS(TH), QG, CISS, CRSS, and RDS(ON). The minimum gate drive
voltage (the supply voltage to the ADP3120A) dictates whether
standard threshold or logic-level threshold MOSFETs must be
used. With VGATE ~10 V, logic-level threshold MOSFETs
(VGS(TH) < 2.5 V) are recommended.
C
X(MAX), the system cannot meet the VID on-the-fly specification
and may require the use of a smaller inductor or more phases
(and may have to increase the switching frequency to keep the
output ripple the same).
This example uses 18, 10 μF 1206 MLC capacitors (CZ = 180 μF).
The VID on-the-fly step change is 450 mV in 230 μs with a
settling error of 2.5 mV. The maximum allowable load release
overshoot for this example is 50 mV. Therefore, solving for the
bulk capacitance yields the following:
The maximum output current (IO) determines the RDS(ON)
requirement for the low-side (synchronous) MOSFETs. With
the ADP3192A, currents are balanced between phases, thus, the
current in each low-side MOSFET is the output current divided
by the total number of MOSFETs (nSF). With conduction losses
being dominant, Equation 24 shows the total power that is
dissipated in each synchronous MOSFET in terms of the ripple
current per phase (IR) and average total output current (IO).
CX
≤
MIN )
(
⎛
⎜
⎞
⎟
⎜
⎜
⎜
⎟
320 nH ×100 A
⎟
−180 ꢁF = 3.92 mF
⎛
⎜
⎝
⎞
⎟
50 mV
100 A
⎜
⎟
⎟
⎠
4 × 1.0 mꢀ +
×1.3 V
⎜
⎜
⎝
⎟
⎟
⎠
2
2
⎡
⎢
⎤
⎥
n I
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
IO
nSF
1
12
R
PSF
=
(
1− D
)
×
+
×
×RDS
(24)
(
SF
)
320 nH × 450 mV
nSF
⎢
⎣
⎥
⎦
CX
≤
MAX
)
×
(
4 × 5.22 ×
(
1.0 mꢀ
)
×1.3 V
2
Knowing the maximum output current being designed for and
the maximum allowed power dissipation, the user can find the
required RDS(ON) for the MOSFET. For D-PAK MOSFETs up to
an ambient temperature of 50°C, a safe limit for PSF is 1 W to 1.5 W
at a 120°C junction temperature. Therefore, for this example
(119 A maximum), RDS(SF) (per MOSFET) < 7.5 mΩ. This RDS(SF)
is also at a junction temperature of about 120°C. As a result,
users need to account for this when making this selection. This
example uses two lower-side MOSFETs at 4.8 mΩ, each at 120°C.
2
⎛
⎜
⎞
⎟
⎛
⎞
⎟
⎟
⎠
230 ꢁs ×1.3 V × 4 × 5.2 ×1.0 mꢀ
450 mV × 320 nH
⎜
⎜
⎝
1+
−1 −180 ꢁF
⎜
⎟
⎜
⎟
⎠
⎝
= 43.0 mF
where K = 5.2.
Using 10, 560 μF Al-Poly capacitors with a typical ESR of 6 mΩ
each yields CX = 5.6 mF with an RX = 0.6 mΩ.
One last check should be made to ensure that the ESL of the
bulk capacitors (LX) is low enough to limit the high frequency
ringing during a load change.
Another important factor for the synchronous MOSFET is the
input capacitance and feedback capacitance. The ratio of the
feedback to input needs to be small (less than 10% is recom-
mended) to prevent accidental turn-on of the synchronous
MOSFETs when the switch node goes high.
Rev. 0 | Page 23 of 32
ADP3192A
Also, the time to switch the synchronous MOSFETs off should
not exceed the nonoverlap dead time of the MOSFET driver
(40 ns typical for the ADP3120A). The output impedance of
the driver is approximately 2 Ω, and the typical MOSFET input
gate resistances are about 1 Ω to 2 Ω. Therefore, a total gate
capacitance of less than 6000 pF should be adhered to. Because
two MOSFETs are in parallel, the input capacitance for each
synchronous MOSFET should be limited to 3000 pF.
Finally, consider the power dissipation in the driver for each
phase. This is best expressed as QG for the MOSFETs and is
given by Equation 27.
⎡
⎢
⎤
fSW
2 ×n
PDRV
=
×
(
nMF ×QGMF + nSF ×QGSF
)
+ ICC ×V (27)
⎥
CC
⎢
⎣
⎥
⎦
where:
QGMF is the total gate charge for each main MOSFET.
The high-side (main) MOSFET has to be able to handle two main
power dissipation components: conduction and switching losses.
The switching loss is related to the amount of time it takes for
the main MOSFET to turn on and off, and to the current and
voltage that are being switched. Basing the switching speed on
the rise and fall time of the gate driver impedance and MOSFET
input capacitance, Equation 25 provides an approximate value
for the switching loss per main MOSFET.
Q
GSF is the total gate charge for each synchronous MOSFET.
Also shown is the standby dissipation factor (ICC × VCC) of the
driver. For the ADP3120A, the maximum dissipation should be
less than 400 mW. In this example, with ICC = 7 mA, QGMF = 5.8 nC,
and QGSF = 48 nC, there is 297 mW in each driver, which is
below the 400 mW dissipation limit. See the ADP3120A data
sheet for more details.
RAMP RESISTOR SELECTION
VCC × IO
nMF
n
PS
= 2 × fSW
MF )
×
× RG ×
×CISS
(25)
(
The ramp resistor (RR) is used for setting the size of the internal
PWM ramp. The value of this resistor is chosen to provide the best
combination of thermal balance, stability, and transient response.
Equation 28 is used for determining the optimum value.
nMF
where:
MF is the total number of main MOSFETs.
RG is the total gate resistance (2 Ω for the ADP3120A and about
1 Ω for typical high speed switching MOSFETs, making RG = 3 Ω).
n
AR × L
RR =
3 × AD × RDS ×CR
C
ISS is the input capacitance of the main MOSFET.
(28)
Adding more main MOSFETs (nMF) does not help the switching
loss per MOSFET because the additional gate capacitance slows
switching. Use lower gate capacitance devices to reduce
switching loss.
0.2 × 320 nH
RR =
= 356 kꢀ
3 × 5 × 2.4 mꢀ × 5 pF
where:
The conduction loss of the main MOSFET is given by
AR is the internal ramp amplifier gain.
AD is the current balancing amplifier gain.
RDS is the total low-side MOSFET on resistance.
CR is the internal ramp capacitor value.
2
⎛ n × IR ⎞2
⎡
⎢
⎤
I
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
O
1
12
⎜
⎜
⎝
⎟ ⎥
× RDS
PC
= D ×
+
×
(26)
(
MF
)
(
MF
)
⎟
nMF
nMF
⎠
⎢
⎣
⎥
⎦
where RDS(MF) is the on resistance of the MOSFET.
The internal ramp voltage magnitude can be calculated by using
Typically, for main MOSFETs, the highest speed (low CISS)
AR ×
(
1− D
)×VVID
VR =
VR =
device is preferred, but these usually have higher on resistance.
Select a device that meets the total power dissipation (about
1.5 W for a single D-PAK) when combining the switching and
conduction losses.
RR × CR × fSW
(29)
0.2 ×
1−0.108 ×1.3 V
( )
= 394 mV
357 kꢀ × 5 pF × 330 kHz
For this example, an NTD40N03 is selected as the main
MOSFET (eight total; nMF = 8), with CISS = 584 pF (maximum)
and RDS(MF) = 19 mΩ (maximum at TJ = 120°C). An NTD110N02is
selected as the synchronous MOSFET (eight total; nSF = 8), with
The size of the internal ramp can be made larger or smaller.
If it is made larger, stability and noise rejection improves, but
transient degrades. Likewise, if the ramp is made smaller,
transient response improves at the sacrifice of noise rejection
and stability.
C
ISS = 2710 pF (maximum) and RDS(SF) = 4.8 mΩ (maximum at
TJ = 120°C). The synchronous MOSFET CISS is less than 3000 pF,
satisfying this requirement.
The factor of 3 in the denominator of Equation 28 sets a ramp
size that gives an optimal balance for good stability, transient
response, and thermal balance.
Solving for the power dissipation per MOSFET at IO = 119 A and
IR = 11 A yields 958 mW for each synchronous MOSFET and
872 mW for each main MOSFET. A guideline to follow is to limit
the MOSFET power dissipation to 1 W. The values calculated in
Equation 25 and Equation 26 comply with this guideline.
Rev. 0 | Page 24 of 32
ADP3192A
For the ADP3192A, the maximum COMP voltage (VCOMP(MAX)
is 4.0 V, and the COMP pin bias voltage (VBIAS) is 1.1 V. In
this example, the maximum duty cycle is 0.61 and the peak
current is 62 A.
)
COMP PIN RAMP
A ramp signal on the COMP pin is due to the droop voltage
and output voltage ramps. This ramp amplitude adds to the
internal ramp to produce the following overall ramp signal
at the PWM input:
The limit of the peak per-phase current described earlier during
the secondary current limit is determined by
VR
VRT
=
(30)
VCOMP
−VBIAS
CLAMPED
)
(
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
2×
(
1−n × D
)
(34)
IPHLIM
≅
1−
AD × RDS
(
MAX )
n × fSW ×CX × RO
For the ADP3192A, the current balancing amplifier gain (AD) is
5 and the clamped COMP pin voltage is 2 V. Using an RDS(MAX)
of 2.8 mΩ (low-side on resistance at 150°C) results in a per-phase
peak current limit of 64 A. This current level can be reached only
with an absolute short at the output and only if the current-limit
latch-off function shuts down the regulator before overheating
can occur.
In this example, the overall ramp signal is 0.46 V. However,
if the ramp size is smaller than 0.5 V, increase the ramp size
to be at least 0.5 V by decreasing the ramp resistor for noise
immunity. Because there is only 0.46 V initially, a ramp resistor
value of 332 kΩ is chosen for this example, yielding an overall
ramp of 0.51 V.
CURRENT-LIMIT SETPOINT
FEEDBACK LOOP COMPENSATION DESIGN
To select the current-limit setpoint, first find the resistor value
for RLIM. The current-limit threshold for the ADP3192A is set
with a constant current source flowing out of the ILIMIT pin,
which sets up a voltage (VLIM) across RLIM with a gain of
82.6 mV/V (ALIM). Thus, increasing RLIM now increases the
current limit. RLIM can be found using
Optimized compensation of the ADP3192A allows the best
possible response of the regulator output to a load change.
The basis for determining the optimum compensation is to
make the regulator and output decoupling appear as an output
impedance that is entirely resistive over the widest possible
frequency range, including dc, and equal to the droop
resistance (RO). With the resistive output impedance, the output
voltage droops in proportion to the load current at any load
current slew rate. This ensures optimal positioning and
minimizes the output decoupling.
VCL
I
LIM ×RCSA
82.6 mV
RLIM
=
=
×RREF
(31)
ALIM × IILIMIT
Here, ILIM is the peak average current limit for the supply output.
The peak average current is the dc current limit plus the output
ripple current. In this example, choosing a dc current limit of
159 A and having a ripple current of 11 A gives an ILIM of 170 A.
This results in an RLIM = 205.8 kΩ, for which 205 kΩ is chosen
as the nearest 1% value.
Because of the multimode feedback structure of the ADP3192A,
the feedback compensation must be set to make the converter
output impedance work in parallel with the output decoupling
to make the load look entirely resistive. Compensation is
needed for several poles and zeros created by the output
inductor and the decoupling capacitors (output filter).
The per-phase initial duty cycle limit and peak current during a
load step are determined by
A type three compensator on the voltage feedback is adequate
for proper compensation of the output filter. Equation 35 to
Equation 39 are intended to yield an optimal starting point for
the design; some adjustments may be necessary to account for
PCB and component parasitic effects (see the Tuning the
ADP3192A section).
VCOMP
−VBIAS
MAX )
(
(32)
(33)
DMAX = D ×
VRT
DMAX
fSW
VIN −VVID
×
IPHMAX
≅
L
Rev. 0 | Page 25 of 32
ADP3192A
First, compute the time constants for all the poles and zeros in the system using Equation 35 to Equation 39.
RL ×VRT 2×L ×
1−n × D ×VRT
RE = n × RO + AD × RDS
+
+
VVID
n × CX × RO ×VVID
1.4 mꢀ × 0.51V 2 × 320 nH ×
(
1−0.432
)
× 0.51 V
(35)
RE = 4 ×1 mꢀ + 5 × 2.4 mꢀ +
+
= 22.9 mꢀ
1.3 V
4 × 5.6 mF ×1 mꢀ ×1.3 V
240 pH 1 mꢀ−0.5mꢀ
RO − R'
RX
LX
RO
TA = CX ×
(
RO −R'
)
+
×
=
= 5.6 mF ×
(
1 mꢀ−0.5 mꢀ
)
+
×
= 3.00 ꢁs
(36)
(37)
1 mꢀ
0.6mꢀ
TB =
RX + R' − RO
× CX
0.6 mꢀ +0.5 mꢀ −1 mꢀ
× 5.6 mF = 560 ns
⎛
⎞
⎟
⎟
⎠
⎛
⎞
⎟
⎟
⎠
AD × RDS
2 × fSW
5 × 2.4 mꢀ
2 × 330 kHz
⎜
⎜
VRT × L −
0.51 V × 320 nH−
⎜
⎜
⎝
⎝
TC =
TD =
=
= 5.17 ꢁs
(38)
(39)
VVID × RE
1.3 V × 22.9 mꢀ
2
CX ×C × R2
5.6 mF×180ꢁF×
+ CZ × RO 5.6 mF× 1mꢀ − 0.5mꢀ
1mꢀ
)
Z
O
=
= 338 ns
CX ×
(
RO − R'
)
(
)
+180ꢁF×1mꢀ
where:
R' is the PCB resistance from the bulk capacitors to the ceramics.
DS is the total low-side MOSFET on resistance per phase.
R
In this example, AD is 5, VRT equals 0.51 V, R' is approximately 0.5 mΩ (assuming a 4-layer, 1 ounce motherboard), and LX is 240 pH for
the 10 Al-Poly capacitors.
Rev. 0 | Page 26 of 32
ADP3192A
The compensation values can then be solved using
CIN SELECTION AND INPUT CURRENT
di/dt REDUCTION
n × RO ×TA 4×1 mꢀ × 3.00 ꢁs
CA
RA
=
=
=
=
= 524 pF
(40)
(41)
(42)
(43)
In continuous inductor current mode, the source current of the
high-side MOSFET is approximately a square wave with a duty
ratio equal to n × VOUT/VIN and an amplitude of one-nth the
maximum output current. To prevent large voltage transients,
a low ESR input capacitor, sized for the maximum rms current,
must be used. The maximum rms capacitor current is given by
RE × RB
5.17 ꢁs
CA 524 pF
22.9mꢀ ×1.00 kꢀ
TC
=
= 9.87 kꢀ
TB
560 ns
CB
CFB
=
= 560 pF
= 34.2 pF
RB 1.00 kꢀ
1
ICRMS = D × IO
×
−1
(44)
TD
338 ns
=
=
N × D
RA 9.87 kꢀ
1
These are the starting values prior to tuning the design that
account for layout and other parasitic effects (see the Tuning the
ICRMS = 0.108 ×119A ×
− 1 =14.7A
4 × 0.108
ADP3192A section). The final values selected after tuning are
The capacitor manufacturer’s ripple current ratings are often
based on only 2000 hours of life. As a result, it advisable to
further derate the capacitor or to choose a capacitor rated at a
higher temperature than required. Several capacitors can be
placed in parallel to meet size or height requirements in the
design. In this example, the input capacitor bank is formed by
two 2700 μF, 16 V aluminum electrolytic capacitors and eight
4.7 μF ceramic capacitors.
CA = 560 pF
RA = 10.0 kΩ
CB = 560 pF
CFB = 27 pF
Figure 13 and Figure 14 show the typical transient response
using these compensation values.
To reduce the input current di/dt to a level below the recom-
mended maximum of 0.1 A/μs, an additional small inductor
(L > 370 nH at 18 A) should be inserted between the converter
and the supply bus. This inductor also acts as a filter between
the converter and the primary power source.
1
THERMAL MONITOR DESIGN
A thermistor is used on the TTSENSE input of the ADP3192A
for monitoring the temperature of the VR. A constant current
of 123 μA is sourced out of this pin and runs through a
thermistor network such as the one shown in Figure 15.
ADP3192A
CH1 50mV
M 10µs
A CH1
–36mV
VRFAN
8
9
Figure 13. Typical Transient Response for Design Example Load Step
VRHOT
OPTIONAL
TEMPERATURE
ADJUST RESISTOR
10
TTSENSE
0.1µF
PLACE
THERMISTOR
NEAR CLOSEST
PHASE
1
R
TTSENSE
Figure 15. VR Thermal Monitor Circuit
A voltage is generated from this current through the thermistor
and sensed inside the IC. When the voltage reaches 1.105 V,
the VRFAN output gets set. When the voltage reaches 0.81 V,
the VRHOT gets set. This corresponds to RTTSENSE values of
8.98 kΩ for VRFAN and 6.58 kΩ for VRHOT.
CH1 50mV
M 10µs
A CH1
–36mV
Figure 14. Typical Transient Response for Design Example Load Release
These values correspond to a thermistor temperature of ~100°C
and ~110°C when using the same type of 100 kΩ NTC thermistor
used in the current sense amplifier.
Rev. 0 | Page 27 of 32
ADP3192A
An additional fixed resistor in parallel with the thermistor allows
tuning of the trip point temperatures to match the hottest tem-
perature in the VR when the thermistor itself is directly sensing
a proportionately lower temperature. Setting this resistor value
is best accomplished with a variable resistor during thermal
validation and then fixing this value for the final design.
where:
VIN(MAX) is the maximum voltage from the 12 V input supply (if
the 12 V input supply is 12 V 5%, VIN(MAX) = 12.6 V; if the 12 V
input supply is 12 V 10%, VIN(MAX) = 13.2 V).
V
CC(MIN) is the minimum VCC voltage of the ADP3192A. This is
specified as 4.75 V.
RSHUNT is the shunt resistor value.
Additionally, a 0.1 μF capacitor should be used for filtering noise.
The CECC standard specification for power rating in surface-
mount resistors is: 0603 = 0.1 W, 0805 = 0.125 W, 1206 = 0.25 W.
SHUNT RESISTOR DESIGN
The ADP3192A uses a shunt to generate 5 V from the 12 V
supply range. A trade-off can be made between the power
dissipated in the shunt resistor and the UVLO threshold.
Figure 16 shows the typical resistor value needed to realize
certain UVLO voltages. It also gives the maximum power
dissipated in the shunt resistor for these UVLO voltages.
TUNING THE ADP3192A
1. Build a circuit based on the compensation values
computed from the design spreadsheet.
2. Hook up the dc load to the circuit, turn it on, and verify its
operation. Also, check for jitter at no load and full load.
550
500
450
400
350
300
250
200
150
0.50
0.45
0.40
0.35
0.30
0.25
0.20
0.15
0.10
DC Load Line Setting
3. Measure the output voltage at no load (VNL). Verify that it
is within tolerance.
4. Measure the output voltage at full load cold (VFLCOLD).
Let the board sit for ~10 minutes at full load, and then
measure the output (VFLHOT). If there is a change of more
than a few millvolts, adjust RCS1 and RCS2 using Equation 46
and Equation 48.
P
R
SHUNT
SHUNT
VNL −VFLCOLD
VNL −VFLHOT
(46)
RCS2
= RCS2
×
OLD )
(
NEW
)
(
5. Repeat Step 4 until the cold and hot voltage measurements
remain the same.
7.0
7.5
8.0
8.5
9.0
(UVLO)
9.5
10.0
10.5
11.0
V
IN
6. Measure the output voltage from no load to full load using
5 A steps. Compute the load line slope for each change, and
then average to get the overall load line slope (ROMEAS).
7. If ROMEAS is off from RO by more than 0.05 mΩ, use
Equation 47 to adjust the RPH values.
Figure 16. Typical Shunt Resistor Value and Power Dissipation
for Different UVLO Voltage
The maximum power dissipated is calculated using Equation 45.
2
V
IN(MAX) −VCC(MIN)
PMAX
=
(45)
RSHUNT
ROMEAS
(47)
RPH
= RPH
×
OLD )
(
NEW
)
(
RO
8. Repeat Step 6 and Step 7 to check the load line. Repeat
adjustments if necessary.
9. When the dc load line adjustment is complete, do not
change RPH, RCS1, RCS2, or RTH for the remainder of the
procedure.
10. Measure the output ripple at no load and full load with
a scope and make sure it is within specifications.
1
RCS1
=
NEW )
(48)
(
RCS1
) + RTH
1
(
OLD
(
25°C
)
−
RCS1
) × RTH
+
(
RCS1
) − RCS2
)
×
(
RCS1
) − RTH
)
RTH
(
OLD
(
25°C
)
(
OLD
(
NEW
)
(
OLD
(
25°C
)
(
25°C
)
Rev. 0 | Page 28 of 32
ADP3192A
AC Load Line Setting
11. Remove the dc load from the circuit and hook up the
dynamic load.
12. Hook up the scope to the output voltage and set it to dc
coupling with the time scale at 100 μs/div.
V
DROOP
13. Set the dynamic load for a transient step of about 40 A at
1 kHz with 50% duty cycle.
14. Measure the output waveform (use dc offset on scope to see
the waveform). Try to use a vertical scale of 100 mV/div or
finer. This waveform should look similar to Figure 17.
V
TRAN1
V
TRAN2
Figure 18. Transient Setting Waveform
19. If both overshoots are larger than desired, try making
adjustments using the following suggestions:
V
ACDRP
•
•
Make the ramp resistor larger by 25% (RRAMP).
For VTRAN1, increase CB or increase the switching
frequency.
V
DCDRP
•
For VTRAN2, increase RA and decrease CA by 25%.
If these adjustments do not change the response, the design
is limited by the output decoupling. Check the output
response every time a change is made, and check the switch-
ing nodes to ensure that the response is still stable.
Figure 17. AC Load Line Waveform
15. Use the horizontal cursors to measure VACDRP and VDCDRP
,
20. For load release (see Figure 19), if VTRANREL is larger
than the allowed overshoot, there is not enough output
capacitance. Either more capacitance is needed, or the
inductor values need to be made smaller. When changing
inductors, start the design again using a spreadsheet and
this tuning procedure.
as shown in Figure 17. Do not measure the undershoot or
overshoot that happens immediately after this step.
16. If VACDRP and VDCDRP are different by more than a few
millivolts, use Equation 49 to adjust CCS. Users may need to
parallel different values to get the right one, because limited
standard capacitor values are available. It is recommended to
have locations for two capacitors in this layout.
VACDRP
VDCDRP
(49)
CCS
= CCS
×
OLD )
(
NEW
)
(
V
TRANREL
V
DROOP
17. Repeat Step 11 to Step 13 and repeat the adjustments, if
necessary. Once complete, do not change CCS for the
remainder of the procedure. Set the dynamic load step to
maximum step size. Do not use a step size larger than
needed. Verify that the output waveform is square, which
means that VACDRP and VDCDRP are equal.
Initial Transient Setting
18. With the dynamic load still set at the maximum step size,
expand the scope time scale to either 2 μs/div or 5 μs/div.
The waveform can have two overshoots and one minor
undershoot (see Figure 18). Here, VDROOP is the final
desired value.
Figure 19. Transient Setting Waveform
Because the ADP3192A turns off all of the phases (switches
inductors to ground), no ripple voltage is present during load
release. Therefore, the user does not have to add headroom for
ripple. This allows load release VTRANREL to be larger than VTRAN1
by the amount of ripple and still meet specifications.
If VTRAN1 and VTRANREL are less than the desired final droop, the
capacitors may be removed. When removing capacitors, check the
output ripple voltage to make sure it is still within specifications.
Rev. 0 | Page 29 of 32
ADP3192A
Power Circuitry Recommendations
LAYOUT AND COMPONENT PLACEMENT
The switching power path should be routed on the PCB to
encompass the shortest possible length to minimize radiated
switching noise energy (EMI) and conduction losses in the
board. Failure to take proper precautions often results in EMI
problems for the entire PC system and noise-related operational
problems in the power converter control circuitry. The
switching power path is the loop formed by the current path
through the input capacitors and the power MOSFETs, including
all interconnecting PCB traces and planes. Using short and wide
interconnection traces is especially critical in this path for two
reasons: it minimizes the inductance in the switching loop that
can cause high energy ringing, and it accommodates the high
current demand with minimal voltage loss.
The guidelines outlined in this section are recommended for
optimal performance of a switching regulator in a PC system.
General Recommendations
For effective results, a PCB with at least four layers is recom-
mended. This provides the needed versatility for control circuitry
interconnections with optimal placement, power planes for
ground, input and output power, and wide interconnection
traces in the remainder of the power delivery current paths.
Keep in mind that each square unit of 1 ounce copper trace
has a resistance of ~0.53 mΩ at room temperature.
Whenever high currents must be routed between PCB layers,
use vias liberally to create several parallel current paths, so the
resistance and inductance introduced by these current paths is
minimized, and the via current rating is not exceeded.
When a power dissipating component (for example, a power
MOSFET) is soldered to a PCB, it is recommended to liberally
use the vias, both directly on the mounting pad and immediately
surrounding it. Two important reasons for this are improved
current rating through the vias and improved thermal perform-
ance from vias extended to the opposite side of the PCB, where a
plane can more readily transfer the heat to the air. Make a
mirror image of any pad being used to heatsink the MOSFETs
on the opposite side of the PCB to achieve the best thermal
dissipation in the air around the board. To further improve
thermal performance, use the largest possible pad area.
If critical signal lines (including the output voltage sense lines of
the ADP3192A) must cross through power circuitry, it is best to
interpose a signal ground plane between those signal lines and
the traces of the power circuitry. This serves as a shield to
minimize noise injection into the signals at the expense of
making signal ground a bit noisier.
An analog ground plane should be used around and under the
ADP3192A as a reference for the components associated with
the controller. This plane should be tied to the nearest output
decoupling capacitor ground and should not be tied to any other
power circuitry to prevent power currents from flowing into it.
The output power path should also be routed to encompass a
short distance. The output power path is formed by the current
path through the inductor, the output capacitors, and the load.
The components around the ADP3192A should be located close
to the controller with short traces. The most important traces to
keep short and away from other traces are the FB pin and CSSUM
pin. The output capacitors should be connected as close as
possible to the load (or connector), for example, a microproc-
essor core, that receives the power. If the load is distributed, the
capacitors should also be distributed and generally be in
For best EMI containment, a solid power ground plane should
be used as one of the inner layers extending fully under all the
power components.
Signal Circuitry Recommendations
The output voltage is sensed and regulated between the FB pin
and the FBRTN pin, which connect to the signal ground at the
load. To avoid differential mode noise pickup in the sensed
signal, the loop area should be small. Thus, the FB trace and
FBRTN trace should be routed adjacent to each other on top
of the power ground plane back to the controller.
proportion to where the load tends to be more dynamic.
Avoid crossing any signal lines over the switching power path loop
(described in the Power Circuitry Recommendations section).
The feedback traces from the switch nodes should be connected
as close as possible to the inductor. The CSREF signal should be
connected to the output voltage at the nearest inductor to the
controller.
Rev. 0 | Page 30 of 32
ADP3192A
OUTLINE DIMENSIONS
6.00
BSC SQ
0.60 MAX
0.60 MAX
PIN 1
INDICATOR
31
40
1
30
PIN 1
INDICATOR
0.50
BSC
TOP
VIEW
4.25
4.10 SQ
3.95
5.75
BCS SQ
EXPOSED
PAD
(BOT TOM VIEW)
0.50
0.40
0.30
21
10
11
20
0.25 MIN
4.50
REF
12° MAX
0.80 MAX
0.65 TYP
0.05 MAX
0.02 NOM
1.00
0.85
0.80
0.30
0.23
0.18
COPLANARITY
0.08
0.20 REF
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-220-VJJD-2
Figure 20. 40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
6 mm × 6 mm Body, Very Thin Quad
(CP-40-1)
Dimensions shown in millimeters
ORDERING GUIDE
Model
Temperature Range Package Description
Package Option Ordering Quantity
CP-40-1 2,500
ADP3192AJCPZ-RL1 0°C to 85°C
40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
1 Z = RoHS Compliant Part.
Rev. 0 | Page 31 of 32
ADP3192A
NOTES
©2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06786-0-5/07(0)
Rev. 0 | Page 32 of 32
相关型号:
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ADP3193JCPZ-RL
IC SWITCHING CONTROLLER, 4000 kHz SWITCHING FREQ-MAX, QCC32, 5 X 5 MM, LEAD FREE, MO-220VHHD-2, LFCSP-32, Switching Regulator or Controller
ADI
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