ADP3163JRU-REEL7 [ADI]
IC SWITCHING CONTROLLER, 3000 kHz SWITCHING FREQ-MAX, PDSO20, TSSOP-20, Switching Regulator or Controller;
| 型号： | ADP3163JRU-REEL7 |
| 厂家： | 
ADI |
| 描述： | IC SWITCHING CONTROLLER, 3000 kHz SWITCHING FREQ-MAX, PDSO20, TSSOP-20, Switching Regulator or Controller 开关 光电二极管 |
| 文件： | 总16页 (文件大小：131K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
5-Bit Programmable 2-/3-Phase
Synchronous Buck Controller
a
ADP3163
FEATURES
FUNCTIO NAL BLO CK D IAGRAM
ADOPT™ Optimal Positioning Technology for Superior
Load Transient Response and Fewest Output
Capacitors
Complies with VRM 9.0 and Intel VR Down Guideline
with Lowest System Cost
PC
VCC
ADP3163
SET
RESET
PWM1
PWM2
PWM3
UVLO
& BIAS
2-/3-PHASE
DRIVER
LOGIC
Digitally Selectable 2- or 3-Phase Operation
at up to 500 kHz per Phase
CROWBAR
Quad Logic-level PWM Outputs for Interface to
External High-Power Drivers
3.0V
REFERENCE
REF
PGND
Active Current Balancing between All Output Phases
Accurate Multiple VRM Module Current Sharing
5-Bit Digitally Programmable 1.1 V to 1.85 V Output
Total Output Accuracy ꢀ0.8% Over Temperature
Current-Mode Operation
Short Circuit Protection
Enhanced Power Good Output Detects Open Outputs in
Multi-VRM Power Systems
GND
DAC+20%
OSCILLATOR
CT
POWER
GOOD
PWRGD
SHARE
CMP
Overvoltage Protection Crowbar Protects Microprocessors
with No Additional External Components
DAC+20%
CS–
CS+
FB
CMP
APPLICATIONS
Desktop PC Power Supplies for:
Intel Pentium® 4 Processors
AMD Athlon Processors
VRM Modules
g
COMP
m
SOFT
START
VID
DAC
VID4 VID3 VID2 VID1 VID0
GENERAL D ESCRIP TIO N
The ADP3163 also uses a unique supplemental regulation tech-
nique called active voltage positioning (ADOPT) to enhance
load transient performance. Active voltage positioning results in
a dc/dc converter that meets the stringent output voltage specifi-
cations for high performance processors, with the minimum
number of output capacitors and smallest footprint. Unlike
voltage-mode and standard current-mode architectures, active
voltage positioning adjusts the output voltage as a function of
the load current so that it is always optimally positioned for a
system transient. The ADP3163 also provides accurate and
reliable short circuit protection, adjustable current limiting, and
an enhanced Power Good output that can detect open outputs
in any phase for single or multi-VRM systems.
The ADP3163 is a highly efficient multiphase synchronous buck
switching regulator controller optimized for converting a 5 V or
12 V main supply into the core supply voltage required by high
performance Intel processors. The ADP3163 uses an internal
5-bit DAC to read a voltage identification (VID) code directly
from the processor, which is used to set the output voltage between
1.1 V and 1.85 V. The ADP3163 uses a current mode PWM
architecture to drive the logic-level outputs at a programmable
switching frequency that can be optimized for VRM size and
efficiency. The phase relationship of the output signals can be
programmed to provide 2- or 3-phase operation, allowing for
the construction of up to three complementary buck switching
stages. These stages share the dc output current to reduce
overall output voltage ripple. An active current balancing func-
tion ensures that all phases carry equal portions of the total load
current, even under large transient loads, to minimize the size of
the inductors.
The ADP3163 is specified over the commercial temperature
range of 0°C to 70°C and is available in a 20-lead TSSOP package.
ADOPT is a trademark of Analog Devices, Inc.
Pentium is a registered trademark of Intel Corporation.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, norforanyinfringementsofpatentsorotherrightsofthirdpartiesthat
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
Fax: 781/326-8703
www.analog.com
© Analog Devices, Inc., 2001
1
ADP3163–SPECIFICATIONS
(VCC = 12 V, IREF = 150 ꢁA, T = 0ꢂC to 70ꢂC, unless otherwise noted.)
A
P aram eter
Sym bol
Conditions
Min
Typ
Max
Unit
FEEDBACK INPUT
Accuracy
VFB
1.1 V Output
1.6 V Output
1.85 V Output
Line Regulation
Input Bias Current
Crowbar Trip Point
Crowbar Reset Point
Crowbar Response Time
1.091
1.587
1.835
1.1
1.6
1.85
0.01
5
120
50
1.109
1.613
1.865
V
V
V
%
nA
%
%
ns
∆VFB
IFB
VCROWBAR
VCC = 10 V to 14 V
50
125
60
% of Nominal Output
% of Nominal Output
Overvoltage to PWM Going Low
115
40
tCROWBAR
400
REFERENCE
Output Voltage
Output Current
VREF
IREF
2.952
300
3.00
3.048
V
µA
VID INPUTS
Input Low Voltage
Input High Voltage
Input Current
Pull-Up Resistance
Internal Pull-Up Voltage
VIL(VID)
VIH(VID)
IVID
0.8
90
V
V
µA
kΩ
V
2.0
VID(X) = 0 V
70
43
3.0
RVID
33
2.7
3.3
OSCILLATOR
Maximum Frequency2
Frequency Variation
fCT(MAX)
fCT
3000
475
850
1100
260
40
kHz
kHz
kHz
kHz
µA
TA = 25°C, CT = 150 pF
TA = 25°C, CT = 68 pF
TA = 25°C, CT = 47 pF
TA = 25°C, VFB in Regulation
TA = 25°C, VFB = 0 V
575
1000
1300
300
65
675
1250
1500
340
80
CT Charge Current
ICT
µA
ERROR AMPLIFIER
Output Resistance
Transconductance
RO(ERR)
gm(ERR)
IO(ERR)
1
MΩ
mmho
µA
V
mV
2.0
2.2
575
3.0
800
500
2.45
875
Output Current
FB Forced to 0 V
Maximum Output Voltage
Output Disable Threshold
–3 dB Bandwidth
VCOMP(MAX) FB Forced to VOUT – 3%
VCOMP(OFF)
BWERR
COMP = Open
kHz
CURRENT SENSE
Threshold Voltage
VCS(TH)
CS+ = VCC,
FB Forced to VOUT – 3%
FB ≤ 750 mV
0.8 V ≤ SHARE ≤ 1 V
CS+ = CS– = VCC
CS+ – (CS–) ≥ 173 mV
143
80
158
173
mV
92
0
1
108
5
5
mV
mV
µA
ns
Input Bias Current
Response Time
ICS+, ICS–
tCS
50
to PWM Going Low
CURRENT SHARING
Output Source Current
Output Sink Current
2
mA
µA
V
300
3.0
400
0.8
Maximum Output Voltage
VSHARE(MAX) FB Forced to VOUT – 3%
PHASE CONTROL
Input Low Voltage
Input High Voltage
VIL(PC)
VIH(PC)
V
V
2.0
POWER GOOD COMPARATOR
Undervoltage Threshold
Overvoltage Threshold
Output Voltage Low
VPWRGD(UV) Percent of Nominal Output
VPWRGD(OV) Percent of Nominal Output
VOL(PWRGD) IPWRGD(SINK) = 1 mA
75
115
80
85
125
525
%
%
mV
ns
120
375
250
Response Time
–2–
REV. 0
ADP3163
P aram eter
Sym bol
Conditions
Min
Typ
Max
Unit
PWM OUTPUTS
Output Voltage Low
Output Voltage High
Duty Cycle Limit Per Phase2
VOL(PWM)
VOH(PWM)
DC
IPWM(SINK) = 400 µA
IPWM(SOURCE) =400 µA
PC = GND
100
5.0
500
mV
V
%
4.0
50
33
PC = REF
%
SUPPLY
DC Supply Current
Normal Mode
No CPU Mode
UVLO Mode
UVLO Threshold Voltage
UVLO Hysteresis
ICC
3.75
3.5
350
6.4
5.5
5.5
500
6.9
1.0
mA
mA
µA
V
ICC(NO CPU)
ICC(UVLO)
VUVLO
VID4 – VID0 = Open
VCC ≤ VUVLO, VCC Rising
5.9
0.5
0.8
V
NOTES
1All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC).
2Guaranteed by design, not tested in production.
Specifications subject to change without notice.
P IN CO NFIGURATIO N
RU-20
ABSO LUTE MAXIMUM RATINGS*
VCC
VID4
VID3
1
2
3
4
20
19
18
17
16
15
VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +15 V
CS+, CS– . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VCC +0.3 V
All Other Inputs and Outputs . . . . . . . . . . . . –0.3 V to +10 V
Operating Ambient Temperature Range . . . . . . . 0°C to 70°C
Operating Junction Temperature . . . . . . . . . . . . . . . . . . 125°C
Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C
θJA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 143°C/W
Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300°C
Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . . 215°C
Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220°C
REF
VID2
PWM1
PWM2
PWM3
PC
VID1
ADP3163
TOPVIEW
(NOTTO SCALE)
VID0
5
6
7
8
9
SHARE
COMP
GND
14 PGND
13 CS–
FB
CS+
12
*This is a stress rating only; operation beyond these limits can cause the device to
be permanently damaged. Unless otherwise specified, all voltages are referenced
to PGND.
CT 10
PWRGD
11
O RD ERING GUID E
Model
Tem perature Range
P ackage D escription
Thin Shrink Small Outline
P ackage O ption
ADP3163JRU
0°C to 70°C
RU-20 (TSSOP-20)
CAUTIO N
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the ADP3163 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
–3–
REV. 0
ADP3163
P IN FUNCTIO N D ESCRIP TIO NS
P in
Nam e
Function
1–5
VID4 –
VID0
Voltage Identification DAC Inputs. These pins are pulled up to an internal 3 V reference, providing a
Logic 1 if left open. The DAC output programs the FB regulation voltage from 1.1 V to 1.85 V. Leaving all five
DAC inputs open results in the ADP3163 going into a “No CPU” mode, shutting off its PWM outputs.
6
SHARE
Current Sharing Output. This pin is connected to the SHARE pins of other ADP3163s in multiple VRM sys-
tems to ensure proper current sharing between the converters. The voltage at this output programs the output
current control level between CS+ and CS–.
7
8
COMP
GND
Error Amplifier Output and Compensation Point.
Ground. FB, REF and the VID DAC of the ADP3163 are referenced to this ground. This is a low current ground
that can also be used as a return for the FB pin in remote voltage sensing applications.
9
10
11
FB
CT
PWRGD
Feedback Input. Error amplifier input for remote sensing of the output voltage.
External capacitor CT connection to ground sets the frequency of the device.
Open drain output that signals when the output voltage is outside of the proper operating range or when a phase
is not supplying current even if the output voltage is in specification.
12
CS+
Current Sense Positive Node. Positive input for the current comparator. The output current is sensed as a volt-
age at this pin with respect to CS–.
13
14
15
CS–
PGND
PC
Current Sense Negative Node. Negative input for the current comparator.
Power Ground. All internal biasing and logic output signals of the ADP3163 are referenced to this ground.
Phase Control Input. This logic-level input determines the number of active phases and the duty cycle limit of
each phase.
16
17
18
19
20
PWM3
PWM2
PWM1
REF
Logic-Level Output for the Phase 3 Driver.
Logic-Level Output for the Phase 2 Driver.
Logic-Level Output for the Phase 1 Driver.
3.0 V Reference Output.
VCC
Supply Voltage for the ADP3163.
ADP3163
Table I. P WM O utputs vs. P hase Control Code
VID4
VCC
REF
1
2
3
4
20
19
18
17
16
15
14
13
12
11
12V
Maxim um
D uty Cycle
1ꢁF
100nF
VID3
VID2
VID1
VID0
SHARE
COMP
GND
FB
P C
P WM3
P WM2
P WM1
20kꢃ
PWM1
PWM2
PWM3
PC
5-BIT CODE
REF
GND
ON
OFF
ON
ON
ON
ON
33%
50%
5
6
PGND
CS–
7
8
100ꢃ
CS+
9
100nF
CT
PWRGD
10
V
FB
AD820
1.2V
Figure1. Closed-LoopOutputVoltageAccuracyTestCircuit
–4–
REV. 0
Typical Performance Characteristics–ADP3163
4.5
4.4
4.3
10
1.0
0.1
4.2
4.1
4.0
0
500
1000
1500
2000
2500
3000
0
50
100
150
200
250
300
OSCILLATOR FREQUENCY – kHz
CT CAPACITANCE – pF
TPC 1. Oscillator Frequency vs. Timing Capacitor (CT)
TPC 2. Supply Current vs. Oscillator Frequency
25
20
T
= 25ꢂC
A
V
= 1.6V
OUT
15
10
5
0
–0.5
0
0.5
OUTPUT ACCURACY – % of Nominal
TPC 3. Output Accuracy Distribution
–5–
REV. 0
ADP3163
phase is inherently limited to 50% for 2-phase operation and
33% for 3-phase operation. While one phase is on, all other
phases remain off. In no case can more than one output be high
at any time.
Table II. O utput Voltage vs. VID Code
VID 4
VID 3
VID 2
VID 1
VID 0
VO UT(NO M)
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
No CPU
1.100 V
1.125 V
1.150 V
1.175 V
1.200 V
1.225 V
1.250 V
1.275 V
1.300 V
1.325 V
1.350 V
1.375 V
1.400 V
1.425 V
1.450 V
1.475 V
1.500 V
1.525 V
1.550 V
1.575 V
1.600 V
1.625 V
1.650 V
1.675 V
1.700 V
1.725 V
1.750 V
1.775 V
1.800 V
1.825 V
1.850 V
O utput Voltage Sensing
The output voltage is sensed at the FB pin allowing for remote
sensing. To maintain the accuracy of the remote sensing, the
GND pin should also be connected close to the load. A voltage
error amplifier (gm) amplifies the difference between the output
voltage and a programmable reference voltage. The reference
voltage is programmed between 1.1 V and 1.85 V by an internal
5-bit DAC, which reads the code at the voltage identification
(VID) pins. (Refer to Table II for the output voltage versus VID
pin code information.)
Active Voltage P ositioning
The ADP3163 uses Analog Devices Optimal Positioning Tech-
nology (ADOPT), a unique supplemental regulation technique
that uses active voltage positioning and provides optimal com-
pensation for load transients. When implemented, ADOPT
adjusts the output voltage as a function of the load current, so
that it is always optimally positioned for a load transient.
Standard (passive) voltage positioning has poor dynamic perfor-
mance, rendering it ineffective under the stringent repetitive
transient conditions required by high performance processors.
ADOPT, however, provides optimal bandwidth for transient
response that yields optimal load transient response with the
minimum number of output capacitors.
Refer ence O utput
A 3.0 V reference is available on the ADP3163. This reference
is normally used to set the voltage positioning accurately using a
resistor divider to the COMP pin. In addition, the reference can
be used for other functions such as generating a regulated volt-
age with an external amplifier. The reference is bypassed with
a 1 nF capacitor to ground. It is not intended to supply large
capacitive loads, and it should not be used to provide more than
300 µA of output current.
TH EO RY O F O P ERATIO N
Cycle-by-Cycle O per ation
The ADP3163 combines a current-mode, fixed frequency PWM
controller with multiphase logic outputs for use in a 2- or 3-phase
synchronous buck power converter. Multiphase operation is
important for switching the high currents required by high
performance microprocessors. Handling the high current in a
single-phase converter would place unreasonable requirements on
the power components such as inductor wire size and MOSFET
ON-resistance and thermal dissipation. The ADP3163’s high-side
current sensing topology ensures that the load currents are
balanced in each phase, such that no single phase has to carry
more than it’s share of the power. An additional benefit of high
side current sensing over output current sensing is that the
average current through the sense resistor is reduced by the duty
cycle of the converter allowing the use of a lower power, lower
cost resistor. The outputs of the ADP3163 are logic drivers
only and are not intended to directly drive external power
MOSFETs. Instead, the ADP3163 should be paired with driv-
ers such as the ADP3413 or ADP3414.
During normal operation (when the output voltage is regulated),
the voltage-error amplifier and the current comparator are the
main control elements. The voltage at the CT pin of the oscilla-
tor ramps between 0 V and 3 V. When that voltage reaches 3 V,
the oscillator sets the driver logic, which sets PWM1 high. Dur-
ing the ON time of Phase 1, the driver IC turns on the Phase 1
high-side MOSFET. The CS+ and CS– pins monitor the current
through the sense resistor that feeds all the high side MOSFETs.
When the voltage between the two pins exceeds the threshold
level, the driver logic is reset and the PWM1 output goes low. This
signals the driver IC to turn off the Phase 1 high side MOSFET
and turn on the Phase 1 low side MOSFET. On the next cycle
of the oscillator, the driver logic toggles and sets PWM2 high.
On each following cycle of the oscillator, the driver logic cycles
between each of the active PWM outputs based on the logic
state of the PC pin. In each case, the current comparator resets
the PWM output low when its threshold is reached. As the load
current increases, the output voltage starts to decrease. This
causes an increase in the output of the gm amplifier, which in
turn leads to an increase in the current comparator threshold,
thus programming more load current to be delivered so that
voltage regulation is maintained.
The frequency of the ADP3163 is set by an external capacitor
connected to the CT pin. The phase relationship and number of
active output phases is determined by the state of the phase
control (PC) pin as shown in Table I. The error amplifier and
current sense comparator control the duty cycle of the PWM
outputs to maintain regulation. The maximum duty cycle per
–6–
REV. 0
ADP3163
Active Cur r ent Shar ing
voltage specification even when one VRM is not functioning.
The ADP3163 addresses this problem by monitoring both the
output voltage and the switch current to determine the state of
the PWRGD output.
The ADP3163 ensures current balance in all the active phases
by sensing the current through a single sense resistor. During
one phase's ON time, the current through the respective high
side MOSFET and inductor is measured through the sense
resistor. When the comparator threshold is reached, the high side
MOSFET turns off. On the next cycle the ADP3163 switches to
the next phase. The current is measured with the same sense
resistor and the same internal comparator, ensuring accurate
matching. This scheme is immune to imbalances in the MOSFET’s
RDS(ON) and inductor parasitic resistance.
The output voltage portion of the Power Good monitor domi-
nates; as long as the output voltage is outside the specified
window, PWRGD will remain low. If the output voltage is
within specification, a second circuit checks to make sure that
current is being delivered to the output by each phase. If no
current is detected in a phase for three consecutive cycles, it is
assumed that an open circuit exists somewhere in the power
path, and PWRGD will be pulled low.
If for some reason one of the phases fails, the other phases will
still be limited to their maximum output current (one over the
total number phases times the total short circuit current limit).
If this is not sufficient to supply the load, the output voltage will
droop and cause the PWRGD output to signal that the output
voltage has fallen out of its specified range. If one of the phases
has an open circuit failure, the ADP3163 will detect the open
phase and signal the problem via the PWRGD pin (see Power
Good Monitoring section).
O utput Cr owbar
The ADP3163 includes a crowbar comparator that senses when
the output voltage rises higher than the specified trip threshold,
VCROWBAR. This comparator overrides the control loop and sets
both PWM outputs low. The driver ICs turn off the high side
MOSFETs and turn on the low side MOSFETs, thus pulling
the output down as the reversed current builds up in the induc-
tors. If the output overvoltage is due to a short of the high side
MOSFET, this action will current limit the input supply or blow
its fuse, protecting the microprocessor from destruction. The
crowbar comparator releases when the output drops below the
specified reset threshold, and the controller returns to normal
operation if the cause of the over voltage failure does not persist.
Cur r ent Shar ing in Multi-VRM Applications
The ADP3163 includes a SHARE pin to allow multiple VRMs
to accurately share load current. In multiple VRM applications,
the SHARE pins should be connected together. This pin is a
low impedance buffered output of the COMP pin voltage. The
output of the buffer is internally connected to set the threshold
of the current sense comparator. The buffer has a 400 µA sink
current, and a 2 mA sourcing capability. The strong pull-up
allows one VRM to control the current threshold set point for
all ADP3163s connected together. The ADP3163’s high accuracy
current set threshold ensures good current balance between
VRMs. Also, the low impedance of the buffer minimizes noise
pick up on this trace which is routed to multiple VRMs. This
circuit operates in addition to the active current sharing between
phases of each VRM described above.
O utput D isable
The ADP3163 includes an output disable function that turns off
the control loop to bring the output voltage to 0 V. Because an
extra pin is not available, the disable feature is accomplished by
pulling the COMP pin to ground. When the COMP pin drops
below 0.8 V, the oscillator stops and all PWM signals are driven
low. When in this state, the reference voltage is still available.
The COMP pin should be pulled down with an open drain
structure capable of sinking at least 2 mA.
Shor t Cir cuit P r otection
AP P LICATIO N INFO RMATIO N
The ADP3163 has multiple levels of short circuit protection to
ensure fail-safe operation. The sense resistor and the maximum
current sense threshold voltage given in the specifications set the
peak current limit.
The design parameters for a typical Intel Pentium4 CPU appli-
cation are as follows:
Input voltage (VIN) = 12 V
VID setting voltage (VVID) = 1.5 V
When the load current exceeds the current limit, the excess current
discharges the output capacitor. When the output voltage is below
the foldback threshold, VFB(LOW), the maximum deliverable output
current is cut by reducing the current sense threshold from
the current limit threshold, VCS(CL), to the foldback threshold,
Nominal output voltage at no load (VONL) = 1.475 V
Nominal output voltage at 65 A load (VOFL) = 1.377 V
Static output voltage drop based on a 1.5 mΩ load line
(ROUT) from no load to full load (V∆) = VONL – VOFL
1.475 V – 1.377 V = 98 mV
=
V
CS(FOLD). Along with the resulting current foldback, the oscilla-
Maximum Output Current (IO) = 65 A
Number of Phases (n) = 3
tor frequency is reduced by a factor of five when the output is
0 V. This further reduces the average current in short circuit.
CT Selection—Choosing the Clock Fr equency
P ower Good Monitor ing
The ADP3163 uses a fixed-frequency control architecture. The
frequency is set by an external timing capacitor, CT. The clock
frequency and the state of the PC pin determine the switching
frequency, which relates directly to switching losses and the
sizes of the inductors and input and output capacitors. With PC
tied to REF, a clock frequency of 600 kHz sets the switching
frequency of each phase, fSW, to 200 kHz, which represents a
practical trade-off between the switching losses and the sizes of
the output filter components. To achieve a 600 kHz oscillator
frequency, the required timing capacitor value is 150 pF. For
good frequency stability and initial accuracy, it is recommended
to use a capacitor with low temperature coefficient and tight
The Power Good comparator monitors the output voltage of the
supply via the FB pin. The PWRGD pin is an open drain output
whose high level (when connected to a pull-up resistor) indicates
that the output voltage is within the specified range of the nomi-
nal output voltage requested by the VID DAC. PWRGD will go
low if the output is outside this range.
Short circuits in a VRM power path are relatively easy to detect
in applications where multiple VRMs are connected to a common
power plane. VRM power train open failures are not as easily
spotted, since the other VRMs may be able to supply enough
total current to keep the output voltage within the Power Good
REV. 0
–7–
ADP3163
tolerance, e.g., an MLC capacitor with NPO dielectric and with
5% or less tolerance.
the circuitry surrounding the inductor. Closed-loop types, such
as pot cores, PQ, U, and E cores, or toroids, cost more, but
have much better EMI/RFI performance. A good compromise
between price and performance are cores with a toroidal shape.
Inductance Selection
The choice of inductance determines the ripple current in the
inductor. Less inductance leads to more ripple current, which
increases the output ripple voltage and the conduction losses in
the MOSFETs, but allows using smaller-size inductors and, for
a specified peak-to-peak transient deviation, output capacitors
with less total capacitance. Conversely, a higher inductance
means lower ripple current and reduced conduction losses, but
requires larger-size inductors and more output capacitance for
the same peak-to-peak transient deviation. In a three-phase
converter, a practical value for the peak-to-peak inductor ripple
current is under 50% of the dc current in the same inductor. A
choice of 50% for this particular design example yields a total
peak-to-peak output ripple current of 12% of the total dc output
current. The following equation shows the relationship between
the inductance, oscillator frequency, peak-to-peak ripple current
in an inductor and input and output voltages.
There are many useful references for quickly designing a power
inductor. Table III gives some examples.
Table III. Magnetics D esign Refer ences
Magnetic Designer Software
Intusoft (http://www.intusoft.com)
Designing Magnetic Components for High-Frequency DC-DC
Converters
McLyman, Kg Magnetics
ISBN 1-883107-00-08
Selecting a Standar d Inductor
The companies listed in Table IV can provide design consulta-
tion and deliver power inductors optimized for high power
applications upon request.
Table IV. P ower Inductor Manufactur er s
(VIN –VOUT ) ×VOUT
VIN × fSW × IL(RIPPLE )
L =
Coilcraft
(1)
(847)639-6400
http://www.coilcraft.com
For an 11 A peak-to-peak ripple current, which corresponds to
50% of the 22 A full-load dc current in an inductor, Equation 1
yields an inductance of:
Coiltronics
(561)752-5000
http://www.coiltronics.com
(12V –1.5V )×1.5V
Sumida Electric Company
(408)982-9660
http://www.sumida.com
L =
= 596 nH
600 kHz
12V ×
×11 A
3
A 600 nH inductor can be used, which gives a calculated ripple
current of 10.9 A at no load. The inductor should not saturate
at the peak current of 27 A, and should be able to handle the
sum of the power dissipation caused by the average current of
22 A in the winding and the core loss.
RSENSE
The value of RSENSE is based on the maximum required output
current. The current comparator of the ADP3163 has a mini-
mum current limit threshold of 143 mV. Note that the 143 mV
value cannot be used for the maximum specified nominal cur-
rent, as headroom is needed for ripple current and tolerances.
The output ripple current is smaller than the inductor ripple
current due to the three phases partially canceling. This can be
calculated as follows:
The current comparator threshold sets the peak of the inductor
current yielding a maximum output current, IO, which equals
the peak inductor current value less half of the peak-to-peak induc-
tor ripple current. From this, the maximum value of RSENSE is
calculated as:
n ×VOUT ×(VIN – n ×VOUT
VIN × L × fOSC
)
IO∆
=
(2)
3 ×1.5V ×(12V – 3 ×1.5V )
12V × 600 nH × 600 kHz
IO∆
=
= 7.81 A
VCSCL(MIN )
143 mV
RSENSE
≤
=
= 5.3 mΩ
IL(RIPPLE )
65 A 10.9 A
IO
n
(3)
D esigning an Inductor
+
+
3
2
2
Once the inductance is known, the next step is either to design
an inductor or find a standard inductor that comes as close as
possible to meeting the overall design goals. The first decision in
designing the inductor is to choose the core material. There are
several possibilities for providing low core loss at high frequen-
cies. Two examples are the powder cores (e.g., Kool-Mµ® from
Magnetics, Inc.) and the gapped soft ferrite cores (e.g., 3F3 or
3F4 from Philips). Low frequency powdered iron cores should
be avoided due to their high core loss, especially when the
inductor value is relatively low and the ripple current is high.
In this case, 5 mΩ was chosen as the closest standard value.
Once RSENSE has been chosen, the output current at the point
where current limit is reached, IOUT(CL), can be calculated using
the maximum current sense threshold of 173 mV:
VCSCL(MAX ) n × IL(RIPPLE )
IOUT (CL ) = n ×
−
RSENSE
2
(4)
173 mV 3 ×10.9 A
5 mΩ
IOUT (CL ) = 3 ×
−
= 87.5 A
2
Two main core types can be used in this application. Open
magnetic loop types, such as beads, beads on leads, and rods
and slugs, provide lower cost but do not have a focused mag-
netic field in the core. The radiated EMI from the distributed
magnetic field may create problems with noise interference in
–8–
REV. 0
ADP3163
Figure 2. 65A Intel Pentium 4CPU Supply Circuit, VR Down Guideline Design
REV. 0
–9–
ADP3163
At output voltages below 750 mV, the current sense threshold is
reduced to 108 mV, and the ripple current is negligible. There-
fore, at dead short the output current is reduced to:
I
L(RIPPLE ) × RSENSE × nI
VIN −VOUT
VGNL =VGNL0
+
−
×
2
L
n × tD × RSENSE × nI
VCS(SC )
RSENSE
108 mV
5 mΩ
IOUT (SC ) = n ×
= 3 ×
= 65 A
(5)
10.9 A × 5 mΩ ×12.5 12V −1.5V
(10)
To safely carry the current under maximum load conditions, the
sense resistor must have a power rating of at least:
VGNL = 1V +
−
×
2
600 nH
2 × 60 ns × 5 mΩ ×12.5 = 1.144V
2
(6)
P
= ISENSE (RMS ) × RSENSE
RSENSE
The divider resistors (RA for the upper and RB for the lower)
can now be calculated, assuming that the internal resistance of
the gm amplifier (ROGM) is 1 MΩ:
where:
ISENSE (RMS )
2
IO
n
VOUT
2
=
×
(7)
η ×VIN
VREF
RB
=
In this formula, n is the number of phases, and η is the con-
verter efficiency, in this case assumed to be 85%. Combining
Equations 6 and 7 yields:
VREF −VGNL
− gm ×(VONL −VVID
)
RT
(11)
3V
− 2.2 mmho ×(1.475V −1.5V )
65 A2
3
1.5V
0.85×12V
RB
=
= 8.59 kΩ
3V −1.144V
6.31kΩ
PR
=
×
× 5 mΩ = 1.0W
SENSE
Choosing the nearest 1% resistor value gives RB = 8.66 kΩ.
O utput Resistance
Finally, RA is calculated:
This design requires that the regulator output voltage measured
at the CPU pins drops when the output current increases. The
specified voltage drop corresponds to a dc output resistance of:
1
1
1
1
RA
=
=
= 23.8 kΩ
1
1
1
1
(12)
−
−
−
−
RT ROGM RB
6.31kΩ 1MΩ 8.66 kΩ
VONL −VOFL 1.475V − 1.377V
ROUT
=
=
= 1.5 mΩ
(8)
IO∆
65 A
Choosing the nearest 1% resistor value gives RA = 23.7 kΩ.
CO UT Selection
The required dc output resistance can be achieved by terminating
the gm amplifier with a resistor. The value of the total termina-
tion resistance that will yield the correct dc output resistance:
The required equivalent series resistance (ESR) and capacitance
drive the selection of the type and quantity of the output capaci-
tors. The ESR must be less than or equal to the specified output
resistance (ROUT), in this case 1.5 mΩ. The capacitance must be
large enough that the voltage across the capacitors, which is the
sum of the resistive and capacitive voltage deviations, does not
deviate beyond the initial resistive step while the inductor cur-
rent ramps up or down to the value corresponding to the new
load current.
nI × RSENSE
12.5 × 5 mΩ
3 × 2.2 mmho ×1.5 mΩ
RT
=
=
= 6.31kΩ
(9)
n × gm × ROUT
where nI is the division ratio from the output voltage signal of the
gm amplifier to the PWM comparator CMP1, gm is the transcon-
ductance of the gm amplifier itself, and n is the number of phases.
O utput O ffset
One can, for example, use nine MBZ-type capacitors from
Rubycon, with 2200 µF capacitance, a 6.3 V voltage rating, and
13 mΩ ESR. The nine capacitors have a maximum total ESR of
1.44 mΩ when connected in parallel.
Intel’s specification requires that at no load the nominal output
voltage of the regulator be offset to a lower value than the nominal
voltage corresponding to the VID code. The offset is introduced
by realizing the total termination resistance of the gm amplifier
with a divider connected between the REF pin and ground. The
resistive divider introduces an offset to the output of the gm
amplifier that, when reflected back through the gain of the gm
stage, accurately positions the output voltage near its allowed
maximum at light load. Furthermore, the output of the gm
amplifier sets the current sense threshold voltage. At no load,
the current sense threshold is increased by the peak of the ripple
current in the inductor and reduced by the delay between sens-
ing when the current threshold has been reached and when the
high side MOSFET actually turns off. These two factors are
combined with the inherent voltage (VGNL0), at the output of the
gm amplifier that commands a current sense threshold of 0 mV:
As long as the capacitance of the output capacitor bank is above
a critical value and the regulating loop is compensated with
Analog Devices’ proprietary compensation technique (ADOPT),
the actual capacitance value has no influence on the peak-to-
peak deviation of the output voltage to a full step change in the
load current. The critical capacitance can be calculated as follows:
IO
L
n
COUT (CRIT )
=
×
=
ROUT ×VOUT
65 A
1.5 mΩ ×1.5V
600 nH
(13)
×
= 5.78 mF
3
–10–
REV. 0
ADP3163
The critical capacitance limit for this circuit is 6.93 mF, while
the actual capacitance of the nine Rubycon capacitors is 9 ×
2200 µF = 19.8 mF. In this case, the capacitance is safely above
the critical value.
The nearest standard value of COC is 4.7 nF. The resistance of the
zero-setting resistor in series with the compensating capacitor is:
n
3
RZ
=
=
= 338 Ω
(15)
π × fOSC × COC π × 600 kHz × 4.7 nF
Multilayer ceramic capacitors are also required for high-frequency
decoupling of the processor. The exact number of these MLC
capacitors is a function of the board layout space and parasitics.
Typical designs use twenty to thirty 10 µF MLC capacitors
located as close to the processor power pins as is practical.
The nearest standard 5% resistor value is 330 Ω. Note that this
resistor is only required when COUT approaches CCRIT (within
25% or less). In this example, COUT >> CCRIT, and RZ can
therefore be omitted.
Feedback Loop Com pensation D esign for AD O P T
P ower MO SFETs
Optimized compensation of the ADP3163 allows the best pos-
sible containment of the peak-to-peak output voltage deviation.
Any practical switching power converter is inherently limited by
the inductor in its output current slew rate to a value much less
than the slew rate of the load. Therefore, any sudden change of
load current will initially flow through the output capacitors,
and assuming that the capacitance of the output capacitor is
larger than the critical value defined by Equation 13, this will
produce a peak output voltage deviation equal to the ESR of the
output capacitor times the load current change.
In this example, six N-channel power MOSFETs must be used;
three as the main (control) switches, and the remaining three as
the synchronous rectifier switches. The main selection parameters
for the power MOSFETs are VGS(TH), QG and RDS(ON). The
minimum gate drive voltage (the supply voltage to the ADP3414)
dictates whether standard threshold or logic-level threshold
MOSFETs must be used. Since VGATE <8 V, logic-level thresh-
old MOSFETs (VGS(TH) < 2.5 V) are strongly recommended.
The maximum output current IO determines the RDS(ON) require-
ment for the power MOSFETs. When the ADP3163 is operating
in continuous mode, the simplifying assumption can be made
that in each phase one of the two MOSFETs is always conduct-
The optimal implementation of voltage positioning, ADOPT,
will create an output impedance of the power converter that is
entirely resistive over the widest possible frequency range, includ-
ing dc, and equal to the maximum acceptable ESR of the output
capacitor array. With the resistive output impedance, the output
voltage will droop in proportion with the load current at any
load current slew rate; this ensures the optimal positioning and
allows the minimization of the output capacitor bank.
ing the average inductor current. For VIN = 12 V and VOUT
1.45 V, the duty ratio of the high-side MOSFET is:
=
VOUT 1.5V
DHSF
=
=
= 12.5%
(16)
VIN
The duty ratio of the low-side (synchronous rectifier) MOSFET is:
(17)
12V
With an ideal current-mode-controlled converter, where the
average inductor current would respond without delay to the
command signal, the resistive output impedance could be
achieved by having a single-pole roll-off of the voltage gain of
the voltage-error amplifier. The pole frequency must coincide
with the ESR zero of the output capacitor bank. The ADP3163
uses constant frequency current-mode control, which is known
to have a nonideal, frequency dependent command signal to
inductor current transfer function. The frequency dependence
manifests in the form of a pair of complex conjugate poles at
one-half of the switching frequency. A purely resistive output
impedance could be achieved by canceling the complex conjugate
poles with zeros at the same complex frequencies and adding a
third pole equal to the ESR zero of the output capacitor. Such a
compensating network would be quite complicated. Fortunately, in
practice it is sufficient to cancel the pair of complex conjugate
poles with a single real zero placed at one-half of the switching
frequency. Although the end result is not a perfectly resistive
output impedance, the remaining frequency dependence causes
only a small percentage of deviation from the ideal resistive
response. The single-pole and single-zero compensation can be
easily implemented by terminating the gm error amplifier with
the parallel combination of a resistor (RT) and a series RC net-
work. The value of the terminating resistor RT was determined
previously; the capacitance and resistance of the series RC net-
work are calculated as follows:
DLSF = 1− DHSF = 87.5%
The maximum rms current of the high-side MOSFET during
normal operation is:
2
IL
IO
n
(RIPPLE )
IHSF (MAX )
=
×
DHSF × 1+
=
2
3 × IO
(18)
65 A
3
10.9 A2
3 × 65 A2
× 0.125× 1+
= 7.7 A
The maximum rms current of the low-side MOSFET during
normal operation is:
DLSF
ILSF (MAX ) = IHFS(M AX )
×
=
DHSF
(19)
0.875
0.125
7.7 A ×
= 20.4 A
The RDS(ON) for each MOSFET can be derived from the allowable
dissipation. If 10% of the maximum output power is allowed for
MOSFET dissipation, the total dissipation in the eight MOSFETs
of the four-phase converter will be:
COUT × ROUT
n
COC
=
−
=
RT
π × fOSC × RT
PFET (TOTAL ) = 0.1×VMIN × IO
=
(14)
19.8 mF ×1.5 mΩ
6.31kΩ
3
(20)
−
= 4.4 nF
0.1×1.394V × 65 A = 9.06W
π × 600 kHz × 6.31kΩ
REV. 0
–11–
ADP3163
CIN Selection and Input Cur r ent di/dt Reduction
Allocating half of the total dissipation for the four high-side
MOSFETs and half for the four low-side MOSFETs, and
assuming that the resistive and switching losses of the high-side
MOSFETs are equal, the required maximum MOSFET resis-
tances will be:
In continuous inductor-current mode, the source current of the
high-side MOSFET is approximately a square wave with a duty
ratio equal to VOUT/VIN and an amplitude of one-half of the
maximum output current. To prevent large voltage transients, a
low ESR input capacitor sized for the maximum rms current
must be used. The maximum rms capacitor current is given by:
PFET (TOTAL )
RDS(ON )HSF
=
=
2
4 × n × IHSF (MAX )
IO
×
n × DHSF −(n × DHSF )2 =
(21)
IC(RMS )
=
9.06W
4 × 3 × 7.7 A2
= 12.7 mΩ
n
(25)
65 A
3
× 3 × 0.125 −(3 × 0.125)2 = 10.5 A
and:
PFET (TOTAL )
RDS(ON )LSF
=
=
2
Note that the capacitor manufacturer’s ripple current ratings are
often based on only 2000 hours of life. This makes it advisable
to further derate the capacitor, or to choose a capacitor rated at
a higher temperature than required. Several capacitors may be
placed in parallel to meet size or height requirements in the
design. In this example, the input capacitor bank is formed by
three 270 µF, 16 V OS-CON capacitors with a ripple current
rating of 4.4 A each.
2 × n × ILSF (MAX )
(22)
9.06W
2 × 3 × 20.4 A2
= 3.63 mΩ
Note that there is a trade-off between converter efficiency and
cost. Larger MOSFETs reduce the conduction losses and allow
higher efficiency, but increase the system cost. A Fairchild
FDB7030L (RDS(ON) = 7 mΩ nominal, 10 mΩ worst-case) for
the high-side and a Fairchild FDB8030L (RDS(ON) = 3.1 mΩ
nominal, 5.6 mΩ worst-case) for the low-side are good choices.
The high-side MOSFET dissipation is:
The ripple voltage across the three paralleled capacitors is:
IO
n
ESRC
nC
DHSF
VC(RIPPLE )
=
×
+
=
nC × CIN × fSW
2
PHSF = RDS(ON )HSF × IHSF (MAX )
(26)
65 A
3
18 mΩ
0.125
3 × 270 µF × 200 kHz
VIN × IL(PK ) × QG × fSW
×
+
= 147 mV
+VIN × QRR × fSW
=
3
2 × IG
12V × 29 A × 35 nC × 200 kHz
2 ×1 A
Multilayer ceramic input capacitors are also required. These
10 mΩ × 7.7 A2 +
(23)
+
capacitors should be placed between the input side of the cur-
rent sense resistor and the sources of the low side synchronous
MOSFETS. These capacitors decouple the high frequency leading
edge current spike which supplies the reverse recovery charge of the
low side MOSFETS body diode. The exact number required is a
function of board layout. Typical designs will use two 10 µF
MLC capacitors.
12V ×150 nC × 200 kHz = 2.17W
Where the first term is the conduction loss of the MOSFET, the
second term represents the turn-off loss of the MOSFET and
the third term represents the turn-on loss due to the stored
charge in the body diode of the low-side MOSFET. In the sec-
ond term, QG is the gate charge to be removed from the gate for
turn-off and IG is the gate turn-off current. From the data sheet,
for the FDB7030L the value of QG is about 35 nC and the peak
gate drive current provided by the ADP3414 is about 1 A. In
the third term, QRR, is the charge stored in the body diode of
the low-side MOSFET at the valley of the inductor current. The
data sheet of the FDB8030L does not give that information, so
an estimated value of 150 nC is used. This estimate is based on
information found on data sheets of similar devices. In both
terms, fSW is the actual switching frequency of the MOSFETs,
or 200 kHz. IL(PK) is the peak current in the inductor, or 27 A.)
To reduce the input-current di/dt to below the recommended
maximum of 0.1 A/µs, an additional small inductor (L > 1 µH
@ 15 A) should be inserted between the converter and the sup-
ply bus. That inductor also acts as a filter between the converter
and the primary power source.
The worst-case low-side MOSFET dissipation is:
2
PLSF = RDS(ON )LSF × ILSF (MAX ) = 5.6 mΩ × 20.4 A2 = 2.33W
(24)
Note that there are no switching losses in the low-side MOSFET.
–12–
REV. 0
ADP3163
LAYO UT AND CO MP O NENT P LACEMENT GUID ELINES
The following guidelines are recommended for optimal perfor-
mance of a switching regulator in a PC system.
P ower Cir cuitr y
9. The switching power path should be routed on the PCB to
encompass the smallest possible area in order to minimize
radiated switching noise energy (i.e., EMI). Failure to take
proper precautions often results in EMI problems for the
entire PC system as well as noise-related operational prob-
lems in the power converter control circuitry. The switching
power path is the loop formed by the current path through
the input capacitors, the power MOSFETs, and the power
Schottky diode, if used (see next), including all intercon-
necting PCB traces and planes. The use of short and wide
interconnection traces is especially critical in this path for
two reasons: it minimizes the inductance in the switching
loop, which can cause high-energy ringing, and it accom-
modates the high current demand with minimal voltage loss.
Gener al Recom m endations
1. For good results, at least a four-layer PCB is recommended.
This should allow the needed versatility for control circuitry
interconnections with optimal placement, a signal ground
plane, power planes for both power ground and the input
power (e.g., 12 V), and wide interconnection traces in the
rest of the power delivery current paths. Keep in mind that
each square unit of 1 ounce copper trace has a resistance of
~0.53 mΩ at room temperature.
2. Whenever high currents must be routed between PCB layers,
vias should be used liberally to create several parallel cur-
rent paths so that the resistance and inductance introduced
by these current paths is minimized and the via current
rating is not exceeded.
10. MLC input capacitors should be placed between VIN and
power ground as close to the sources of the low-side
MOSFETS as possible.
3. If critical signal lines (including the voltage and current sense
lines of the ADP3163) must cross through power circuitry,
it is best if a signal ground plane can be interposed between
those signal lines and the traces of the power circuitry. This
serves as a shield to minimize noise injection into the signals at
the expense of making signal ground a bit noisier.
11. To dampen ringing, an RC snubber circuit should be placed
from the SW hole of each phase to ground.
12. An optional power Schottky diode (3 A–5 A dc rating) from
each lower MOSFET’s source (anode) to drain (cathode)
will help to minimize switching power dissipation in the
upper MOSFETs. In the absence of an effective Schottky
diode, this dissipation occurs through the following sequence
of switching events. The lower MOSFET turns off in advance
of the upper MOSFET turning on (necessary to prevent
cross-conduction). The circulating current in the power
converter, no longer finding a path for current through the
channel of the lower MOSFET, draws current through the
inherent body diode of the MOSFET. The upper MOSFET
turns on, and the reverse recovery characteristic of the
lower MOSFET’s body diode prevents the drain voltage
from being pulled high quickly. The upper MOSFET then
conducts very large current while it momentarily has a high
voltage forced across it, which translates into added power
dissipation in the upper MOSFET. The Schottky diode
minimizes this problem by carrying a majority of the circu-
lating current when the lower MOSFET is turned off, and
by virtue of its essentially nonexistent reverse recovery time.
The Schottky diode has to be connected with very short
copper traces to the MOSFET to be effective.
4. The power ground plane should not extend under signal
components, including the ADP3163 itself. If necessary,
follow the preceding guideline to use the signal ground
plane as a shield between the power ground plane and the
signal circuitry.
5. The GND pin of the ADP3163 should be connected first to
the timing capacitor (on the CT pin), and then into the
signal ground plane. In cases where no signal ground plane
can be used, short interconnections to other signal ground
circuitry in the power converter should be used.
6. The output capacitors of the power converter should be
connected to the signal ground plane even though power
current flows in the ground of these capacitors. For this
reason, it is advised to avoid critical ground connections
(e.g., the signal circuitry of the power converter) in the
signal ground plane between the input and output capaci-
tors. It is also advised to keep the planar interconnection
path short (i.e., have input and output capacitors close
together).
13. Whenever a power dissipating component (e.g., a power
MOSFET) is soldered to a PCB, the liberal use of vias,
both directly on the mounting pad and immediately sur-
rounding it, is recommended. Two important reasons for
this are: improved current rating through the vias, and
improved thermal performance from vias extended to the
opposite side of the PCB where a plane can more readily
transfer the heat to the air.
7. The output capacitors should also be connected as closely
as possible to the load (or connector) that receives the power
(e.g., a microprocessor core). If the load is distributed, the
capacitors should also be distributed, and generally in pro-
portion to where the load tends to be more dynamic.
8. Absolutely avoid crossing any signal lines over the switching
power path loop, described below.
REV. 0
–13–
ADP3163
14. The output power path, though not as critical as the switch-
ing power path, should also be routed to encompass a small
area. The output power path is formed by the current path
through the current sensing resistor, the inductors, the output
capacitors, and back to the input capacitors.
Signal Cir cuitr y
16. The output voltage is sensed and regulated between the FB
pin and the GND pin (which connects to the signal ground
plane). The output current is sensed (as a voltage) by the
CS+ and CS– pins. In order to avoid differential mode
noise pickup in the sensed signal, the loop area should be
small. Thus the FB trace should be routed atop the signal
ground plane and the CS+ and CS– pins. (The CS+ pin
should be over the signal ground plane as well.)
15. For best EMI containment, the power ground plane should
extend fully under all the power components except the out-
put capacitors. These components are: the input capacitors,
the power MOSFETs and Schottky diodes, the inductors, the
current sense resistor, and any snubbing element that might
be added to dampen ringing. Avoid extending the power
ground under any other circuitry or signal lines, including
the voltage and current sense lines.
17. The CS+ and CS– traces should be Kelvin-connected to
the current sense resistor, so that the additional voltage
drop due to current flow on the PCB at the current sense
resistor connections, does not affect the sensed voltage.
–14–
REV. 0
ADP3163
O UTLINE D IMENSIO NS
Dimensions shown in inches and (mm).
20-Lead TSSO P
(RU-20)
0.260 (6.60)
0.252 (6.40)
20
11
0.177 (4.50)
0.169 (4.30)
0.256 (6.50)
0.246 (6.25)
10
1
PIN 1
0.006 (0.15)
0.002 (0.05)
0.0433 (1.10)
MAX
8ꢂ
0ꢂ
0.0256 (0.65)
BSC
0.0118 (0.30)
0.0075 (0.19)
0.028 (0.70)
0.020 (0.50)
SEATING
PLANE
0.0079 (0.20)
0.0035 (0.090)
REV. 0
–15–
–16–
相关型号：
ADP3163JRUZ-REEL7
IC SWITCHING CONTROLLER, 3000 kHz SWITCHING FREQ-MAX, PDSO20, TSSOP-20, Switching Regulator or Controller
ADI
ADP3164JRU-REEL
IC SWITCHING CONTROLLER, 4000 kHz SWITCHING FREQ-MAX, PDSO20, TSSOP-20, Switching Regulator or Controller
ADI
ADP3164JRUZ-REEL7
IC SWITCHING CONTROLLER, 4000 kHz SWITCHING FREQ-MAX, PDSO20, TSSOP-20, Switching Regulator or Controller
ADI
©2020 ICPDF网 联系我们和版权申明