AD8251ARMZ [ADI]
10 MHz, 20 V/レs, G = 1, 2, 4, 8 i CMOS㈢ Programmable Gain Instrumentation Amplifier; 10兆赫, 20 V /レS,G = 1 , 2 , 4 , 8我CMOS㈢可编程增益仪表放大器型号: | AD8251ARMZ |
厂家: | ADI |
描述: | 10 MHz, 20 V/レs, G = 1, 2, 4, 8 i CMOS㈢ Programmable Gain Instrumentation Amplifier |
文件: | 总24页 (文件大小:637K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
10 MHz, 20 V/μs, G = 1, 2, 4, 8 iCMOS®
Programmable Gain Instrumentation Amplifier
AD8251
FEATURES
FUNCTIONAL BLOCK DIAGRAM
DGND WR
A1
5
A0
4
Small package: 10-lead MSOP
Programmable gains: 1, 2, 4, 8
Digital or pin-programmable gain setting
Wide supply: 5 V to 15 V
2
6
LOGIC
1
–IN
Excellent dc performance
High CMRR: 98 dB (minimum), G = 8
Low gain drift: 10 ppm/°C (maximum)
Low offset drift: 1.8 μV/°C (maximum), G = 8
Excellent ac performance
7
OUT
10
+IN
Fast settling time: 785 ns to 0.001% (maximum)
High slew rate: 20 V/ꢀs (minimum)
Low distortion: −110 dB THD at 1 kHz,10 V swing
High CMRR over frequency: 80 dB to 50 kHz (minimum)
Low noise: 18 nV/√Hz, G = 10 (maximum)
Low power: 4 mA
AD8251
8
3
–V
9
+V
REF
S
S
Figure 1.
25
APPLICATIONS
20
15
10
5
G = 8
G = 4
Data acquisition
Biomedical analysis
Test and measurement
G = 2
G = 1
GENERAL DESCRIPTION
The AD8251 is an instrumentation amplifier with digitally
programmable gains that has GΩ input impedance, low output
noise, and low distortion, making it suitable for interfacing with
sensors and driving high sample rate analog-to-digital converters
(ADCs). It has high bandwidth of 10 MHz, low THD of −110 dB,
and fast settling time of 785 ns (maximum) to 0.001%. Offset
drift and gain drift are guaranteed to 1.8 μV/°C and 10 ppm/°C,
respectively, for G = 8. In addition to its wide input common
voltage range, it boasts a high common-mode rejection of 80 dB
at G = 1 from dc to 50 kHz. The combination of precision dc
performance coupled with high speed capabilities makes the
AD8251 an excellent candidate for data acquisition. Furthermore,
this monolithic solution simplifies design and manufacturing
and boosts performance of instrumentation by maintaining a
tight match of internal resistors and amplifiers.
0
–5
–10
1k
100M
10k
100k
1M
10M
FREQUENCY (Hz)
Figure 2. Gain vs. Frequency
Table 1. Instrumentation and Difference Amplifiers
by Category
High
Low
High
Mil
Low
Digital
Gain
AD6271 AD82311
Performance Cost
Voltage Grade Power
AD82201
AD8221
AD8222
AD82241
AD6231
AD85531
AD628
AD629
AD620
AD621
AD524
AD526
AD624
AD8250
AD85551
AD85561
AD85571
The AD8251 user interface consists of a parallel port that allows
users to set the gain in one of two different ways (see Figure 1
for the functional block diagram). A 2-bit word sent via a bus
1 Rail-to-rail output.
WR
can be latched using the
input. An alternative is to use
The AD8251 is available in a 10-lead MSOP package and is
specified over the −40°C to +85°C temperature range, making it
an excellent solution for applications where size and packing
density are important considerations.
transparent gain mode where the state of logic levels at the gain
port determines the gain.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registeredtrademarks arethe property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.461.3113
www.analog.com
©2007 Analog Devices, Inc. All rights reserved.
AD8251
TABLE OF CONTENTS
Features .............................................................................................. 1
Power Supply Regulation and Bypassing ................................ 18
Input Bias Current Return Path ............................................... 18
Input Protection ......................................................................... 18
Reference Terminal .................................................................... 19
Common-Mode Input Voltage Range..................................... 19
Layout .......................................................................................... 19
RF Interference ........................................................................... 19
Driving an Analog-to-Digital Converter ................................ 20
Applications..................................................................................... 21
Differential Output .................................................................... 21
Setting Gains with a Microcontroller ...................................... 21
Data Acquisition......................................................................... 22
Outline Dimensions....................................................................... 23
Ordering Guide .......................................................................... 23
Applications....................................................................................... 1
General Description......................................................................... 1
Functional Block Diagram .............................................................. 1
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Timing Diagram ........................................................................... 5
Absolute Maximum Ratings............................................................ 6
Maximum Power Dissipation ..................................................... 6
ESD Caution.................................................................................. 6
Pin Configuration and Function Descriptions............................. 7
Typical Performance Characteristics ............................................. 8
Theory of Operation ...................................................................... 16
Gain Selection............................................................................. 16
REVISION HISTORY
5/07—Revision 0: Initial Version
Rev. 0 | Page 2 of 24
AD8251
SPECIFICATIONS
+VS = +15 V, −VS = −15 V, VREF = 0 V @ TA = 25°C, G = 1, RL = 2 kΩ, unless otherwise noted.
Table 2.
Parameter
Conditions
Min
Typ
Max
Unit
COMMON-MODE REJECTION RATIO (CMRR)
CMRR to 60 Hz with 1 kΩ Source Imbalance
+IN = −IN = −10 V to +10 V
G = 1
G = 2
G = 4
G = 8
80
86
92
98
94
dB
dB
dB
dB
104
105
105
CMRR to 50 kHz
+IN = −IN = −10 V to +10 V
G = 1
G = 2
G = 4
G = 8
80
84
86
86
dB
dB
dB
dB
NOISE
Voltage Noise, 1 kHz, RTI
G = 1
G = 2
G = 4
G = 8
40
27
22
18
nV/√Hz
nV/√Hz
nV/√Hz
nV/√Hz
0.1 Hz to 10 Hz, RTI
G = 1
G = 2
G = 4
G = 8
2.5
2.5
1.8
1.2
μV p-p
μV p-p
μV p-p
μV p-p
pA/√Hz
pA p-p
Current Noise, 1 kHz
Current Noise, 0.1 Hz to 10 Hz
VOLTAGE OFFSET
Offset RTI VOS
5
60
G = 1, 2, 4, 8
200 + 600/G
μV
Over Temperature
Average TC
T = −40°C to +85°C
T = −40°C to +85°C
VS = 5 V to 15 V
260 + 900/G
1.2 + 5/G
6 + 20/G
μV
μV/°C
μV/V
Offset Referred to the Input vs. Supply (PSR)
INPUT CURRENT
Input Bias Current
Over Temperature
Average TC
Input Offset Current
Over Temperature
Average TC
5
5
30
40
400
30
30
nA
nA
pA/°C
nA
nA
T = −40°C to +85°C
T = −40°C to +85°C
T = −40°C to +85°C
T = −40°C to +85°C
160
pA/°C
DYNAMIC RESPONSE
Small Signal −3 dB Bandwidth
G = 1
10
10
8
MHz
MHz
MHz
MHz
G = 2
G = 4
G = 8
2.5
Settling Time 0.01%
G = 1
G = 2
G = 4
G = 8
ΔOUT = 10 V step
615
460
460
625
ns
ns
ns
ns
Rev. 0 | Page 3 of 24
AD8251
Parameter
Conditions
Min
Typ
Max
Unit
Settling Time 0.001%
ΔOUT = 10 V step
G = 1
G = 2
G = 4
G = 8
785
700
700
770
ns
ns
ns
ns
Slew Rate
G = 1
G = 2
G = 4
G = 8
20
30
30
30
V/μs
V/μs
V/μs
V/μs
dB
Total Harmonic Distortion + Noise
f = 1 kHz, RL = 10 kΩ, 10V,
G = 1, 10 Hz to 22 kHz band-
pass filter
−110
GAIN
Gain Range
Gain Error
G = 1, 2, 4, 8
OUT = 10 V
1
8
V/V
G = 1
G = 2, 4, 8
0.03
0.04
%
%
Gain Nonlinearity
G = 1
G = 2
G = 4
G = 8
OUT = −10 V to +10 V
RL = 10 kΩ, 2 kΩ, 600 Ω
RL = 10 kΩ, 2 kΩ, 600 Ω
RL = 10 kΩ, 2 kΩ, 600 Ω
RL = 10 kΩ, 2 kΩ, 600 Ω
All gains
9
ppm
ppm
ppm
ppm
12
12
15
10
Gain vs. Temperature
INPUT
3
ppm/°C
Input Impedance
Differential
1||2pF
1||2pF
GΩ||pF
Common Mode
Input Operating Voltage Range
Over Temperature
OUTPUT
GΩ||pF
V
V
VS = 5 V to 15 V
T = −40°C to +85°C
−VS + 1.5
−VS + 1.6
+VS − 1.5
+VS − 1.7
Output Swing
Over Temperature
Short-Circuit Current
REFERENCE INPUT
RIN
−13.5
−13.5
+13.5
+13.5
V
V
mA
T = −40°C to +85°C
+IN, −IN, REF = 0
37
20
kΩ
μA
V
IIN
1
+VS
Voltage Range
Gain to Output
DIGITAL LOGIC
Digital Ground Voltage, DGND
Digital Input Voltage Low
Digital Input Voltage High
Digital Input Current
Gain Switching Time1
tSU
−VS
1
0
0.0001
V/V
Referred to GND
Referred to GND
Referred to GND
−VS + 4.25
DGND
2.8
+VS − 2.7
2.1
+VS
V
V
V
μA
ns
ns
ns
ns
ns
1
325
See Figure 3 timing diagram
20
10
20
40
tHD
t WR -LOW
t WR -HIGH
Rev. 0 | Page 4 of 24
AD8251
Parameter
Conditions
Min
Typ
Max
Unit
POWER SUPPLY
Operating Range
Quiescent Current, +IS
Quiescent Current, −IS
Over Temperature
TEMPERATURE RANGE
Specified Performance
5
15
4.5
4.5
4.5
V
4.1
3.7
mA
mA
mA
T = −40°C to +85°C
−40
+85
°C
1 Add time for the output to slew and settle to calculate the total time for a gain change.
TIMING DIAGRAM
tWR-HIGH
tWR-LOW
WR
tSU
tHD
A0, A1
Figure 3. Timing Diagram for Latched Gain Mode (See the Timing for Latched Gain Mode Section)
Rev. 0 | Page 5 of 24
AD8251
ABSOLUTE MAXIMUM RATINGS
package due to the load drive for all outputs. The quiescent
power is the voltage between the supply pins (VS) times the
quiescent current (IS). Assuming the load (RL) is referenced to
midsupply, the total drive power is VS/2 × IOUT, some of which is
dissipated in the package and some in the load (VOUT × IOUT).
Table 3.
Parameter
Rating
17 V
Supply Voltage
Power Dissipation
See Figure 4
Indefinite1
VS
VS
VS
Output Short-Circuit Current
Common-Mode Input Voltage
Differential Input Voltage
Digital Logic Inputs
The difference between the total drive power and the load
power is the drive power dissipated in the package.
PD = Quiescent Power + (Total Drive Power − Load Power)
Storage Temperature Range
Operating Temperature Range2
Lead Temperature (Soldering 10 sec)
Junction Temperature
–65°C to +125°C
–40°C to +85°C
300°C
2
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
VS VOUT
VOUT
RL
PD
=
(
VS × IS
)
+
×
–
2
RL
140°C
In single-supply operation with RL referenced to −VS, the worst
case is VOUT = VS/2.
θJA (4-Layer JEDEC Standard Board)
Package Glass Transition Temperature
112°C/W
140°C
Airflow increases heat dissipation, effectively reducing θJA. In
addition, more metal directly in contact with the package leads
from metal traces, through holes, ground, and power planes
reduces the θJA.
1 Assumes the load is referenced to midsupply.
2 Temperature for specified performance is −40°C to +85°C. For performance
to +125°C, see the Typical Performance Characteristics section.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Figure 4 shows the maximum safe power dissipation in the
package vs. the ambient temperature on a 4-layer JEDEC
standard board.
2.00
1.75
1.50
1.25
1.00
0.75
0.50
0.25
0
MAXIMUM POWER DISSIPATION
The maximum safe power dissipation in the AD8251 package is
limited by the associated rise in junction temperature (TJ) on
the die. The plastic encapsulating the die locally reaches the
junction temperature. At approximately 140°C, which is the
glass transition temperature, the plastic changes its properties.
Even temporarily exceeding this temperature limit can change
the stresses that the package exerts on the die, permanently
shifting the parametric performance of the AD8251. Exceeding
a junction temperature of 140°C for an extended period can
result in changes in silicon devices, potentially causing failure.
–40
–20
0
20
40
60
80
100
120
AMBIENT TEMPERATURE (°C)
Figure 4. Maximum Power Dissipation vs. Ambient Temperature
ESD CAUTION
The still-air thermal properties of the package and PCB (θJA),
the ambient temperature (TA), and the total power dissipated in
the package (PD) determine the junction temperature of the die.
The junction temperature is calculated as
TJ = TA
+
PD × θJA
The power dissipated in the package (PD) is the sum of the
quiescent power dissipation and the power dissipated in the
Rev. 0 | Page 6 of 24
AD8251
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
–IN
1
2
3
4
5
10
9
+IN
DGND
REF
AD8251
TOP VIEW
(Not to Scale)
–V
8
+V
S
S
A0
A1
7
OUT
WR
6
Figure 5. 10-Lead MSOP (RM-10) Pin Configuration
Table 4. Pin Function Descriptions
Pin No.
Mnemonic
Description
1
2
3
4
−IN
DGND
−VS
Inverting Input Terminal. True differential input.
Digital Ground.
Negative Supply Terminal.
Gain Setting Pin (LSB).
A0
5
A1
Gain Setting Pin (MSB).
6
WR
Write Enable.
7
8
9
10
OUT
+VS
REF
Output Terminal.
Positive Supply Terminal.
Reference Voltage Terminal.
Noninverting Input Terminal. True differential input.
+IN
Rev. 0 | Page 7 of 24
AD8251
TYPICAL PERFORMANCE CHARACTERISTICS
TA @ 25°C, +VS = +15 V, −VS = −15 V, RL = 10 kꢀ, unless otherwise noted.
2700
2400
2100
1800
1500
1200
900
800
700
600
500
400
300
200
100
0
600
300
0
–30
–20
–10
0
10
20
30
–120
–90
–60
–30
0
30
60
90
120
INPUT OFFSET CURRENT (nA)
CMRR (µV/V)
Figure 9. Typical Distribution of Input Offset Current
Figure 6. Typical Distribution of CMRR, G = 1
90
80
70
60
50
40
30
20
10
0
500
400
300
200
100
0
G = 1
G = 2
G = 4
G = 8
–200
–100
0
100
200
1
100k
10
100
1k
10k
INPUT OFFSET VOLTAGE, V
, RTI (µV)
OSI
FREQUENCY (Hz)
Figure 7. Typical Distribution of Offset Voltage, VOSI
Figure 10. Voltage Spectral Density Noise vs. Frequency
800
600
400
200
0
2µV/DIV
1s/DIV
–30
–20
–10
0
10
20
30
INPUT BIAS CURRENT (nA)
Figure 11. 0.1 Hz to 10 Hz RTI Voltage Noise, G = 1
Figure 8. Typical Distribution of Input Bias Current
Rev. 0 | Page 8 of 24
AD8251
150
130
110
90
G = 4
G = 2
G = 1
70
G = 8
50
30
1.25µV/DIV
1s/DIV
10
10
100
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 12. 0.1 Hz to 10 Hz RTI Voltage Noise, G = 8
Figure 15. Positive PSRR vs. Frequency, RTI
150
130
110
90
18
16
14
12
10
8
G = 4
G = 8
G = 1
70
6
50
4
G = 2
30
2
10
10
0
100
1k
10k
100k
1M
1
100k
10
100
1k
10k
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 13. Current Noise Spectral Density vs. Frequency
Figure 16. Negative PSRR vs. Frequency, RTI
20
15
10
5
I
+
B
I
–
B
0
I
OS
–5
–10
140pA/DIV
1s/DIV
–60 –40 –20
0
20
40
60
80
100 120 140
TEMPERATURE (ºC)
Figure 14. 0.1 Hz to 10 Hz Current Noise
Figure 17. Input Bias Current and Offset Current vs. Temperature
Rev. 0 | Page 9 of 24
AD8251
140
120
100
80
25
20
15
10
5
V
V
R
= ±15V
S
G = 4
G = 8
= 200m Vp-p
= 2kΩ
IN
G = 8
G = 4
LOAD
G = 2
G = 2
G = 1
G = 1
0
60
–5
–10
40
10
100
1k
10k
100k
1M
1M
130
1k
100M
10k
100k
1M
10M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 18. CMRR vs. Frequency
Figure 21. Gain vs. Frequency
140
120
100
40
30
20
G = 8
10
G = 4
0
G = 2
80
60
40
G = 1
–10
–20
–30
–40
–10
–8
–6
–4
–2
0
2
4
6
8
10
10
100
1k
10k
100k
OUTPUT VOLTAGE (V)
FREQUENCY (Hz)
Figure 19. CMRR vs. Frequency, 1 kΩ Source Imbalance
Figure 22. Gain Nonlinearity, G = 1, RL = 10 kΩ, 2 kΩ, 600 Ω
40
30
20
15
10
5
10
0
0
–10
–5
–10
–15
–20
–30
–40
–10
–8
–6
–4
–2
0
2
4
6
8
10
–50
–30
–10
10
30
50
70
90
110
OUTPUT VOLTAGE (V)
TEMPERATURE (°C)
Figure 23. Gain Nonlinearity, G = 2, RL = 10 kΩ, 2 kΩ, 600 Ω
Figure 20. CMRR vs. Temperature, G = 1
Rev. 0 | Page 10 of 24
AD8251
16
12
8
40
30
20
–13V, +13.5V
0V, +13.5V
±15V
+13V, +13V
V
S
0V, +4V
–4V, +4V
+4V, +3.9V
4
10
0
0
V = ±5V
S
–10
–4
–8
–12
–16
–4V, –3.9V 0V, –3.9V +4V, –4V
–20
–30
–40
–13V, –13.1V
0V, –13.5V
0
+13V, –13.5V
16
–10
–8
–6
–4
–2
0
2
4
6
8
10
–16
–12
–8
–4
4
8
12
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
Figure 27. Input Common-Mode Voltage Range vs. Output Voltage, G = 8
Figure 24. Gain Nonlinearity, G = 4, RL = 10 kΩ, 2 kΩ, 600 Ω
+V
S
40
30
20
+85°C
+125°C
–40°C
–1
–2
+25°C
10
0
–10
+2
+1
–40°C
–20
+25°C
–30
–40
+125°C
+85°C
12
–V
S
–10
–8
–6
–4
–2
0
2
4
6
8
10
4
16
6
8
10
14
OUTPUT VOLTAGE (V)
SUPPLY VOLTAGE (±V )
S
Figure 25. Gain Nonlinearity, G = 8, RL = 10 kΩ, 2 kΩ, 600 Ω
Figure 28. Input Voltage Limit vs. Supply Voltage, G = 1, VREF = 0 V, RL = 10 kΩ
16
15
0V, +13.5V
+V
S
12
8
–14.2V, +7.1V
+14V, +7V
0V, ±15V
10
5
FAULT CONDITION
(OVER DRIVEN INPUT)
G = 8
FAULT CONDITION
(OVER DRIVEN INPUT)
G = 8
0V, +3.85V
–4V, +2.2V
+4V, +2V
+4V, –2V
4
+IN
–IN
0
V = ±5V
S
0
–4V, –2V
–4
–8
–12
–16
0V, –3.9V
–5
–10
–15
–V
S
–14.2V, –7.1V
–12
+14V, –7V
12
0V, –13.5V
0
–16
16
–8
–4
4
8
–16
–12
–8
–4
0
4
8
12
16
OUTPUT VOLTAGE (V)
DIFFERENTIAL INPUT VOLTAGE (V)
Figure 26. Input Common-Mode Voltage Range vs. Output Voltage, G = 1
Figure 29. Fault Current Draw vs. Input Voltage, G = 8, RL = 10 kΩ
Rev. 0 | Page 11 of 24
AD8251
+V
+V
S
S
–0.2
–0.4
–0.6
–0.8
–1.0
+85°C
–0.4
–0.8
–1.2
–1.6
–2.0
+2.0
+1.6
+1.2
+0.8
+0.4
+125°C
+125°C
+25°C
–40°C
–40°C
–40°C
+25°C
+25°C
–40°C
+1.0
+0.8
+0.6
+0.4
+0.2
+25°C
+125°C
+125°C
+85°C
–V
–V
S
S
4
16
4
16
6
8
10
12
14
6
8
10
12
14
SUPPLY VOLTAGE (±V )
OUTPUT CURRENT (mA)
S
Figure 30. Output Voltage Swing vs. Supply Voltage, G = 8, RL = 2 kΩ
Figure 33. Output Voltage Swing vs. Output Current
+V
S
100pF
–0.2
–0.4
–0.6
–0.8
–1.0
NO
LOAD
+125°C
47pF
+85°C
–40°C
–40°C
+25°C
+1.0
+0.8
+0.6
+0.4
+0.2
+25°C +85°C
+125°C
20mV/DIV
2µs/DIV
–V
S
4
16
6
8
10
12
14
SUPPLY VOLTAGE (±V )
S
Figure 31. Output Voltage Swing vs. Supply Voltage, G = 8, RL = 10 kΩ
Figure 34. Small Signal Pulse Response for Various Capacitive Loads
15
+25°C
+85°C
10
5
–40°C
+125°C
5V/DIV
0
585ns TO 0.01%
723ns TO 0.001%
–5
–10
–15
0.002%/DIV
+125°C
+85°C
–40°C
+25°C
2µs/DIV
100
10k
1k
LOAD RESISTANCE (Ω)
Figure 32. Output Voltage Swing vs. Load Resistance
Figure 35. Large Signal Pulse Response and Settling Time, G = 1, RL = 10 kΩ
Rev. 0 | Page 12 of 24
AD8251
5V/DIV
400ns TO 0.01%
600ns TO 0.001%
0.002%/DIV
2µs/DIV
25mV/DIV
2µs/DIV
Figure 39. Small Signal Response,
G = 1, RL = 2 kΩ, CL = 100 pF
Figure 36. Large Signal Pulse Response and Settling Time,
G = 2, RL = 10 kΩ
5V/DIV
376ns TO 0.01%
640ns TO 0.001%
0.002%/DIV
2µs/DIV
25mV/DIV
2µs/DIV
Figure 40. Small Signal Response,
G = 2, RL = 2 kΩ, CL = 100 pF
Figure 37. Large Signal Pulse Response and Settling Time,
G = 4, RL = 10 kΩ
5V/DIV
364ns TO 0.01%
522ns TO 0.001%
0.002%/DIV
2µs/DIV
25mV/DIV
2µs/DIV
Figure 38. Large Signal Pulse Response and Settling Time,
G = 8, RL = 10 kΩ
Figure 41. Small Signal Response,
G = 4, RL = 2 kΩ, CL = 100 pF
Rev. 0 | Page 13 of 24
AD8251
1200
1000
800
600
400
200
0
SETTLED TO 0.001%
SETTLED TO 0.01%
25mV/DIV
2µs/DIV
2
20
4
6
8
10
12
14
16
18
STEP SIZE (V)
Figure 42. Small Signal Response, G = 8, RL = 2 kΩ, CL = 100 pF
Figure 45. Settling Time vs. Step Size, G = 4, RL = 10 kΩ
1200
1000
1200
1000
800
600
400
200
0
SETTLED TO 0.001%
800
SETTLED TO 0.001%
SETTLED TO 0.01%
600
SETTLED TO 0.01%
400
200
0
2
20
4
6
8
10
12
14
16
18
2
20
4
6
8
10
12
14
16
18
STEP SIZE (V)
STEP SIZE (V)
Figure 43. Settling Time vs. Step Size, G = 1, RL = 10 kΩ
Figure 46. Settling Time vs. Step Size, G = 8, RL = 10 kΩ
1200
1000
800
600
400
200
0
–50
–55
–60
–65
–70
–75
–80
SETTLED TO 0.001%
–85
G = 8
–90
G = 4
–95
SETTLED TO 0.01%
–100
–105
–110
–115
–120
G = 2
G = 1
2
20
4
6
8
10
12
14
16
18
10
1M
100
1k
10k
100k
STEP SIZE (V)
FREQUENCY (Hz)
Figure 44. Settling Time vs. Step Size, G = 2, RL = 10 kΩ
Figure 47. Total Harmonic Distortion vs. Frequency,
10 Hz to 22 kHz Band-Pass Filter, 2 kΩ Load
Rev. 0 | Page 14 of 24
AD8251
–50
–55
–60
–65
–70
–75
–80
G = 8
G = 4
–85
G = 2
–90
–95
–100
–105
–110
–115
–120
G = 1
1k
10
100
10k
100k
1M
FREQUENCY (Hz)
Figure 48. Total Harmonic Distortion vs. Frequency,
10 Hz to 500 kHz Band-Pass Filter, 2 kΩ Load
Rev. 0 | Page 15 of 24
AD8251
THEORY OF OPERATION
+V
–V
+V
–V
S
S
A0
A1
2.2kΩ
+V
–V
S
S
S
2.2kΩ
–IN
10kΩ
10kΩ
A1
S
+V
S
DIGITAL
GAIN
OUT
REF
A3
CONTROL
–V
+V
S
S
+V
–V
S
10kΩ
10kΩ
A2
+IN
2.2kΩ
–V
S
+V
–V
+V
S
S
S
2.2kΩ
DGND
WR
–V
S
S
Figure 49. Simplified Schematic
The AD8251 is a monolithic instrumentation amplifier based
on the classic, three op amp topology, as shown in Figure 49.
It is fabricated on the Analog Devices, Inc. proprietary iCMOS
process that provides precision, linear performance, and a robust
digital interface. A parallel interface allows users to digitally
program gains of 1, 2, 4, and 8. Gain control is achieved by
switching resistors in an internal, precision, resistor array (as
shown in Figure 49). Although the AD8251 has a voltage feed-
back topology, gain bandwidth product increases for gains of 1,
2, and 4 because each gain has its own frequency compensation.
This results in maximum bandwidth at higher gains.
Transparent Gain Mode
The easiest way to set the gain is to program it directly via a
logic high or logic low voltage applied to A0 and A1. Figure 50
shows an example of this gain setting method, referred to through-
WR
out the data sheet as transparent gain mode. Tie
to the
negative supply to engage transparent gain mode. In this mode,
any change in voltage applied to A0 and A1 from logic low to
logic high, or vice versa, immediately results in a gain change.
Table 5 is the truth table for transparent gain mode, and Figure 50
shows the AD8251 configured in transparent gain mode.
+15V
All internal amplifiers employ distortion cancellation circuitry
and achieve high linearity and ultralow THD. Laser trimmed
resistors allow for a maximum gain error of less than 0.03% for
G = 1 and minimum CMRR of 98 dB for G = 8. A pinout opti-
mized for high CMRR over frequency enables the AD8251 to
offer a guaranteed minimum CMRR over frequency of 80 dB at
50 kHz (G = 1). The balanced input reduces the parasitics that,
in the past, had adversely affected CMRR performance.
10μF
0.1µF
WR
–15V
+5V
A1
A0
+IN
+5V
G = 8
AD8251
REF
–IN
DGND
DGND
GAIN SELECTION
10μF
0.1µF
This section shows users how to configure the AD8251 for
basic operation. Logic low and logic high voltage limits are
listed in the Specifications section. Typically, logic low is 0 V
and logic high is 5 V; both voltages are measured with respect
to DGND. Refer to the specifications table (Table 2) for the
permissible voltage range of DGND. The gain of the AD8251
can be set using two methods.
–15V
NOTE:
1. IN TRANSPARENT GAIN MODE, WR IS TIED TO −V .
S
THE VOLTAGE LEVELS ON A0 AND A1 DETERMINE
THE GAIN. IN THIS EXAMPLE, BOTH A0 AND A1 ARE
SET TO LOGIC HIGH, RESULTING IN A GAIN OF 8.
Figure 50. Transparent Gain Mode, A0 and A1 = High, G = 8
Rev. 0 | Page 16 of 24
AD8251
Table 5. Truth Table Logic Levels for Transparent Gain Mode
Table 6. Truth Table Logic Levels for Latched Gain Mode
WR
A1
A0
Gain
WR
A1
A0
Gain
−VS
−VS
−VS
−VS
Low
Low
High
High
Low
High
Low
High
1
2
4
8
High to Low
High to Low
High to Low
High to Low
Low to Low
Low to High
High to High
Low
Low
High
High
X1
Low
High
Low
High
X1
Change to 1
Change to 2
Change to 4
Change to 8
No Change
No Change
No Change
X1
X1
Latched Gain Mode
X1
X1
Some applications have multiple programmable devices such as
multiplexers or other programmable gain instrumentation
amplifiers on the same PCB. In such cases, devices can share a
data bus. The gain of the AD8251 can be set using
allowing other devices to share A0 and A1. Figure 51 shows a
schematic using this method, known as latched gain mode. The
1 X = don’t care.
On power-up, the AD8251 defaults to a gain of 1 when in
latched gain mode. In contrast, if the AD8251 is configured in
transparent gain mode, it starts at the gain indicated by the
voltage levels on A0 and A1 on power-up.
WR
as a latch,
WR
AD8251 is in this mode when
low, typically 5 V and 0 V, respectively. The voltages on A0 and
WR
is held at logic high or logic
Timing for Latched Gain Mode
A1 are read on the downward edge of the
signal as it
In latched gain mode, logic levels at A0 and A1 have to be held
for a minimum setup time, tSU, before the downward edge of
transitions from logic high to logic low. This latches in the logic
levels on A0 and A1, resulting in a gain change. See the truth
table listing in Table 6 for more on these gain changes.
+15V
WR
latches in the gain. Similarly, they must be held for a
WR
minimum hold time of tHD after the downward edge of
to
ensure that the gain is latched in correctly. After tHD, A0 and A1
may change logic levels but the gain does not change (until the
next downward edge of
can be held high is t WR-HIGH, and t WR-LOW is the minimum
WR
+5V
0V
WR
A1
10μF
0.1µF
WR WR
+5V
0V
A1
). The minimum duration that
+5V
0V
A0
A0
+IN
WR
duration that
can be held low. Digital timing specifications
+
–
G = PREVIOUS G = 8
STATE
are listed in Table 2. The time required for a gain change is
dominated by the settling time of the amplifier. A timing
diagram is shown in Figure 52.
AD8251
REF
–IN
When sharing a data bus with other devices, logic levels applied
to those devices can potentially feed through to the output of
the AD8251. Feedthrough can be minimized by decreasing the
edge rate of the logic signals. Furthermore, careful layout of the
PCB also reduces coupling between the digital and analog
portions of the board.
DGND
DGND
10μF
0.1µF
–15V
NOTE:
1. ON THE DOWNWARD EDGE OF WR, AS IT TRANSITIONS
FROM LOGIC HIGH TO LOGIC LOW, THE VOLTAGES ON A0
AND A1 ARE READ AND LATCHED IN, RESULTING IN A
GAIN CHANGE. IN THIS EXAMPLE, THE GAIN SWITCHES TO G = 8.
Figure 51. Latched Gain Mode, G = 8
tWR-HIGH
tWR-LOW
WR
tSU
tHD
A0, A1
Figure 52. Timing Diagram for Latched Gain Mode
Rev. 0 | Page 17 of 24
AD8251
INCORRECT
+V
CORRECT
+V
POWER SUPPLY REGULATION AND BYPASSING
S
S
The AD8251 has high PSRR. However, for optimal performance,
a stable dc voltage should be used to power the instrumentation
amplifier. Noise on the supply pins can adversely affect per-
formance. As in all linear circuits, bypass capacitors must be
used to decouple the amplifier.
AD8251
AD8251
REF
REF
REF
REF
Place a 0.1 ꢁF capacitor close to each supply pin. A 10 ꢁF tanta-
lum capacitor can be used farther away from the part (see
Figure 53) and, in most cases, it can be shared by other
precision integrated circuits.
–V
–V
S
S
TRANSFORMER
+V
TRANSFORMER
+V
S
S
+V
S
0.1µF
WR
A1
10µF
AD8251
AD8251
A0
+IN
–IN
REF
V
OUT
10Mꢀ
AD8251
LOAD
–V
–V
S
S
REF
THERMOCOUPLE
THERMOCOUPLE
DGND
+V
+V
S
S
0.1µF
10µF
C
C
C
–V
DGND
S
R
1
fHIGH-PASS
=
Figure 53. Supply Decoupling, REF, and Output Referred to Ground
AD8251
2πRC
AD8251
C
REF
INPUT BIAS CURRENT RETURN PATH
R
The AD8251 input bias current must have a return path to its
local analog ground. When the source, such as a thermocouple,
cannot provide a return current path, one should be created
(see Figure 54).
–V
S
–V
S
CAPACITIVELY COUPLED
CAPACITIVELY COUPLED
Figure 54. Creating an IBIAS Path
INPUT PROTECTION
All terminals of the AD8251 are protected against ESD. Note
that 2.2 kꢀ series resistors precede the ESD diodes as shown in
Figure 49. They limit current into the diodes and allow for dc
overload conditions 13 V above the positive supply and 13 V
below the negative supply. An external resistor should be used
in series with each of the inputs to limit current for voltages
greater than 13 V beyond either supply rail. In either scenario,
the AD8251 safely handles a continuous 6 mA current at room
temperature. For applications where the AD8251 encounters
extreme overload voltages, external series resistors and low
leakage diode clamps such as BAV199Ls, FJH1100s, or SP720s
should be used.
Rev. 0 | Page 18 of 24
AD8251
Coupling Noise
REFERENCE TERMINAL
To prevent coupling noise onto the AD8251, follow these
guidelines:
The reference terminal, REF, is at one end of a 10 kꢀ resistor
(see Figure 49). The instrumentation amplifier output is
referenced to the voltage on the REF terminal; this is useful
when the output signal needs to be offset to voltages other than
its local analog ground. For example, a voltage source can be
tied to the REF pin to level shift the output so that the AD8251
can interface with a single-supply ADC. The allowable reference
voltage range is a function of the gain, common-mode input,
and supply voltages. The REF pin should not exceed either +VS
or −VS by more than 0.5 V.
•
•
•
Do not run digital lines under the device.
Run the analog ground plane under the AD8251.
Shield fast switching signals with digital ground to avoid
radiating noise to other sections of the board, and never
run them near analog signal paths.
•
•
Avoid crossover of digital and analog signals.
Connect digital and analog ground at one point only
(typically under the ADC).
For best performance, especially in cases where the output is
not measured with respect to the REF terminal, source imped-
ance to the REF terminal should be kept low because parasitic
resistance can adversely affect CMRR and gain accuracy.
•
Power supply lines should use large traces to ensure a low
impedance path. Decoupling is necessary; follow the
guidelines listed in the Power Supply Regulation and
Bypassing section.
INCORRECT
CORRECT
Common-Mode Rejection
AD8251
AD8251
The AD8251 has high CMRR over frequency, giving it greater
immunity to disturbances, such as line noise and its associated
harmonics, in contrast to typical in amps whose CMRR falls off
around 200 Hz. They often need common-mode filters at the
inputs to compensate for this shortcoming. The AD8251 is able
to reject CMRR over a greater frequency range, reducing the
need for input common-mode filtering.
V
REF
V
REF
+
OP1177
–
Figure 55. Driving the Reference Pin
Careful board layout maximizes system performance. To
maintain high CMRR over frequency, lay out the input traces
symmetrically. Ensure that the traces maintain resistive and
capacitive balance; this holds for additional PCB metal layers
under the input pins and traces. Source resistance and capaci-
tance should be placed as close to the inputs as possible. Should
a trace cross the inputs (from another layer), it should be routed
perpendicular to the input traces.
COMMON-MODE INPUT VOLTAGE RANGE
The three op amp architecture of the AD8251 applies gain and
then removes the common-mode voltage. Therefore, internal
nodes in the AD8251 experience a combination of both the
gained signal and the common-mode signal. This combined
signal can be limited by the voltage supplies even when the
individual input and output signals are not. Figure 26 and
Figure 27 show the allowable common-mode input voltage
ranges for various output voltages, supply voltages, and gains.
RF INTERFERENCE
RF rectification is often a problem when amplifiers are used in
applications where there are strong RF signals. The disturbance
can appear as a small dc offset voltage. High frequency signals
can be filtered with a low-pass, RC network placed at the input
of the instrumentation amplifier, as shown in Figure 56. The
filter limits the input signal bandwidth according to the
following relationship:
LAYOUT
Grounding
In mixed-signal circuits, low level analog signals need to be
isolated from the noisy digital environment. Designing with the
AD8251 is no exception. Its supply voltages are referenced to an
analog ground. Its digital circuit is referenced to a digital ground.
Although it is convenient to tie both grounds to a single ground
plane, the current traveling through the ground wires and PC
board can cause an error. Therefore, use separate analog and
digital ground planes. Only at one point, star ground, should
analog and digital ground meet.
1
FilterFreqDIFF
=
2 π R(2CD + CC )
1
FilterFreqCM
=
2 π RCC
The output voltage of the AD8251 develops with respect to the
potential on the reference terminal. Take care to tie REF to the
appropriate local analog ground or to connect it to a voltage that
is referenced to the local analog ground.
where CD ≥ 10 CC.
Rev. 0 | Page 19 of 24
AD8251
+15V
In this example, a 1 nF capacitor and a 49.9 Ω resistor create an
antialiasing filter for the AD7612. The 1 nF capacitor also serves
to store and deliver necessary charge to the switched capacitor
input of the ADC. The 49.9 ꢀ series resistor reduces the burden
of the 1 nF load from the amplifier and isolates it from the
kickback current injected from the switched capacitor input of
the AD7612. Selecting too small a resistor improves the
correlation between the voltage at the output of the AD8251
and the voltage at the input of the AD7612 but may destabilize
the AD8251. A trade-off must be made between selecting a
resistor small enough to maintain accuracy and large enough to
maintain stability.
0.1µF
+IN
10µF
C
C
C
C
D
C
R
R
V
OUT
AD8251
REF
–IN
0.1µF
10µF
–15V
+15V
Figure 56. RFI Suppression
10μF
0.1µF
Values of R and CC should be chosen to minimize RFI.
Mismatch between the R × CC at the positive input and the
R × CC at negative input degrades the CMRR of the AD8251.
By using a value of CD that is 10 times larger than the value of
CC, the effect of the mismatch is reduced and performance is
improved.
WR
+12V
0.1μF
–12V
0.1μF
A1
A0
+IN
49.9ꢀ
AD8251
AD7612
1nF
REF
+5V
ADR435
–IN
DRIVING AN ANALOG-TO-DIGITAL CONVERTER
DGND
DGND
An instrumentation amplifier is often used in front of an
analog-to-digital converter to provide CMRR. Usually,
instrumentation amplifiers require a buffer to drive an ADC.
However, the low output noise, low distortion, and low settle
time of the AD8251 make it an excellent ADC driver.
10μF
0.1µF
–15V
Figure 57. Driving an ADC
Rev. 0 | Page 20 of 24
AD8251
APPLICATIONS
When using this circuit to drive a differential ADC, VREF can be
set using a resistor divider from the ADC reference to make the
output ratiometric with the ADC.
DIFFERENTIAL OUTPUT
In certain applications, it is necessary to create a differential
signal. High resolution analog-to-digital converters often
require a differential input. In other cases, transmission over
a long distance can require differential signals for better
immunity to interference.
SETTING GAINS WITH A MICROCONTROLLER
+15V
10μF
0.1µF
Figure 59 shows how to configure the AD8251 to output a
differential signal. An op amp, the AD817, is used in an
inverting topology to create a differential voltage. VREF sets the
output midpoint according to the equation shown in the figure.
Errors from the op amp are common to both outputs and are
thus common mode. Likewise, errors from using mismatched
resistors cause a common-mode dc offset error. Such errors are
rejected in differential signal processing by differential input
ADCs or instrumentation amplifiers.
WR
A1
MICRO-
CONTROLLER
A0
+IN
+
–
AD8251
REF
–IN
DGND
DGND
10μF
0.1µF
–15V
Figure 58. Programming Gain Using a Microcontroller
+12V
0.1μF
AMPLITUDE
WR
+5V
A1
A0
+IN
AMPLITUDE
–5V
+
V
A = V + V
IN REF
OUT
2
+2.5V
0V
–2.5V
AD8251
V
G = 1
IN
TIME
REF
–
4.99kꢀ
0.1μF
DGND
–12V
V
–
+
REF
0V
–12V
10pF
+12V
AD817
4.99kꢀ
AMPLITUDE
0.1µF
0.1µF
–12V
+12V
10μF
+2.5V
0V
–2.5V
10μF
DGND
V
B = –V + V
IN
OUT
REF
TIME
2
Figure 59. Differential Output with Level Shift
Rev. 0 | Page 21 of 24
AD8251
–70
–80
DATA ACQUISITION
The AD8251 makes an excellent instrumentation amplifier
for use in data acquisition systems. Its wide bandwidth, low
distortion, low settling time, and low noise enable it to
condition signals in front of a variety of 16-bit ADCs.
–90
–100
–110
–120
–130
–140
–150
–160
–170
–180
Figure 61 shows a schematic of the AD825x data acquisition
demonstration board. The quick slew rate of the AD8251 allows
it to condition rapidly changing signals from the multiplexed
inputs. An FPGA controls the AD7612, AD8251, and ADG1209.
In addition, mechanical switches and jumpers allow users to pin
strap the gains when in transparent gain mode.
This system achieved −106 dB of THD at 1 kHz and a signal-to-
noise ratio of 91 dB during testing, as shown in Figure 60.
0
50
5
10
15
20
25
30
35
40
45
FREQUENCY (kHz)
Figure 60. FFT of the AD825x DAQ Demo Board
Using the AD8251 1 kHz Signal
JMP
JMP
–V
S
+12V
–12V
+12V
14
+
+
+5V
0.1µF
10µF
10µF
2kꢀ
GND
2
V
DD
DGND
806ꢀ
806ꢀ
EN
DGND
2
JMP
4
5
S1A
S2A
+CH1
+CH2
+5V
2kꢀ
DGND
806ꢀ
806ꢀ
ALTERA
EPF6010ATC144-3
+CH3
+CH4
6
7
S3A
S4A
6
DGND
0ꢀ 0ꢀ
0ꢀ 0ꢀ
C
C
5
WR
DGND
VOUT
+IN
8
10
1
+
4
A1
ADG1209
S4B
+IN
A0
VREF
9
806ꢀ
7
AD7612
ADR435
AD8251
C
–CH4
10
11
12
D
0ꢀ 49.9ꢀ
–IN
1nF
806ꢀ
806ꢀ
9
–V
3
–
S
S3B
S2B
–CH3
–CH2
–CH1
C
+V
8
C
S
15
A0
806ꢀ
S1B
1
A1
16
C4
0.1µF
C3
0.1µF
V
SS
3
+12V –12V
JMP
0.1µF
+5V
–12V
2kꢀ
DGND
JMP
+5V
R8
2kꢀ
DGND
Figure 61. Schematic of ADG1209, AD8251, and AD7612 in the AD825x DAQ Demo Board
Rev. 0 | Page 22 of 24
AD8251
OUTLINE DIMENSIONS
3.10
3.00
2.90
10
6
5.15
4.90
4.65
3.10
3.00
2.90
1
5
PIN 1
0.50 BSC
0.95
0.85
0.75
1.10 MAX
0.80
0.60
0.40
8°
0°
0.15
0.05
0.33
0.17
SEATING
PLANE
0.23
0.08
COPLANARITY
0.10
COMPLIANT TO JEDEC STANDARDS MO-187-BA
Figure 62. 10-Lead Mini Small Outline Package [MSOP]
(RM-10)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD8251ARMZ1
AD8251ARMZ-RL1
AD8251ARMZ-R71
AD8251-EVALZ1
Temperature Range
–40°C to +85°C
–40°C to +85°C
Package Description
10-Lead MSOP
10-Lead MSOP
10-Lead MSOP
Evaluation Board
Package Option
RM-10
RM-10
Branding
H0T
H0T
–40°C to +85°C
RM-10
H0T
1 Z = RoHS Compliant Part.
Rev. 0 | Page 23 of 24
AD8251
NOTES
©2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06287-0-5/07(0)
Rev. 0 | Page 24 of 24
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ADI
AD8253ARMZ-R71
10 MHz, 20 V/レs, G = 1, 10, 100, 1000 i CMOS㈢ Programmable Gain Instrumentation Amplifier
ADI
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