TPS40060PWPR [TI]
WIDE-INPUT SYNCHRONOUS BUCK CONTROLLER; 宽输入同步降压控制器
| 型号： | TPS40060PWPR |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | WIDE-INPUT SYNCHRONOUS BUCK CONTROLLER |
| 文件： | 总28页 (文件大小：363K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
TPS40060
8
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
WIDE-INPUT SYNCHRONOUS BUCK CONTROLLER
DESCRIPTION
FEATURES
•
•
•
•
Operating Input Voltage 10 V to 55 V
Input Voltage Feed-Forward Compensation
< 1% Internal 0.7-V Reference
The TPS40060 and TPS40061 are high-voltage, wide
input (10 V to 55 V) synchronous, step-down con-
verters.
This family of devices offers design flexibility with a
variety of user programmable functions, including;
soft-start, UVLO, operating frequency, voltage
feed-forward, high-side current limit, and loop com-
pensation. These devices are also synchronizable to
an external supply.
Programmable Fixed-Frequency, Up to 1-MHz
Voltage Mode Controller
Internal Gate Drive Outputs for High-Side
P-Channel and Synchronous N-Channel
MOSFETs
•
•
•
•
•
16-Pin PowerPAD™ Package (θJC = 2°C/W)
Thermal Shutdown
Externally Synchronizable
Programmable High-Side Sense Short Circuit
Protection
Programmable Closed-Loop Soft-Start
The TPS40060 and TPS40061 incorporate MOSFET
gate drivers for external P-channel high-side and
N-channel synchronous rectifier (SR) MOSFETs.
Gate drive logic incorporates anti-cross conduction
circuitry to prevent simultaneous high-side and
synchronous rectifier conduction.
•
•
TPS40060 Source Only/TPS40061 Source/Sink
APPLICATIONS
•
•
•
•
Networking Equipment
Telecom Equipment
Base Stations
Servers
SIMPLIFIED APPLICATION DIAGRAM
TPS40060PWP
1
2
KFF
16
15
ILIM
RT
VIN
V
IN
3
4
5
6
7
8
BP5
SYNC
HDRV 14
BPN10 13
12
SGND
SS/SD
VFB
SW
+
BP10 11
LDRV 10
V
OUT
COMP
PGND
9
-
UDG-02157
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2002–2004, Texas Instruments Incorporated
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
These devices have limited built-in ESD protection. The leads should be shorted together or the device
placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
TA
LOAD CURRENT
SOURCE(2)
SOURCE/SIN(2)
PACKAGE(1)
PART NUMBER
TPS40060PWP
TPS40061PWP
Plastic HTSSOP (PWP)
Plastic HTSSOP (PWP)
–40°C to 85°C
(1) The PWP package is also available taped and reeled. Add an R suffix to the device type (i.e., TPS40060PWPR). See the application
section of the data sheet for PowerPAD drawing and layout information.
(2) See Application Information section.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range unless otherwise noted(1)
TPS40060
TPS40061
VIN
60 V
VFB, SS/SD, SYNC
–0.3 V to 6 V
VIN
Input voltage range
–0.3 V to 60 V or VIN+5 V
(whichever is less)
SW
SW. transient < 50 ns
–2.5 V
–0.3 V to 6 V
5 mA
VOUT
IIN
Output voltage range
Input current
COMP, RT, KFF, SS
KFF
RT
IOUT
TJ
Output current
200 µA
Operating junction temperature range
Storage temperature
–40°C to 125°C
–55°C to 150°C
260°C
Tstg
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN NOM MAX UNIT
VIN
TA
Input voltage
10
55
85
V
Operating free-air temperature
–40
°C
(1)(2)
PWP PACKAGE
(TOP VIEW)
1
2
3
16
15
14
13
12
11
10
9
KFF
RT
BP5
SYNC
SGND
SS/SD
VFB
ILIM
VIN
HDRV
BPN10
SW
BP10
LDRV
PGND
THERMAL
PAD
4
5
6
7
8
COMP
(1) For more information on the PWP package, refer to TI Technical Brief (SLMA002).
(2) PowerPAD™ heat slug must be connected to SGND (Pin 5), or electrically isolated from all other pins.
2
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
ELECTRICAL CHARACTERISTICS
TA = –40°C to 85°C, VIN = 24 Vdc, RT = 165 kΩ, IKFF = 113 µA, fSW = 300 kHz, all parameters at zero power dissipation (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
INPUT SUPPLY
VIN Input voltage range, VIN
OPERATING CURRENT
IDD Quiescent current
5-V REFERENCE
10
55
2.5
5.5
V
mA
V
Output drivers not switching
1.5
5.0
VBP5
Input voltage
4.5
270
2
OSCILLATOR/RAMP GENERATOR(1)
fOSC
Frequency
300
2
330 kHz
VRAMP PWM ramp voltage(2)
VIH
VIL
High-level input voltage, SYNC
Low-level input voltage, SYNC
V
0.8
ISYNC Input current, SYNC
Pulse width, SYNC
5
10
µA
ns
V
Pulse amplitude = 5 V
50
2.32
85%
VRT
RT voltage
2.50
2.68
98%
0%
Maximum duty cycle
VFB = 0 V, 100 kHz ≤ fSW≤ 1 MHz
Minumum duty cycle
Feed-forward voltage
Feed-forward current operating range(3)
VFB ≥ 0.75 V
VKFF
IKFF
3.35
20
3.50
3.65
1100
V
µA
SS/SD (SOFT START)
ISS
Soft-start source current
Soft-start clamp voltage
1.8
3.1
2.3
3.7
2.9
4.0
µA
V
VSS
tDSCH Discharge time
tSS Soft-start time
SS/SD (SHUTDOWN)
CSS = 220 pF
1.8
2.2
2.8
µs
CSS = 220 pF, 0 V ≤ VSS ≤ 1.6 V
120
155
190
VSD
VEN
Shutdown threshold voltage
Device action threshold voltage
90
120
210
145
260
mV
mV
160
10-V REFERENCE
VBP10 Input voltage
ERROR AMPLIFIER
9.0
9.7
10.7
V
V
TA = 25°C
0.698 0.700 0.704
0.690 0.700 0.707
0.690 0.700 0.715
VFB
Feedback regulation voltage
0°C ≤ TA ≤ 85°C
GBW
AVOL
IOH
Gain bandwidth
3
60
5
80
MHz
dB
Open loop gain
High-level output source current
Low-level output sink current
Input bias current
VCOMP = 2.0 V, VFB = 0 V
VCOMP = 2.0 V, VFB = 1 V
VFB = 0.7 V
1.5
2.5
4.0
mA
nA
V
IOL
4.0
IBIAS
VOH
VOL
100
3.45
300
High-level output voltage
Low-level output voltage
IOH = 0.5 mA, VFB = 0 V
IOL = 0.5 mA, VFB = 1 V
3.25
3.60
0.050 0.215 0.350
(1) KFF current (IKFF) increases with SYNC frequency (fSYNC) and decreases with maximum duty cycle (DMAX).
(2) Ensured by design. Not production tested.
(3) Ensured by design. Not production tested.
3
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
ELECTRICAL CHARACTERISTICS (continued)
TA = –40°C to 85°C, VIN = 24 Vdc, RT = 165 kΩ, IKFF = 113 µA, fSW = 300 kHz, all parameters at zero power dissipation (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
CURRENT LIMIT
ISINK Current limit sink current
8.3
10.0
330
275
11.5
500
375
µA
ns
VILIM = 23.7 V, VSW = (VILIM– 0.5 V)
VILIM = 23.7 V, VSW = (VILIM– 2 V)
tDELAY Propagation delay to output
tON
Switch leading-edge blanking pulse time(4)
Off time during a fault
100
tOFF
VOS
7
cycles
mV
Overcurrent comparator offset voltage
-200
-60
50
OUTPUT DRIVER
tHFALL High-side driver fall time(4)
tHRISE High-side driver rise time(4)
tLFALL Low-side driver fall time(4)
tLRISE Low-side driver rise time(4)
CHDRV = 2200 pF, (VIN– VBPN10
CHDRV = 2200 pF, (VIN– VBPN10
CLDRV = 2200 pF, BP10
)
)
48
36
24
48
1.0
96
72
ns
V
48
CLDRV = 2200 pF, BP10
96
VOH
VOL
VOH
VOL
High-level ouput voltage, HDRV
Low-level ouput voltage, HDRV
High-level ouput voltage, LDRV
Low-level ouput voltage, LDRV
Minimum controllable pulse width
IHDRV = 0.1 A , (VIN– VHDRV
)
1.4
0.75
1.5
0.5
150
IHDRV = 0.1 A , (VHDRV– VBPN10
)
ILDRV = 0.1 A, (VBP10– VLDRV
ILDRV = 0.1 A
)
1.0
100
ns
V
BPN10 REGULATOR
VBPN1
0
Output voltage
Outputs off
–7.5
–6
–8.5
0
–9.5
RECTIFIER ZERO CURRENT COMPARATOR (TPS40060 ONLY)
VSW
Switch voltage
LDRV output OFF
6
1
mV
µA
SW NODE
ILEAK
Leakage current(4)
THERMAL SHUTDOWN
Shutdown temperature(4)
Hysteresis(4)
165
25
TSD
°C
UNDERVOLTAGE LOCKOUT
VUVLO Undervoltage lockout threshold voltage, BP10
Undervoltage lockout hysteresis
RKFF = 10 kΩ
6.25
9
6.5
0.4
10
7.5
11
V
VKFF
KFF programmable threshold voltage
RKFF = 82.5 kΩ
(4) Ensured by design. Not production tested.
4
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
Terminal Functions
TERMINAL
I/O
DESCRIPTION
NAME
NO.
5-V reference. This pin should be bypassed to ground with a 0.1-µF ceramic capacitor. This pin may be used with
an external DC load of 1 mA or less.
BP5
3
O
O
O
10-V reference used for gate drive of the N-channel synchronous rectifier. This pin should be bypassed by a 1-µF
ceramic capacitor. This pin may be used with an external DC load of 1 mA or less.
BP10
BPN10
11
13
Negative 8-V reference with respect to VIN. This voltage is used to provide gate drive for the high side P-channel
MOSFET. This pin should be bypassed to VIN with a 0.1-µF capacitor
Output of the error amplifier, input to the PWM comparator. A feedback network is connected from this pin to the
VFB pin to compensate the overall loop. The comp pin is internally clamped above the peak of the ramp to
improve large signal transient response.
COMP
HDRV
ILIM
8
I
O
I
Floating gate drive for the high-side P-channel MOSFET. This pin switches from VIN (MOSFET off) to BPN10
(MOSFET on).
14
16
Current limit pin, used to set the overcurrent threshold. An internal current sink from this pin to ground sets a
voltage drop across an external resistor connected from this pin to VIN. The voltage on this pin is compared to the
voltage drop (VIN -SW) across the high side MOSFET during conduction.
A resistor is connected from this pin to VIN to program the amount of voltage feed-forward. The current fed into
this pin is internally divided and used to control the slope of the PWM ramp.
KFF
1
10
9
I
I
Gate drive for the N-channel synchronous rectifier. This pin switches from BP10 (MOSFET on) to ground
(MOSFET off).
LDRV
PGND
Power ground reference for the device. There should be a low-impedance connection from this point to the source
of the power MOSFET.
A resistor is connected from this pin to ground to set the internal oscillator ramp charging current and switching
frequency.
RT
2
5
I
I
SGND
Signal ground reference for the device.
Soft-start programming pin. A capacitor connected from this pin to ground programs the soft-start time. The
capacitor is charged with an internal current source of 2.3 µA. The resulting voltage ramp on the SS pin is used as
a second non-inverting input to the error amplifier. The output voltage begins to rise when VSS/SD is approximately
0.85 V. The output continues to rise and reaches regulation when VSS/SD is approximately 1.55 V. The controller is
considered shut down when VSS/SD is 125 mV or less. All internal circuitry is inactive. The internal circuitry is
enabled when VSS/SD is 210 mV or greater. When VSS/SD is less than approximately 0.85 V, the outputs cease
switching and the output voltage (VOUT) decays while the internal circuitry remains active.
SS/SD
6
This pin is connected to the switched node of the converter and used for overcurrent sensing. This pin is used for
zero current sensing in the TPS40060.
SW
12
4
I
I
Synchronization input for the device. This pin can be used to synchronize the oscillator to an external master
frequency.
SYNC
Inverting input to the error amplifier. In normal operation the voltage on this pin is equal to the internal reference
voltage, 0.7 V.
VFB
VIN
7
I
I
15
Supply voltage for the device.
5
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
SIMPLIFIED BLOCK DIAGRAM
ILIM
16
BP10
VIN 15
11
BP10
+
CLK
CLK
7
13 BPN10
7
RT
2
4
Clock
10−V Regulator
VIN
7
Oscillator
SYNC
1V5REF
7 HDRV
CL
7
HDRV
7
07VREF
7
Ramp Generator
7
7
7
7
1V5REF
Reference
Voltages
3−bit up/down
Fault Counter
3V5REF
BP5
P-Channel
Driver
HDRV
14
KFF
1
7
Restart Fault
BPN10
12 SW
BP5
BP5
3
8
7
BP10
7
Fault
COMP
7
7
S
R
Q
Q
07VREF
CL
7
+
+
VFB
7
N-Channel
Driver
10
LDRV
0.85 V
+
+
SW
7
07VREF
S
R
Q
Q
7
SS/SD
6
CLK
7
Zero Current Detector
(TPS40060 Only)
9
PGND
Restart
5
UDG−02160
SGND
6
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION
The TPS40060/61 family of parts allows the user to optimize the PWM controller to the specific application.
The TPS40061 is the controller of choice for synchronous buck designs which will include most applications. It
has two quadrant operation and will source or sink output current. This provides the best transient response.
The TPS40060 operates in one quadrant and sources output current only, allowing for paralleling of converters
and ensures that one converter does not sink current from another converter. This controller also emulates a
standard buck converter at light loads where the inductor current goes discontinuous. At continuous output
inductor currents the controller operates as a synchronous buck converter to optimize efficiency.
SW NODE RESISTOR
The SW node of the converter will be negative during the dead time when both the upper and lower MOSFETs
are off. The magnitude of this negative voltage is dependent on the lower MOSFET body diode and the output
current which flows during this dead time. This negative voltage could affect the operation of the controller,
especially at low input voltages.
Therefore, a 10-Ω resistor must be placed between the lower MOSFET drain and pin 12 (SW) of the controller as
shown in Figure 13 as RSW
.
SETTING THE SWITCHING FREQUENCY (PROGRAMMING THE CLOCK OSCILLATOR)
The TPS40060 and TPS40061 have independent clock oscillator and ramp generator circuits. The clock
oscillator serves as the master clock to the ramp generator circuit. The switching frequency, fSW in kHz, of the
clock oscillator is set by a single resistor (RT) to ground. The clock frequency is related to RT, in kΩ by
Equation 1 and the relationship is charted in Figure 2.
1
R + ǒ
* 23ǓkW
T
*6
f
17.82 10
SW
(1)
PROGRAMMING THE RAMP GENERATOR CIRCUIT
The ramp generator circuit provides the actual ramp used by the PWM comparator. The ramp generator provides
voltage feed-forward control by varying the PWM ramp slope with line voltage, while maintaining a constant ramp
magnitude. Varying the PWM ramp directly with line voltage provides excellent response to line variations since
the PWM does not have to wait for loop delays before changing the duty cycle. (See Figure 1).
VIN
VIN
SW
SW
RAMP
V
PEAK
COMP
COMP
RAMP
V
VALLEY
T
1
T
2
t
ON2
t
ON1
tON
T
d +
t
> t
and d > d
ON2 1 2
ON1
UDG-02131
Figure 1. Voltage Feed-Forward Effect on PWM Duty Cycle
7
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
The PWM ramp must be faster than the master clock frequency or the PWM is prevented from starting. The
PWM ramp time is programmed via a single resistor (RKFF) pulled up to VIN. RKFF is related to RT, and the
minimum input voltage, VIN(min) through the following:
* 3.5 ǒ65.27 R ) 1502
Ǔ
+ ǒV
Ǔ
R
(W)
KFF
IN (min)
T
(2)
where:
•
VIN is the desired start-up (UVLO) input voltage
•
RT is the timing resistor in kΩ
See the section on UVLO operation for further description.
The curve showing the feedforward impedance required for a given switching frequency, fSW, at various input
voltages is shown in Figure 3.
For low input voltage and high duty cycle applications, the voltage feed-forward may limit the duty cycle
prematurely. This does not occur for most applications. The voltage control loop controls the duty cycle and
regulates the output voltages. For more information on large duty cycle operation, refer to Application Note
(SLUA310).
TIMING RESISTANCE
vs
SWITCHING FREQUENCY
FEED-FORWARD IMPEDANCE
vs
SWITCHING FREQUENCY
600
800
700
600
V
IN
= 25 V
500
400
300
200
500
400
V
IN
= 15 V
300
200
100
0
V
IN
= 9 V
100
0
0
200
f
400
600
800
1000
200
400
600
800
1000
- Switching Frequency - kHz
SW
f
SW
- Switching Frequency - kHz
Figure 3.
Figure 2.
UVLO OPERATION
The TPS40060 and TPS40061 use both fixed and variable (user programmable) UVLO protection. The fixed
UVLO monitors the BP10 and BP5 bypass voltages. The UVLO circuit holds the soft-start low until the BP5 and
BP10 voltage rails have exceeded their thresholds and the input voltage has exceed the user programmable
undervoltage threshold.
The TPS40060 and TPS40061 use the feed-forward pin, KFF, as a user programmable low-line UVLO detection.
This variable low-line UVLO threshold compares the PWM ramp duration to the oscillator clock period. An
undervoltage condition existis if the device receives a clock pulse before the ramp has reached 90% of its full
amplitude. The ramp duration is a function of the ramp slope, which is directly related to the current into the KFF
pin. The KFF current is a function of the input voltage and the resistance from KFF to the input voltage. The KFF
resistor can be referenced to the oscillator frequency as descibed in Equation 3:
8
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
* 3.5 ǒ65.27 R ) 1502
Ǔ
+ ǒV
Ǔ
R
(W)
KFF
IN (min)
T
(3)
where:
•
VIN is the desired start-up (UVLO) input voltage
•
RT is the timing resistor in kΩ
The variable UVLO function utilizes a 3-bit full adder to prevent spurious shut-downs or turn-ons due to spikes or
fast line transients. When the adder reaches a total of seven counts in which the ramp duration is shorter the
clock cycle a powergood signal is asserted, a soft-start initiated, and the upper and lower MOSFETs are turned
off.
Once the soft-start is initiated, the UVLO cicruit must see a total count of seven cycles in which the ramp
duration is longer than the clock cycle before an undervoltage condition is declared (See Figure 4).
UVLO Threshold
VIN
Clock
PWM RAMP
1
2
3
4
5
6
7
1
2
1 2 3 4 5 6 7
PowerGood
UDG-02132
Figure 4. Undervoltage Lockout Operation
UNDERVOLTAGE LOCKOUT
vs
HYSTERESIS
3.0
2.5
2.0
1.5
1.0
0.5
0
10
15
20
25
30
35
40
45
50
45
V
UVLO
- Undervoltage Lockout Threshold - V
Figure 5.
9
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
The impedance of the input voltage can cause the input voltage, at the TPS4006x, to sag when the converter
starts to operate and draw current from the input source. Therefore, there is voltage hysteresis that prevents
nuisance shutdowns at the UVLO point.
With RT chosen to select the operating frequency and RKFF chosen to select the start-up voltage, the amount of
hysteresis voltage is shown in Figure 5.
PROGRAMMING SOFT START
TPS4006x uses a closed-loop approach to ensure a controlled ramp on the output during start-up. Soft-start is
programmed by charging an external capacitor (CSS) via an internally generated current source. The voltage on
CSS minus 0.85 V, is fed into a separate non-inverting input to the error amplifier (in addition to FB and 0.7-V
VREF). The loop is closed on the lower of the (CSS– 0.85 V) voltage or the internal reference voltage ( 0.7-V
VREF). Once the (CSS– 0.85 V) voltage rises above the internal reference voltage, regulation is based on the
internal reference. To ensure a controlled ramp-up of the output voltage the soft-start time should be greater than
the L-CO time constant as described in Equation 4.
ǸL C
t
w 2p
(seconds)
START
O
(4)
There is a direct correlation between tSTART and the input current required during start-up. The faster tSTART, the
higher the input current required during start-up. This relationship is describe in more detail in the section titled,
Programming the Current Limit which follows. The soft-start capacitance, CSS, is described in Equation 5.
For applications in which the VIN supply ramps up slowly, (typically between 50 ms and 100 ms) it may be
necessary to increase the soft-start time to between approximately 2 ms and 5 ms to prevent nuisance UVLO
tripping. The soft-start time should be longer than the time that the VINsupply transitions between 6 V and 7 V.
2.3 mA
0.7 V
C
+
t
(Farads)
START
SS
(5)
10
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
PROGRAMMING CURRENT LIMIT
This device uses a two-tier approach for overcurrent protection. The first tier is a pulse-by-pulse protection
scheme. Current limit is implemented on the high-side MOSFET by sensing the voltage drop across the
MOSFET when the gate is driven low. The MOSFET voltage is compared to the voltage dropped across a
resistor connected from VIN pin to the ILIM pin when driven by a constant current sink. If the voltage drop across
the MOSFET exceeds the voltage drop across the ILIM resistor, the switching pulse is immediately terminated.
The MOSFET remains off until the next switching cycle is initiated.
The second tier consists of a fault counter. The fault counter is incremented on an overcurrent pulse and
decremented on a clock cycle without an overcurrent pulse. When the counter reaches seven (7) a restart is
issued and seven soft-start cycles are initiated. Both the upper and lower MOSFETs are turned off during this
period. The counter is decremented on each soft-start cycle. When the counter is decremented to zero, the PWM
is re-enabled. If the fault has been removed the output starts up normally. If the output is still present the counter
counts seven overcurrent pulses and re-enters the second-tier fault mode. See Figure 6 for typical overcurrent
protection waveforms.
The minimum current limit setpoint (ILIM) depends on tSTART, CO, VO, and the load current at turn-on (IL).
ǒC
OǓ
V
O
I
+
) I (A)
L
ƪ ƫ
LIM
t
SS
(6)
The current limit programming resistor (RILIM) is calculated using Equation 7. Care must be taken in choosing the
values used fro VOS and ISINK in the equation. In order to ensure the output current at the overcurrent level, the
minimum value of ISINK and the maximum value of VOS must be used.
I
R
V
OC
DS(on)[max]
OS
R
+
)
(W)
ILIM
I
I
SINK
SINK
(7)
where:
•
•
•
ISINK is the current into the ILIM pin and is nominally 8.3 µA, minimum
OC is the overcurrent setpoint which is the DC output current plus one-half of the peak inductor current
OS is the overcurrent comparator offset and is 50 mV maximum
I
V
BP5, BPN10 AND BPN10 INTERNAL VOLTAGE REGULATOR
Start-up characteristics of the BP5, BP10 and BPN10 regulators are shown in Figure 6. Slight variations in the
BP5 occurs dependent upon the switching frequency. Variation in the BPN10 and BP10 regulation characteristics
is also based on the load presented by switching the external MOSFETs.
11
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
INTERNAL REGULATOR OUTPUT VOLTAGE
vs
INPUT VOLTAGE
12
BP10
10
8
BP5
6
BPN10
4
2
0
2
4
6
8
10
12
V
IN
- Input Voltage - V
Figure 6.
HDRV
CLOCK
t
BLANKING
V
V
ILIM
-V
VIN SW
SS
7 CURRENT LIMIT TRIPS
(HDRV CYCLE TERMINATED BY CURRENT LIMIT TRIP)
UDG-02136
7 SOFT-START CYCLES
Figure 7. Typical Current Limit Protection Waveforms
CALCULATING THE BPN10 AN BP10V BYPASS CAPACITOR
The BPN10 capacitance provides a local, low impedance source for the high-side driver. The BPN10 capacitor
should be a good quality, high-frequency capacitor. The size of the bypass capacitor depends on the total gate
charge of the MOSFET and the amount of droop allowed on the bypass capacitor. The BPN10 capacitance is
described in Equation 8.
12
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
Q
g
C
+
(F)
BPN10
DV
(8)
The 10-V reference pin, BP10V needs to provide energy for the synchronous MOSFET gate drive via the BP10V
capacitor. Neglecting any efficiency penalty, the BP10V capacitance is described in Equation 9.
Q
gSR
C
+
(F)
BP10V
DV
(9)
SYNCHRONIZING TO AN EXTERNAL SUPPLY
The TPS4006x can be synchronized to an external clock through the SYNC pin. The SW node rises on the
falling edge of the SYNC signal. The synchronization frequency should be in the range of 20% to 30% higher
than its programmed free-run frequency. The clock frequency at the SYNC pin replaces the master clock
generated by the oscillator circuit. Pulling the SYNC pin low programs the TPS4006x to freely run at the
frequency programmed by RT.
Internally, the SYNC pin has a pull-down current between 5 µA and 10 µA. In order to synchronize the device to
an external clock signal, the SYNC pin has to be overdriven from the external clock circuit. Normal logic gates or
an external MOSFET with a pull-up resistor of 10 kΩ is adequate.
Internally there is a delay of between approximately 50 ns and 100 ns from the time the SYNC pin is pulled low
and the HDRV signal goes high to turn on the upper MOSFET. Additionally, there is some delay as the MOSFET
gate charges to turn on the upper MOSFET, typically between 20 ns and 50 ns.
The higher synchronization must be factored in when programming the PWM ramp generator circuit. If the PWM
ramp is interrupted by the SYNC pulse, a UVLO condition is declared and the PWM becomes disabled. Typically
this is of concern under low-line conditions only. In any case, RKFF needs to be adjusted for the higher switching
frequency. In order to specify the correct value for RKFF at the synchronizing frequency, calculate a 'dummy'
value for RT that would cause the oscillator to run at the synchronizing frequency. Do not use this value of RT in
the design.
1
+ ǒ
* 23ǓkW
R
T(dummy)
*6
f
17.82 10
SYNC
(10)
(11)
Use the value of RT(dummy) to calculate the value for RKFF
.
+ ǒV
* 3.5 VǓ ǒ65.27 R
) 1502Ǔ kW
R
KFF
IN(min)
T(dummy)
where:
•
RT is in kΩ
This value of RKFF ensures that UVLO is not engaged when operating at the synchronization frequency.
SELECTING THE INDUCTOR VALUE
The inductor value determines the magnitude of ripple current in the output capacitors as well as the load current
at which the converter enters discontinuous mode. Too large an inductance results in lower ripple current but is
physically larger for the same load current. Too small an inductance results in larger ripple currents and a greater
number of (or more expensive output capacitors for) the same output ripple voltage requirement. A good
compromise is to select the inductance value such that the converter doesn't enter discontinuous mode until the
load approximated somewhere between 10% and 30% of the rated output. The inductance value is described in
Equation 12.
ǒV
V
Ǔ
* V V
IN
IN
O
O
L +
(H)
DI f
SW
(12)
where:
•
VO is the output voltage
•
∆I is the peak-to-peak inductor current
13
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
CALCULATING THE OUTPUT CAPACITANCE
The output capacitance depends on the output ripple voltage requirement, output ripple current, as well as any
output voltage deviation requirement during a load transient.
The output ripple voltage is a function of both the output capacitance and capacitor ESR. The worst case output
ripple is described in Equation 13.
1
ǒVP*PǓ
DV + DI ESR )
ǒ
Ǔ
ƪ
ƫ
8 C f
O
SW
(13)
The output ripple voltage is typically between 90% and 95% due to the ESR component.
The output capacitance requirement typically increases in the presence of a load transient requirement. During a
step load, the output capacitance must provide energy to the load (light to heavy load step) or absorb excess
inductor energy (heavy-to-light load step) while maintaining the output voltage within acceptable limits. The
amount of capacitance depends on the magnitude of the load step, the speed of the loop and the size of the
inductor.
Stepping the load from a heavy load to a light load results in an output overshoot. Excess energy stored in the
inductor must be absorbed by the output capacitance. The energy stored in the inductor is described in
Equation 14 and Equation 15.
1
2
E + L I (J)
L
2
(14)
where:
OHǓ2 ǒ Ǔ2
2
2
ǒ
ǒ(
) Ǔ
I + ƪI
ƫ
* I
Amperes
OL
(15)
where:
•
IOH is the output current under heavy load conditions
OL is the output current under light load conditions
•
I
Energy in the capacitor is given by the following equation:
1
2
E
+
C V (J)
C
2
(16)
(17)
where:
V + ǒV Ǔ2 * ǒV Ǔ2 ǒVolts
2
2
Ǔ
f
i
where:
•
Vf is the final peak capacitor voltage
Vi is the initial capacitor voltage
•
By substituting Equation 14 into Equation 13, substituting Equation 16 into Equation 15, setting Equation 13
equal to Equation 15 and solving for CO yields the following equation.
ǒ
OHǓ2 ǒ Ǔ2
ƪI
ƫ
(F)
L
* I
OL
C
+
O
ǒ Ǔ2 ǒ Ǔ2
ƪV
ƫ
* V
f
i
(18)
Loop Compensation
Voltage-mode buck-type converters are typically compensated using Type III networks. Since the TPS40060 and
TPS40061 use voltage feedforward control, the gain of the PWM modulator with voltage feedforward circuit must
be included. The modulator gain is described in Figure 7, with VIN being the minimum input voltage required to
cause the ramp excursion to cover the entire switching period.
14
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
V
V
IN
V
S
IN
+ 20 log ǒ Ǔ
A
+
or
A
MOD(db)
MOD
V
S
(19)
Duty dycle, D, varies from 0 to 1 as the control voltage, VC, varies from the minimum ramp voltage to the
maximum ramp voltage, VS. Also, for a synchronous buck converter, D = VO / VIN. To get the control voltage to
output voltage modulator gain in terms of the input voltage and ramp voltage,
V
V
V
V
O
C
S
O
C
IN
D +
+
or
+
V
V
V
V
IN
S
(20)
Calculate the Poles and Zeros
For a buck converter using voltage mode control there is a double pole due to the output L-CO. The double pole
is located at the frequency calculated in Equation 21.
1
f
+
(Hz)
LC
ǸL C
2p
O
(21)
There is also a zero created by the output capacitance, CO, and its associated ESR. The ESR zero is located at
the frequency calculated in Equation 22.
1
f +
(Hz)
Z
2p ESR C
O
(22)
(23)
(24)
Calculate the value of RBIASto set the output voltage, VOUT
.
0.7 R1
R
+
W
BIAS
V
* 0.7
OUT
The maximum crossover frequency (0 dB loop gain) is calculated in Equation 24.
f
SW
4
f
+
(Hertz)
C
Typically, fC is selected to be close to the midpoint between the L-CO double pole and the ESR zero. At this
frequency, the control to output gain has a –2 slope (-40 dB/decade), while the Type III topology has a +1 slope
(20 dB/decade), resulting in an overall closed loop –1 slope (–20 dB/decade). Figure 9 shows the modulator
gain, L-C filter, output capacitor ESR zero, and the resulting response to be compensated.
A Type III topology, shown in Figure 10, has two zero-pole pairs in addition to a pole at the origin. The gain and
phase boost of a Type III topology is shown in Figure 11. The two zeros are used to compensate the L-CO
double pole and provide phase boost. The double pole is used to compensate for the ESR zero and provide
controlled gain roll-off. In many cases the second pole can be eliminated and the amplifier's gain roll-off used to
roll-off the overall gain at higher frequencies.
15
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
PWM MODULATOR RELATIONSHIPS
MODULATOR GAIN
vs
SWITCHING FREQUENCY
ESR Zero, + 1
A
MOD
= V / V
IN
S
V
S
Resultant, - 1
V
C
D = V / V
C
S
LC Filter, - 2
100
1 k
10 k
100 k
f
SW
- Switching Frequency - Hz
Figure 8.
Figure 9.
C2
(optional)
− 1
+ 1
0 dB
C1
R2
R3
− 1
GAIN
−90°
C3
VFB
7
R1
180°
PHASE
8
COMP
V
OUT
−270°
+
R
BIAS
VREF
UDG−02189
Figure 10. Type III Compensation of Configuration
Figure 11. Type III Compensation Gain and Phase
The poles and zeros for a type III network are described in Equation 25.
1
1
f
+
(Hz)
f
+
Z2
(Hz)
Z1
2p R2 C1
2p R1 C3
(25)
1
1
f
+
(Hz)
f
+
(Hz)
P1
P2
2p R2 C2
2p R3 C3
The value of R1 is somewhat arbitraty, but influences other component values. A value between 50kΩ and
100kΩ usually yields reasonable values.
The unity gain frequency is described in Equation 26.
1
f
+
(Hertz)
C
2p R1 C2 G
(26)
where G is the reciprocal of the modulator gain at fC.
16
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
The modulator gain as a function of frequency at fC, is described in Equation 27.
2
f
LC
1
ǒ Ǔ
AMOD(f) + AMOD
and G +
f
AMOD(f)
C
(27)
Care must be taken not to load down the output of the error amplifier with the feedback resistor, R2, that is too
small. The error amplifier has a finite output source and sink current which must be considered when sizing R2.
Too small a value does not allow the output to swing over its full range.
V
C (max)
3.45 V
2.0 mA
R2
+
(W) +
+ 1.725 kW
(MIN)
I
SOURCE (min)
(28)
dv/dt INDUCED TURN-ON
MOSFETs are susceptible to dv/dt turn-on particularly in high-voltage (VDS) applications. The turn-on is caused
by the capacitor divider that is formed by CGD and CGS. High dv/dt conditions and drain-to-source voltage, on the
MOSFET causes current flow through CGD and causes the gate-to-source voltage to rise. If the gate-to-source
voltage rises above the MOSFET threshold voltage, the MOSFET turns on, resulting in large shoot-through
currents. Therefore the SR MOSFET should be chosen so that the CGD capacitance is smaller than the CGS
capacitance. A 2-Ω to 5-Ω resistor in the upper MOSFET gate lead shapes the turn-on and dv/dt of the SW node
and helps reduce the induced turn-on.
HIGH-SIDE MOSFET POWER DISSIPATION
The power dissipated in the external high-side MOSFET is comprised of conduction and switching losses. The
conduction losses are a function of the IRMS current through the MOSFET and the RDS(on) of the MOSFET. The
high-side MOSFET conduction losses are defined by Equation 29.
+ ǒIRMSǓ2
O
1 ) TC ƪT * 25 C
ƫ
ǒ
Ǔ
P
R
(W)
COND
DS(on)
R
J
(29)
where:
•
TCR is the temperature coefficient of the MOSFET RDS(on)
The TCR varies depending on MOSFET technology and manufacturer but is typically ranges between 0.0035
ppm/°C and 0.010 ppm/°C.
The IRMS current for the high side MOSFET is described in Equation 30.
Ǹ
ǒAmperesRMSǓ
I
+ I d
RMS
O
(30)
(31)
The switching losses for the high-side MOSFET are descibed in Equation 31.
+ ǒV
Ǔ
f
P
I
t
(Watts)
SW
SW(fsw)
IN
OUT
SW
where:
•
•
•
IO is the DC output current
tSW is the switching rise time, typically < 20 ns
SW is the switching frequency
f
Typical switching waveforms are shown in Figure 12.
17
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
I
D2
I
O
∆I
}
I
D1
d
1-d
BODY DIODE
CONDUCTION
BODY DIODE
CONDUCTION
SW
0
SYNCHRONOUS
RECTIFIER ON
ANTI-CROSS
CONDUCTION
HIGH SIDE ON
UDG-02179
Figure 12. Inductor Current and SW Node Waveforms
The maximum allowable power dissipation in the MOSFET is determined by the following equation.
ǒT * TAǓ
J
P +
(W)
T
q
JA
(32)
where:
P + P
) P
(W)
T
COND
SW(fsw)
(33)
and ΘJA is the package thermal impedance.
SYNCHRONOUS RECTIFIER MOSFET POWER DISSIPATION
The power dissipated in the synchronous rectifier MOSFET is comprised of three components: RDS(on) conduction
losses, body diode conduction losses, and reverse recovery losses. RDS(on) conduction losses can be found
using Equation 29 and the RMS current through the synchronous rectifier MOSFET is described in Equation 34.
Ǹ
ǒARMSǓ
+ I 1 * d
O
I
RMS
(34)
The body-diode conduction losses are due to forward conduction of the body diode during the anti-cross
conduction delay time. The body diode conduction losses are described by Equation 35.
P
+ 2 I V t
f
(W)
DC
O
F
DELAY
SW
(35)
where:
•
VF is the body diode forward voltage
DELAY is the delay time just before the SW node rises
•
t
The 2-multiplier is used because the body-diode conducts twice during each cycle (once on the rising edge and
once on the falling edge)
The reverse recovery losses are due to the time it takes for the body diode to recovery from a forward bias to a
reverse blocking state. The reverse recovery losses are described in Equation 36.
P
+ 0.5 Q V f
(W)
RR
RR
IN
SW
(36)
where:
•
QRR is the reverse recovery charge of the body diode
The total synchronous rectifier MOSFET power dissipation is described in Equation 37.
18
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
P
+ P ) P ) P
(W)
COND
SR
DC
RR
(37)
TPS40060/TPS40061 POWER DISSIPATION
The power dissipation in the TPS40060 and TPS40061 is largely dependent on the MOSFET driver currents and
the input voltage. The driver current is proportional to the total gate charge, Qg, of the external MOSFETs. Driver
power (neglecting external gate resistance, refer to [2] can be calculated from Equation 38.
P
+ Q V f
(W)
g
D
DR
SW
(38)
And the total power dissipation in the device, assuming MOSFETs with similar gate charges for both the
high-side and synchronous rectifier is described in Equation 39.
2 P
D
P + ǒ Ǔ
) I
V
(W)
T
Q
IN
V
DR
(39)
(40)
or
P + ƪǒ2 Q f Ǔ ) I
ƫ
V (W)
IN
g
T
SW
Q
where:
•
IQ is the quiescent operating current (neglecting drivers)
The maximum power capability of the device's PowerPad package is dependent on the layout as well as air flow.
The thermal impedance from junction to air, assuming 2 oz. copper trace and thermal pad with solder and no air
flow.
ΘJA = 36.51°C/W
The maximum allowable package power dissipation is related to ambient temperature by Equation 36.
Substituting Equation 32 into Equation 40 and solving for fSW yields the maximum operating frequency for the
TPS40060 and TPS40061. The result is:
ǒ
AǓ
DDǓ
V
T *T
J
ƪ ƫ* I
ǒ Ǔ
Q
ǒ
q
JA
f
+
(Hz)
SW
ǒ2 QgǓ
(41)
19
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
LAYOUT CONSIDERATIONS
THE PowerPAD™ PACKAGE
The PowerPAD package provides low thermal impedance for heat removal from the device. The PowerPAD
derives its name and low thermal impedance from the large bonding pad on the bottom of the device. For
maximum thermal performance, the circuit board must have an area of solder-tinned-copper underneath the
package. The dimensions of this area depends on the size of the PowerPAD package. For a 16-pin TSSOP
(PWP) package the dimensions of the circuit board pad are 5 mm x 3.4 mm. The dimensions of the package pad
are shown in Figure 13.
Thermal vias connect this area to internal or external copper planes and should have a drill diameter sufficiently
small so that the via hole is effectively plugged when the barrel of the via is plated with copper. This plug is
needed to prevent wicking the solder away from the interface between the package body and the solder-tinned
area under the device during solder reflow. Drill diameters of 0.33 mm (13 mils) works well when 1-oz copper is
plated at the surface of the board while simultaneously plating the barrel of the via. If the thermal vias are not
plugged when the copper plating is performed, then a solder mask material should be used to cap the vias with a
diameter equal to the via diameter of 0.1 mm minimum. This capping prevents the solder from being wicked
through the thermal vias and potentially creating a solder void under the package. Refer to PowerPAD Thermally
Enhanced Package[2] for more information on the PowerPAD package.
X: Minimum PowerPAD = 1,86 mm
Y: Minimum PowerPAD = 1,75 mm
Thermal Pad
6,60 mm
4,30 mm 6,20 mm
4,50 mm
X
1
10
Y
Figure 13. PowerPAD Dimensions
MOSFET PACKAGING
MOSFET package selection depends on MOSFET power dissipation and the projected operating conditions. In
general, for a surface-mount applications, the DPAK style package provides the lowest thermal impedance (θJA)
and, therefore, the highest power dissipation capability. However, the effectiveness of the DPAK depends on
proper layout and thermal management. The θJAspecified in the MOSFET data sheet refers to a given copper
area and thickness. In most cases, a thermal impedance of 40°C/W requires one square inch of 2-ounce copper
on a G-10/FR-4 board. Lower thermal impedances can be achieved at the expense of board area. Please refer
to the selected MOSFET's data sheet for more information regarding proper mounting.
GROUNDING AND CIRCUIT LAYOUT CONSIDERATIONS
The device provides separate signal ground (SGND) and power ground (PGND) pins. It is important that circuit
grounds are properly separated. Each ground should consist of a plane to minimize its impedance if possible.
The high power noisy circuits such as the output, synchronous rectifier, MOSFET driver decoupling capacitor
(BP10), and the input capacitor should be connected to PGND plane at the input capacitor.
Sensitive nodes such as the FB resistor divider, RT, and ILIM should be connected to the SGND plane. The
SGND plane should only make a single point connection to the PGND plane.
Component placement should ensure that bypass capacitors (BP10, BP5, and BPN10) are located as close as
possible to their respective power and ground pins. Also, sensitive circuits such as FB, RT and ILIM should not
be located near high dv/dt nodes such as HDRV, LDRV, BPN10, and the switch node (SW).
20
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
DESIGN EXAMPLE
•
•
•
•
•
•
•
Input voltage: 18 VDC to 55 VDC
Output voltage: 3.3 V ±2%
Output current: 5 A (maximum, steady-state), 7 A (surge, 10-ms duration, 10% duty cycle maximum)
Output ripple: 33 mVP-P at 5 A
Output load response: 0.3 V => 10% to 90% step load change
Operating temperature: –40°C to 85°C
fSW = 130 kHz
1. Calculate maximum and minimum duty cycles
V
V
O(min)
O(max)
d
+
+ 0.0588
d
+
MAX
++ 0.187
MIN
V
V
IN(max)
IN(min)
(42)
2. Select switching frequency
The switching frequency is based on the minimum duty cycle ratio and the propagation delay of the current limit
comparator. In order to maintain current limit capability, the on time of the upper MOSFET, tON, must be greater
than 330 ns (see Electrical Characteristics table). Therefore
V
t
O(min)
ON
+
or
V
T
SW
IN(max)
(43)
V
O(min)
ȡǒ Ǔȣ
V
IN(max)
+ȧ
ȧ
ȧ
ȧ
1
+ f
ȧ
SW
T
T
SW
ON
ȧ
Ȣ
Ȥ
(44)
(45)
Using 400 ns to provide margin,
0.0588
f
+
+ 147 kHz
SW
400 ns
Since the oscillator can vary by 10%, decrease fSW, by 10%
fSW = 0.9 × 147 kHz = 130 kHz
and therefore choose a frequency of 130 kHz.
3. Select∆ I
In this case ∆I is chosen so that the converter enters discontinuous mode at 20% of nominal load.
DI + I 2 0.2 + 5 2 0.2 + 2.0 A
O
(46)
4. Calculate the power losses
Power losses in the high-side MOSFET (Si9407AGY) at 55-VIN where switching losses dominate can be
calculated from Equation 46 through Equation 49.
Ǹ
Ǹ
I
+ I d + 5 0.0588 + 1.2 A
O
RMS
(47)
substituting Equation 30 into Equation 29 yields
2
(
(
))
P
+ 1.2 0.12 1 ) 0.007 150 * 25 + 0.324 W
COND
(48)
and from Equation 31, the switching losses can be determined.
21
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
DESIGN EXAMPLE (continued)
+ ǒV
Ǔ
f
P
I t
+ 55 V 5 A 20 ns 130 kHz + 0.715 W
SW
SW(fsw)
IN
O
SW
(49)
(50)
The MOSFET junction temperature can be found by substituting Equation 33 into Equation 32
O
T + ǒP
Ǔ
(
)
) P
q ) T + 0.324 ) 0.715 40 ) 85 + 127 C
J
COND
SW
JA
A
5. Calculate synchronous rectifier losses
The synchronous rectifier MOSFET has two (2) loss components, conduction, and diode reverse recovery
losses. The conduction losses are due to IRMS losses as well as body diode conduction losses during the dead
time associated with the anti-cross conduction delay.
The IRMS current through the synchronous rectifier from Equation 51
Ǹ
Ǹ
I
+ I 1 * d + 5 1 * 0.0588 + 4.85 A
O RMS
RMS
(51)
The synchronous MOSFET conduction loss from Equation 29 is:
2
2
(
(
))
P
+ I
R
RMS DS(on)
+ 4.85 0.011 1 ) 0.007 150 * 25 + 0.10 W
COND
(52)
(53)
(54)
(55)
(56)
The body diode conduction loss from Equation 35 is:
+ 2 I V t f + 2 5.0 A 0.8 V 100 ns 130 kHz + 0.104 W
P
DC
O
FD
DELAY
SW
The body diode reverse recovery loss from Equation 36 is:
+ 0.5 Q V f + 0.5 30 nC 55 V 130 kHz + 0.107 W
P
RR
RR
IN
SW
The total power dissipated in the synchronous rectifier MOSFET from Equation 37 is:
+ P ) P ) P + 0.107 ) 0.1 ) 0.104 + 0.311 W
P
SR
RR
COND
DC
The junction temperature of the synchronous rectifier at 85°C is:
o
(
)
T + P q ) T + 0.311 40 ) 85 + 97 C
J
SR
JA
A
In typical applications, paralleling the synchronous rectifier MOSFET with a Schottky rectifier increases the
overall converter efficiency by approximately 2% due to the lower power dissipation during the body diode
conduction and reverse recovery periods.
6. Calculate the Inductor Value
The inductor value is calculated from Equation 11
(
)
48 * 3.3 3.3
48 2.0 130 kHz
L +
+ 11.8 mH
(57)
A standard inductor value of 10-µH is chosen. A Coev DXM1306-10RO or Panasonic ETQPF102HFA could be
used.
7. Setting the switching frequency
The clock frequency is set with a resistor (RT) from the RT pin to ground. The value of RT can be derived from
following Equation 58, with fSW in kHz.
1
R + ǒ
Ǔ
* 23 kW + 408 kW, use 412 kW
T
f
17.82 E * 06
SW
(58)
8. Programming the Ramp Generator Circuit
The PWM ramp is programmed through a resistor (RKFF) from the KFF pin to VIN. The ramp generator also
controls the input UVLO voltage. For an undervoltage level of 14.4V (20% below the 18 VIN(min), RKFF is
calculated in Equation 59.
22
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
DESIGN EXAMPLE (continued)
* 3.5 ǒ65.27 R ) 1502Ǔ
W + 133.7 kW, use 133 kW
T
+ ǒV
Ǔ
R
KFF
IN(min)
(59)
9. Calculating the Output Capacitance (CO)
In this example. the output capacitance is determined by the load response requirement of ∆V = 0.3 V for a 1 A
to 5 A step load. CO can be calculated using Equation 18.
2
2
Ǔ
ǒ
10 mH 5 * 1
C
+
+ 127 mF
O
2
2
Ǔ
ǒ
3.3 * 3.0
(60)
(61)
Using Equation 13 calculate the ESR required to meet the output ripple requirements.
1
ǒ
Ǔ
33 mV + 2.0 ESR )
8 127 mF 130 kHz
ESR = 12.7 mΩ
In order to get the required ESR, the capacitance needs to be greater than the 127-µF calculated. For example,
a single Panasonic SP capacitor, 180-µF with ESR of 12 mΩ can be used. Re-calculating the ESR required with
the ne3w3 vmaVlu+e o2f.108ǒ0E-SµRF i)s shown in Equation 62. Ǔ
1
8 180 mF 130 kHz
(62)
ESR = 13.8 mΩ
10. Calculate the Soft-Start Capacitor (CSS)
This design requires a soft-start time (tSTART) of 1 ms. CSS is calculated in Equation 63.
2.3 mA
0.7 V
C
+
1 ms + 3.28 nF + 3300 pF
SS
(63)
11. Calculate the Current Limit Resistor (RILIM
)
The current limit set point depends on tSTART, VO, CO and ILOAD at start up as shown in Equation 7.
180 mF 3.3
I
u
) 7.0 + 7.6 A
LIM
1 m
(64)
(65)
Set ILIM for 10.0 A minimum, then from Equation 7
V
(
)
50 mV
8.3 mA
10 0.14
OS
10 0.14
8.3 mA
R
+
)
W +
)
W + 175 kW + 174 kW
ILIM
I
I
SINK
SINK
12. Calculate Loop Compensation Values
Calculate the DC modulator gain (AMOD) from Equation 19.
10
A
+
+ 5
MOD
2
(66)
(67)
( )
+ 20 log 5.0 + 14 dB
A
MOD(dB)
Calculate the output poles and zeros from Equation 21 and Equation 22 of the L-C filter.
1
f
+
+ 3.7 kHz
LC
Ǹ
2p 10 mH 180 mF
(68)
(69)
and
1
f +
+ 74 kHz
Z
2p 0.012 180 mF
Select the close-loop 0 dB crossover frequency, fC. For this example fC = 10 kHz.
23
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
DESIGN EXAMPLE (continued)
Select the double zero location for the Type III compensation network at the output filter double pole at 3.7 kHz.
Select the double pole location for the Type III compensation network at the output capacitor ESR zero at 73.7
kHz.
The amplifier gain at the crossover frequency of 10 kHz is determined by the reciprocal of the modulator gain
AMOD at the crossover frequency from Equation 27.
2
2
f
LC
3.7 kHz
ǒ Ǔ
ǒ Ǔ
A
+ A
MOD
+ 5
+ 0.68
MOD(f)
f
10 kHz
C
(70)
(71)
And also from Equation 27.
1
1
0.68
G +
+
+ 1.46
A
MOD(f)
Choose R1 = 100 kΩ
The poles and zeros for a Type III network are described in Equation 25 and Equation 26.
1
1
f
+
N C3 +
+ 430 pF, choose 470 pF
Z2
2p R1 C3
2p 100 kW 3.7 kHz
(72)
1
1
f
+
N R3 +
+ 4.62 kW, choose 4.64 kW
P2
2p R3 C3
2p 470 pF 73.3 kHz
(73)
(74)
1
1
f
f
+
N C2 +
+ 109 pF, choose 100 pF
C
2p R1 C2 G
2p 100 kW 3.29 1.46 kHz
1
1
+
+
N R2 +
+ 21.7 kW, choose 21.5 kW
P1
2p R2 C2
2p 100 pF 73.3 kHz
(75)
(76)
1
1
f
N C1 +
+ 2000 pF, choose 1800 pF
Z1
2p R2 C1
2p 21.5 kW 3.7 kHz
Calculate the value of RBIAS from Equation 23 with R1 = 100 kΩ.
0.7 V R1
0.7 V 100kW
3.3 V * 0.7 V
R
+
+
+ 26.9 kW, choose 26.7 kW
BIAS
V
* 0.7 V
O
(77)
CALCULATING THE BPN10 AND BP10V BYPASS CAPACITANCE
The size of the bypass capacitor depends on the total gate charge of the MOSFET being used and the amount
of droop allowed on the bypass capacitor. The BPN10 capacitance, allowing for a 0.5-V droop on the BPN10 pin
from Equation 8 is shown in Equation 78.
Q
g
30 nC
0.5
C
+
+
+ 60 nF
BPN10
DV
(78)
and the BP10V capacitance from Equation 9 is shown in Equation 79.
Q
gSR
57 nC
0.5
C
+
+
+ 114 nF
BP10V
DV
(79)
For this application, a 0.1-µF capacitor was used for the BPN10V and a 1.0-µF was used for the BP10V bypass
ccapacitor. Figure 14 shows component selection for the 18-V through 55-V to 3.3-V at 5-A dc-to-dc converter
specified in the design example.
GATE DRIVE CONFIGURATION
Due to the possibility of dv/dt induced turn-on from the fast MOSFET switching times, high VDS voltage and low
gate threshold voltage of the Si4470, the design includes a 2-Ω in the gate lead of the upper MOSFET. The
resistor can be used to shape the low-to-high transition of the SWitch node and reduce the tendancy of
dv/dt-induced turn on.
24
TPS40060
TPS40061
www.ti.com
SLUS543D–DECEMBER 2002–REVISED SEPTEMBER 2004
DESIGN EXAMPLE (continued)
R
R
KFF
ILIM
133 kΩ
174 kΩ
TPS40060PWP
1
2
3
KFF
16
15
+
ILIM
R
T
412 kΩ
RT
VIN
V
2 Ω
IN
BP5
HDRV 14
Si9470
0.1 µF
−
BPN10
13
0.1 µF
SYNC
4
R
SW
30BQ060
10 Ω
10 µH
5
6
7
8
12
SGND
SS/SD
VFB
SW
C
SS
1.0 µF
+
BP10 11
LDRV 10
R1
100kΩ
C
O
Si4470
R3
4.64 kΩ
180 µF
3300 pF
C1 1800 pF
V
OUT
C3
470 pF
COMP
PGND
9
R2
PGND
−
21.5 kΩ
R
BIAS
C2
100 pF
26.7 kΩ
UDG−02161
Figure 14. Design Example, 48 V to 3.3 V at 5 A dc-to-dc Converter
REFERENCES
1. Balogh, Laszlo, Design and Application Guide for High Speed MOSFET Gate Drive Circuits, Texas
Instruments/Unitrode Corporation, Power Supply Design Seminar, SEM-1400 Topic 2.
2. PowerPAD Thermally Enhanced Package Texas Instruments, Semiconductor Group, Technical Brief: TI
Literature No. SLMA002
25
PACKAGE OPTION ADDENDUM
www.ti.com
8-Mar-2005
PACKAGING INFORMATION
Orderable Device
TPS40060PWP
TPS40060PWPR
TPS40061PWP
TPS40061PWPR
Status (1)
ACTIVE
ACTIVE
ACTIVE
ACTIVE
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
Drawing
HTSSOP
PWP
16
16
16
16
90 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR
no Sb/Br)
HTSSOP
HTSSOP
HTSSOP
PWP
PWP
PWP
2000 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR
no Sb/Br)
90 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR
no Sb/Br)
2000 Green (RoHS & CU NIPDAU Level-2-260C-1 YEAR
no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - May not be currently available - please check http://www.ti.com/productcontent for the latest availability information and additional
product content details.
None: Not yet available Lead (Pb-Free).
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens,
including bromine (Br) or antimony (Sb) above 0.1% of total product weight.
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
Addendum-Page 1
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