THS3001IDGNR [TI]
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| 型号： | THS3001IDGNR |
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TEXAS INSTRUMENTS |
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THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
THS3001
THS3002
D AND DGN PACKAGE
(TOP VIEW)
High Speed
†
D AND DGN PACKAGE
– 420 MHz Bandwidth (G = 1, –3 dB)
– 6500 V/µs Slew Rate
– 40-ns Settling Time (0.1%)
(TOP VIEW)
NULL
IN–
NULL
1OUT
1IN–
1IN+
V
CC+
1
2
3
4
8
7
6
5
1
2
3
4
8
7
6
5
V
2OUT
2IN–
2IN+
High Output Drive, I = 100 mA
O
CC+
IN+
OUT
NC
Excellent Video Performance
– 115 MHz Bandwidth (0.1 dB, G = 2)
– 0.01% Differential Gain
V
–V
CC
CC–
NC – No internal connection
– 0.02° Differential Phase
†
The THS3001 implemented in the DGN package is in the
product preview stage of development. Contact your local TI
sales office for availability.
Low 3-mV (max) Input Offset Voltage
Very Low Distortion
– THD = –96 dBc at f = 1 MHz
– THD = –80 dBc at f = 10 MHz
OUTPUT AMPLITUDE
vs
FREQUENCY
Wide Range of Power Supplies
8
– V
= ±4.5 V to ±16 V
CC
V
R
= ±15 V
= 680 Ω
CC
F
Evaluation Module Available
7
6
description
5
The THS300x is a high-speed current-feedback
operational amplifier, ideal for communication,
imaging, and high-quality video applications. This
device offers a very fast 6500-V/µs slew rate, a
420-MHz bandwidth, and 40-ns settling time for
large-signal applications requiring excellent tran-
sient response. In addition, the THS300x
operates with a very low distortion of –96 dBc,
making it well suited for applications such as
wireless communication basestations or ultrafast
ADC or DAC buffers.
V
R
= ±5 V
= 750 Ω
CC
F
4
3
2
1
G = 2
= 150 Ω
0
R
L
V = 200 mV RMS
I
–1
100k
1M
10M
f – Frequency – Hz
100M
1G
HIGH-SPEED AMPLIFIER FAMILY
THD
SUPPLY
VOLTAGE
t
s
ARCHITECTURE
BW
SR
DIFF.
GAIN
DIFF.
PHASE
V
n
(nV/√Hz)
f = 1 MHz 0.1%
(dB)
DEVICE
(MHz)
(V/µs)
(ns)
VFB
CFB
5 V ±5 V ±15 V
THS3001/02
THS4001
•
•
•
•
•
•
•
•
•
•
•
420
270
290
100
180
6500
400
310
100
400
–96
–72
–80
–72
–72
40
40
37
60
40
0.01%
0.04%
0.006%
0.02%
0.02%
0.02°
0.15°
0.01°
0.03°
0.02°
1.6
12.5
7.5
•
•
•
•
•
THS4011/12
THS4031/32
THS4061/62
1.6
14.5
CAUTION: The THS300x provides ESD protection circuitry. However, permanent damage can still occur if this device is subjected
to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance degradation or loss
of functionality.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Copyright 1999, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
1
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
AVAILABLE OPTIONS
PACKAGED DEVICE
EVALUATION
MODULE
T
A
†
MSOP (DGN)
DEVICE SYMBOL
SOIC
(D)
‡
‡
THS3001CD
THS3002CD
THS3001CDGN
THS3002CDGN
TIADP
TIADI
THS3001EVM
THS3002EVM
0°C to 70°C
‡
‡
‡
‡
THS3001ID
THS3002ID
THS3001IDGN
THS3002IDGN
TIADQ
TIADJ
–40°C to 85°C
—
‡
†
‡
The D package is available taped and reeled. Add an R suffix to the device type (i.e.,
THS3001CDR)
Product Preview
†
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)
Supply voltage, V
to V
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 V
CC–
CC+
Input voltage, V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±V
I
CC
Output Current, I
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175 mA
O
Differential input voltage, V
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±6 V
ID
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Dissipation Rating Table
Operating free-air temperature, T , THS300xC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to 70°C
A
THS300xI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to 85°C
Storage temperature, T
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 125°C
stg
Lead temperature, 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300°C
†
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
T
≤ 25°C
DERATING FACTOR
T
= 70°C
T = 85°C
A
POWER RATING
A
A
PACKAGE
POWER RATING
ABOVE T = 25°C
POWER RATING
A
D
740 mW
6 mW/°C
470 mW
380 mW
recommended operating conditions
MIN NOM
MAX
±16
32
UNIT
Split supply
Single supply
THS300xC
±4.5
9
Supply voltage, V
and V
V
CC+
CC–
0
70
Operating free-air temperature, T
°C
A
THS300xI
–40
85
2
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
electrical characteristics, T = 25°C, R = 150 Ω, R = 1 kΩ (unless otherwise noted)
A
L
F
†
PARAMETER
TEST CONDITIONS
MIN
±4.5
9
TYP
MAX
±16.5
33
UNIT
Split supply
Single supply
V
CC
Power supply operating range
Quiescent current
V
T
= 25°C
5.5
6.6
7.5
A
V
V
V
V
= ±5 V
= ±15 V
= ±5 V
= ±15 V
CC
CC
CC
CC
T
A
= full range
= 25°C
8.5
I
mA
CC
T
A
9
T
A
= full range
= 150 Ω
= 1 kΩ
10
R
R
R
R
R
R
±2.9
±3
±3.2
±3.3
L
L
L
L
L
L
V
O
Output voltage swing
V
= 150 Ω
= 1 kΩ
±12.1 ±12.8
±12.8 ±13.1
100
V
V
= ±5 V,
= 20 Ω
CC
I
O
Output current (see Note 1)
mA
= ±15 V,
= 75 Ω
85
120
1
CC
T
= 25°C
3
4
A
V
IO
Input offset voltage
V
= ±5 V or ±15 V
= ±5 V or ±15 V
mV
CC
CC
T
A
= full range
Input offset voltage drift
V
5
2
µV/°C
T
A
= 25°C
10
15
10
15
–Input
T
= full range
= 25°C
A
I
IB
Input bias current
V
CC
= ±5 V or ±15 V
µA
T
A
1
+Input
T
A
= full range
V
V
V
= ±5 V
= ±15 V
= ±5 V,
±3
±3.2
CC
CC
CC
V
ICR
Common-mode input voltage range
Open loop transresistance
V
±12.9 ±13.2
V
= ±2.5 V,
= ±7.5 V,
O
1.3
R
= 1 kΩ
L
MΩ
V
R
= ±15 V,
= 1 kΩ
V
O
CC
2.4
L
V
V
= ±5 V,
V
V
= ±2.5 V
= ±10 V
62
65
65
63
69
67
70
73
76
CC
CM
CMRR Common-mode rejection ratio
dB
= ±15 V,
CC
CM
T
A
= 25°C
V
= ±5 V
dB
dB
CC
CC
T
A
= full range
= 25°C
PSRR
Power supply rejection ratio
Input resistance
T
A
76
V
= ±15 V
T
A
= full range
+Input
–Input
1.5
15
MΩ
Ω
R
I
C
R
Differential input capacitance
Output resistance
7.5
10
pF
Ω
I
Open loop at 5 MHz
O
V
= ±5 V or ±15 V, f = 10 kHz,
CC
G = 2
V
Input voltage noise
Input current noise
1.6
nV/√Hz
pA/√Hz
n
Positive (IN+)
Negative (IN–)
13
16
V
CC
G = 2
= ±5 V or ±15 V, f = 10 kHz,
I
n
†
Full range = 0°C to 70°C for the THS300xC and –40°C to 85°C for the THS300xI.
NOTE 1: Observe power dissipation ratings to keep the junction temperature below absolute maximum when the output is heavily loaded or
shorted. See absolute maximum ratings section.
3
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
operating characteristics, T = 25°C, R = 150 Ω, R = 1 kΩ (unless otherwise noted)
A
L
F
PARAMETER
TEST CONDITIONS
G = –5
= ±5 V,
MIN
TYP
1700
1300
6500
6300
MAX
UNIT
V
V
CC
= 4 V
G = 5
O(PP)
SR
Slew rate (see Note 2)
V/µs
G = –5
G = 5
V
CC
V
= ±15 V,
= 20 V
O(PP)
V
= ±15 V,
Gain = –1,
CC
Settling time to 0.1%
Settling time to 0.1%
Total harmonic distortion
40
25
0 V to 10 V Step
t
s
ns
V
= ±5 V,
Gain = –1,
CC
0 V to 2 V Step,
V
CC
= ±15 V,
V
= 2 V,
O(PP)
G = 2
THD
–80
0.015%
0.01%
0.01°
dBc
f = 10 MHz,
c
G = 2,
V
CC
V
CC
V
CC
V
CC
= ±5 V
40 IRE modulation,
±100 IRE Ramp,
NTSC and PAL
A
D
Differential gain error
Differential phase error
= ±15 V
= ±5 V
G = 2,
40 IRE modulation,
±100 IRE Ramp,
NTSC and PAL
θ
D
= ±15 V
0.02°
V
CC
V
CC
V
CC
V
CC
V
CC
V
CC
V
CC
= ±5 V,
= ±15 V,
= ±5 V
330
420
300
385
350
85
MHz
MHz
G = 1,
R
= 1 kΩ,
F
Small signal bandwidth (–3 dB)
Bandwidth for 0.1 dB flatness
G = 2,
G = 2,
R
R
= 750 Ω,
= 680 Ω,
F
F
BW
= ±15 V
= ±15 V
= ±5 V
MHz
MHz
G = 5, R = 560 Ω,
F
G = 2,
G = 2,
R
R
= 750 Ω,
= 680 Ω,
F
F
= ±15 V
115
V
V
R
= ±5 V,
G = –5
G = 5
65
62
32
MHz
MHz
MHz
CC
O(PP)
= 500 Ω
= 4 V,
L
Full power bandwidth (see Note 3)
Crosstalk (THS3002 only)
V
= ±15 V,
= 20 V
= 500 Ω
G = –5
G = 5
CC
V
O(PP)
31
MHz
dB
R
L
TBD
NOTES: 2. Slew rate is measured from an output level range of 25% to 75%.
3. Full power bandwidth is defined as the frequency at which the output has 3% THD.
PARAMETER MEASUREMENT INFORMATION
R
R
F
G
V
CC
+
–
+
V
O
V
I
50 Ω
R
L
V
CC
–
Figure 1. Test Circuit, Gain = 1 + (R /R )
F
G
4
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
|V
|
Output voltage swing
Current supply
vs Free-air temperature
vs Free-air temperature
vs Free-air temperature
vs Free-air temperature
vs Common-mode input voltage
vs Common-mode input voltage
vs Frequency
2
3
O
I
I
CC
Input bias current
Input offset voltage
4
IB
V
IO
5
6
CMRR Common-mode rejection ratio
7
8
Transresistance
vs Free-air temperature
vs Frequency
9
Closed-loop output impedance
10
V
Voltage noise
Current noise
vs Frequency
11
n
I
n
vs Frequency
11
vs Frequency
12
PSRR
SR
Power supply rejection ratio
vs Free-air temperature
vs Supply voltage
vs Output step peak-to-peak
vs Gain
13
14
Slew rate
15, 16
17
Normalized slew rate
Harmonic distortion
vs Peak-to-peak output voltage swing
vs Frequency
18, 19
20, 21
22, 23
24, 25
26–30
31–34
35, 36
37, 38
39 – 46
Differential gain
vs Loading
Differential phase
vs Loading
Output amplitude
vs Frequency
Normalized output response
Small and large signal frequency response
Small signal pulse response
Large signal pulse response
vs Frequency
5
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
TYPICAL CHARACTERISTICS
OUTPUT VOLTAGE SWING
vs
FREE-AIR TEMPERATURE
CURRENT SUPPLY
vs
FREE-AIR TEMPERATURE
14
9
8
V
= ±15 V
CC
No Load
13.5
13
12.5
12
V
CC
= ±15 V
V
R
= ±15 V
= 150 Ω
CC
L
7
6
5
V
CC
= ±10 V
4
3.5
3
V
= ±5 V
CC
No Load
V
CC
= ±5 V
V
R
= ±5 V
= 150 Ω
CC
4
3
L
2.5
2
–40 –20
0
20
40
60
80
100
–40 –20
0
20
40
60
80
100
T
A
– Free-Air Temperature – °C
T
A
– Free-Air Temperature – °C
Figure 2
Figure 3
INPUT BIAS CURRENT
vs
FREE-AIR TEMPERATURE
INPUT OFFSET VOLTAGE
vs
FREE-AIR TEMPERATURE
–0.5
–1
0
I
IB+
–0.2
V
CC
= ±5 V
V
CC
= ±5 V
I
IB+
–0.4
–0.6
–0.8
V
= ±15 V
CC
–1.5
V
CC
= ±5 V
I
IB–
–2
–2.5
–3
V
CC
= ±15 V
V
= ±15 V
CC
–1
I
IB–
Gain = 1
R
= 1 kΩ
F
–1.2
–40
–20
0
20
40
60
80
100
–40 –20
0
20
40
60
80
100
T
A
– Free-Air Temperature – °C
T
A
– Free-Air Temperature – °C
Figure 4
Figure 5
6
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
TYPICAL CHARACTERISTICS
COMMON-MODE REJECTION RATIO
COMMON-MODE REJECTION RATIO
vs
COMMON-MODE INPUT VOLTAGE
vs
COMMON-MODE INPUT VOLTAGE
80
70
60
80
70
60
50
40
T
= –40°C
A
T
= –40°C
= 85°C
A
T
A
T
A
= 85°C
T
= 25°C
A
T
A
= 25°C
50
40
30
30
20
V
CC
= ±15 V
V
CC
= ±5 V
0
2
4
6
8
10
12
14
0
0.5
1
1.5
2
2.5
3
3.5
4
|V | – Common-Mode Input Voltage – V
IC
|V | – Common-Mode Input Voltage – V
IC
Figure 6
Figure 7
COMMON-MODE REJECTION RATIO
TRANSRESISTANCE
vs
FREE-AIR TEMPERATURE
vs
FREQUENCY
80
70
2.8
2.6
2.4
2.2
2
V
= ±15 V
CC
V
CC
= ±5 V
V
CC
= ±15 V
60
50
40
V
CC
= ±10 V
1.8
30
20
10
0
1 kΩ
1.6
1.4
1.2
1
1 kΩ
–
+
V
O
V
I
1 kΩ
V
= ±5 V
CC
1 kΩ
V
R
= V /2
CC
O
L
= 1 kΩ
1k
10k
100k
1M
10M
100M
–40 –20
0
20
40
60
80
100
f – Frequency – Hz
T
A
– Free-Air Temperature – °C
Figure 8
Figure 9
7
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
TYPICAL CHARACTERISTICS
CLOSED-LOOP OUTPUT IMPEDANCE
VOLTAGE NOISE AND CURRENT NOISE
vs
vs
FREQUENCY
FREQUENCY
100
10
1
1000
100
V
R
= ±15 V
= 750 Ω
CC
F
V
= ±15 V and ±5 V
CC
T = 25°C
A
Gain = +2
= 25°C
T
A
V
= 2 V
I(PP)
I
I
n–
V
O
750 Ω
10
750 Ω
n+
1 kΩ
–
V
0.1
I
+
THS300x
1000
50 Ω
V
O
V
n
Z
=
– 1
o
)
(
V
I
0.01
1
100k
1M
10M
100M
1G
10
100
1k
10k
100k
f – Frequency – Hz
f – Frequency – Hz
Figure 10
Figure 11
POWER SUPPLY REJECTION RATIO
POWER SUPPLY REJECTION RATIO
vs
vs
FREQUENCY
FREE-AIR TEMPERATURE
90
90
85
80
V
= ±5 V
CC
80
70
60
50
40
30
V
CC
= ±15 V
V
CC
= ±15 V
V
CC
= –5 V
V
CC
= ±5 V
–PSRR
V
CC
= –15 V
+PSRR
V
CC
= +5 V
75
70
20
10
0
V
= +15 V
CC
G = 1
R
= 1 kΩ
F
1k
10k
100k
1M
10M
100M
–40 –20
0
20
40
60
80
100
f – Frequency – Hz
T
A
– Free-Air Temperature – °C
Figure 12
Figure 13
8
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
TYPICAL CHARACTERISTICS
SLEW RATE
vs
SUPPLY VOLTAGE
SLEW RATE
vs
OUTPUT STEP
7000
10000
1000
100
G = +5
= 150 Ω
+SR
R
L
t /t = 300 ps
r f
F
6000
5000
4000
R
= 1 kΩ
–SR
3000
2000
1000
+SR
V
= ±15 V
CC
G = +5
= 150 Ω
R
–SR
L
t /t = 300 ps
r f
R
= 1 kΩ
F
5
7
9
11
13
15
0
5
10
15
20
|V | – Supply Voltage – V
CC
V
– Output Step – V
O(PP)
Figure 14
Figure 15
SLEW RATE
vs
OUTPUT STEP
NORMALIZED SLEW RATE
vs
GAIN
2000
1000
1.5
1.4
1.3
1.2
1.1
+SR
V
V
R
R
= ±5 V
= 4 V
= 150 Ω
CC
O(PP)
L
F
= 1 kΩ
t /t = 300 ps
r f
–SR
–Gain
1
0.9
0.8
0.7
+Gain
V
= ±5 V
CC
G = +5
= 150 Ω
R
L
t /t = 300 ps
r f
R = 1 kΩ
F
100
0
1
2
3
4
5
1
2
3
4
5
6
7
8
9
10
V
– Output Step – V
G – Gain – V/V
O(PP)
Figure 16
Figure 17
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TYPICAL CHARACTERISTICS
HARMONIC DISTORTION
vs
PEAK-TO-PEAK OUTPUT VOLTAGE SWING
HARMONIC DISTORTION
vs
PEAK-TO-PEAK OUTPUT VOLTAGE SWING
–50
–55
–60
–50
–55
–60
8 MHz
Gain = 2
4 MHz
Gain = 2
V
R
R
= ±15 V
= 150 Ω
= 750 Ω
CC
L
F
V
R
R
= ±15 V
= 150 Ω
= 750 Ω
CC
L
F
3rd Harmonic
3rd Harmonic
–65
–70
–75
–80
–65
–70
–75
2nd Harmonic
2nd Harmonic
–85
–80
–85
–90
–95
0
2
4
6
8
10 12 14 16 18 20
0
2
4
6
8
10 12 14 16 18 20
V
– Peak-to-Peak Output Voltage Swing – V
O(PP)
V
– Peak-to-Peak Output Voltage Swing – V
O(PP)
Figure 18
Figure 19
HARMONIC DISTORTION
HARMONIC DISTORTION
vs
vs
FREQUENCY
FREQUENCY
–70
–60
Gain = 2
Gain = 2
V
V
R
R
= ±15 V
CC
O
L
F
V
V
R
R
= ±5 V
= 2 V
= 150 Ω
CC
O
L
F
–65
–70
= 2 V
= 150 Ω
PP
–75
–80
–85
–90
PP
= 750 Ω
= 750 Ω
–75
–80
–85
–90
3rd Harmonic
2nd Harmonic
2nd Harmonic
–95
–95
3rd Harmonic
–100
–100
100k
1M
10M
100k
1M
10M
f – Frequency – Hz
f – Frequency – Hz
Figure 20
Figure 21
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TYPICAL CHARACTERISTICS
DIFFERENTIAL GAIN
DIFFERENTIAL GAIN
vs
vs
LOADING
LOADING
0.04
0.04
Gain = 2
= 750 Ω
40 IRE NTSC Modulation
Gain = 2
R = 750 Ω
F
R
F
40 IRE PAL Modulation
Worst Case: ±100 IRE Ramp
Worst Case: ±100 IRE Ramp
0.03
0.02
0.01
0
0.03
0.02
0.01
0
V
= ±15 V
CC
V
= ±15 V
CC
V
= ±5 V
CC
V
= ±5 V
CC
1
2
3
4
5
6
7
8
1
2
3
4
5
6
7
8
Number of 150 Ω Loads
Number of 150 Ω Loads
Figure 22
Figure 23
DIFFERENTIAL PHASE
DIFFERENTIAL PHASE
vs
vs
LOADING
LOADING
0.3
0.35
0.3
Gain = 2
Gain = 2
R = 750 Ω
F
40 IRE PAL Modulation
R
= 750 Ω
F
40 IRE NTSC Modulation
0.25
Worst Case: ±100 IRE Ramp
Worst Case: ±100 IRE Ramp
0.25
0.2
0.2
0.15
V
CC
= ±15 V
0.15
0.1
V
CC
= ±15 V
0.1
0.05
0
V
CC
= ±5 V
V
CC
= ±5 V
0.05
0
1
2
3
4
5
6
7
8
1
2
3
4
5
6
7
8
Number of 150 Ω Loads
Number of 150 Ω Loads
Figure 24
Figure 25
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TYPICAL CHARACTERISTICS
OUTPUT AMPLITUDE
vs
OUTPUT AMPLITUDE
vs
FREQUENCY
FREQUENCY
3
3
Gain = 1
= ±15 V
Gain = 1
V = ±5 V
V
R
= 750 Ω
F
CC
= 150 Ω
R
= 750 Ω
2
1
CC
R = 150 Ω
L
F
2
1
R
L
V = 200 mV RMS
I
V = 200 mV RMS
I
0
0
–1
–2
–3
–4
–1
–2
–3
–4
R
= 1 kΩ
R
= 1 kΩ
F
F
R
= 1.5 kΩ
F
R
= 1.5 kΩ
F
–5
–6
–5
–6
100k
1M
10M
100M
1G
100k
1M
10M
100M
1G
f – Frequency – Hz
f – Frequency – Hz
Figure 26
Figure 27
OUTPUT AMPLITUDE
vs
OUTPUT AMPLITUDE
vs
FREQUENCY
FREQUENCY
9
8
7
6
5
4
3
9
8
7
6
5
4
3
Gain = 2
= ±15 V
Gain = 2
V = ±5 V
R
= 560 Ω
F
V
CC
= 150 Ω
CC
R = 150 Ω
L
R
= 560 Ω
R
F
L
V = 200 mV RMS
I
V = 200 mV RMS
I
R
= 680 Ω
F
R = 750 Ω
F
R
= 1 kΩ
F
R = 1 kΩ
F
2
1
2
1
0
0
–1
100k
–1
100k
1M
10M
100M
1G
1M
10M
100M
1G
f – Frequency – Hz
f – Frequency – Hz
Figure 28
Figure 29
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TYPICAL CHARACTERISTICS
OUTPUT AMPLITUDE
vs
FREQUENCY
70
60
50
40
30
20
10
V
= ±15 V
CC
V
CC
= ±5 V
G = +1000
R
R
= 10 kΩ
= 150 Ω
= 200 mV RMS
F
L
0
V
O
–10
100k
1M
10M
100M
1G
f – Frequency – Hz
Figure 30
NORMALIZED OUTPUT RESPONSE
NORMALIZED OUTPUT RESPONSE
vs
vs
FREQUENCY
FREQUENCY
3
3
Gain = –1
Gain = –1
R
= 560 Ω
V
R
= ±15 V
= 150 Ω
F
CC
L
V
R
= ±5 V
= 150 Ω
2
1
CC
L
2
1
R
= 560 Ω
F
V = 200 mV RMS
I
V = 200 mV RMS
I
0
0
–1
–2
–3
–4
–1
–2
–3
–4
–5
R
= 680 Ω
F
R
= 750 Ω
F
R
= 1 kΩ
F
R = 1 kΩ
F
–5
–6
–6
100k
100k
1M
10M
100M
1G
1M
10M
100M
1G
f – Frequency – Hz
f – Frequency – Hz
Figure 31
Figure 32
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TYPICAL CHARACTERISTICS
NORMALIZED OUTPUT RESPONSE
NORMALIZED OUTPUT RESPONSE
vs
vs
FREQUENCY
FREQUENCY
3
0
4
R = 390 Ω
F
R
= 390 Ω
F
2
0
–3
–6
–9
R = 560 Ω
F
–2
–4
–6
–8
–10
R
= 620 Ω
F
R
= 1 kΩ
F
R
= 1 kΩ
F
Gain = +5
Gain = +5
–12
–15
V
R
V
= ±15 V
= 150 Ω
= 200 mV RMS
CC
L
O
V
R
V
= ±5 V
= 150 Ω
= 200 mV RMS
CC
L
O
–12
–14
100k
1M
10M
100M
1G
100k
1M
10M
100M
1G
f – Frequency – Hz
f – Frequency – Hz
Figure 33
Figure 34
SMALL AND LARGE SIGNAL
FREQUENCY RESPONSE
SMALL AND LARGE SIGNAL
FREQUENCY RESPONSE
–3
3
V = 500 mV
I
V = 500 mV
I
–6
–9
0
–3
V = 250 mV
I
V = 250 mV
I
–12
–15
–18
–21
–24
–6
–9
V = 125 mV
I
V = 125 mV
I
–12
–15
–18
V = 62.5 mV
I
V = 62.5 mV
I
Gain = 1
Gain = 2
V
R
R
= ±15 V
= 1 kΩ
= 150 Ω
V
R
R
= ±15 V
= 680 Ω
= 150 Ω
CC
F
L
CC
F
L
–27
–30
–21
–24
100k
1M
10M
100M
1G
100k
1M
10M
100M
1G
f – Frequency – Hz
f – Frequency – Hz
Figure 35
Figure 36
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TYPICAL CHARACTERISTICS
SMALL SIGNAL PULSE RESPONSE
SMALL SIGNAL PULSE RESPONSE
300
100
60
20
–100
–200
–20
–60
200
100
0
200
100
0
Gain = 1
Gain = 5
V
R
R
= ±5 V
= 150 Ω
= 1 kΩ
V
R
R
= ±5 V
= 150 Ω
= 1 kΩ
CC
L
F
CC
L
F
–100
–100
–200
–300
–200
–300
t /t = 300 ps
r f
t /t = 300 ps
r f
0
10 20 30 40
50 60 70 80
90 100
0
10 20 30 40
50 60 70 80
90 100
t – Time – ns
t – Time – ns
Figure 37
Figure 38
LARGE SIGNAL PULSE RESPONSE
LARGE SIGNAL PULSE RESPONSE
3
1
3
1
–1
–3
–1
–3
2
1
2
1
0
0
Gain = 1
Gain = +1
V
R
R
= ±5 V
= 150 Ω
= 1 kΩ
V
R
R
= ±15 V
= 150 Ω
= 1 kΩ
CC
L
F
CC
L
F
–1
–1
–2
–3
–2
–3
t /t = 2.5 ns
r f
t /t = 2.5 ns
r f
0
10 20 30 40
50 60 70 80
90 100
0
10 20 30 40
50 60 70 80
90 100
t – Time – ns
t – Time – ns
Figure 39
Figure 40
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TYPICAL CHARACTERISTICS
LARGE SIGNAL PULSE RESPONSE
LARGE SIGNAL PULSE RESPONSE
3
1
600
200
–1
–3
10
–200
–600
2
1
5
0
0
Gain = 5
= ±5 V
Gain = +5
V
CC
V
R
R
= ±15 V
= 150 Ω
= 1 kΩ
CC
L
F
–1
–5
R
R
= 150 Ω
= 1 kΩ
L
F
–2
–3
–10
–15
t /t = 300 ps
r f
t /t = 300 ps
r f
0
10 20 30 40
50 60 70 80
90 100
0
10 20 30 40
50 60 70 80
90 100
t – Time – ns
t – Time – ns
Figure 41
Figure 42
LARGE SIGNAL PULSE RESPONSE
LARGE SIGNAL PULSE RESPONSE
3
3
1
1
–1
2
–1
2
1
1
Gain = –1
Gain = –1
0
0
V
R
R
= ±15 V
= 150 Ω
= 1 kΩ
V
R
R
= ±5 V
= 150 Ω
= 1 kΩ
CC
L
F
CC
L
F
–1
–1
t /t = 2.5 ns
r f
t /t = 300 ps
r f
–2
–3
–2
–3
0
10 20 30 40
50 60 70 80
90 100
0
10 20 30 40
50 60 70 80
90 100
t – Time – ns
t – Time – ns
Figure 43
Figure 44
16
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TYPICAL CHARACTERISTICS
LARGE SIGNAL PULSE RESPONSE
LARGE SIGNAL PULSE RESPONSE
600
200
–200
–600
2
3
1
–1
–2
10
1
0
5
0
Gain = –5
Gain = –5
V
R
R
= ±5 V
= 150 Ω
= 1 kΩ
CC
L
F
V
R
R
= ±15 V
= 150 Ω
= 1 kΩ
CC
L
F
–1
–5
–2
–3
–10
–15
t /t = 300 ps
r f
t /t = 300 ps
r f
0
10 20 30 40
50 60 70 80
90 100
0
10 20 30 40
50 60 70 80
90 100
t – Time – ns
t – Time – ns
Figure 45
Figure 46
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APPLICATION INFORMATION
theory of operation
The THS300x is a high-speed, operational amplifier configured in a voltage-feedback architecture. The device
is built using a 30-V, dielectrically isolated, complementary bipolar process with NPN and PNP transistors
possessing f s of several GHz. This configuration implements an exceptionally high-performance amplifier that
T
has a wide bandwidth, high slew rate, fast settling time, and low distortion. A simplified schematic is shown in
Figure 47.
V
CC+
7
I
IB
3
2
IN+
IN–
6
OUT
I
IB
4
V
CC–
Figure 47. Simplified Schematic
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APPLICATION INFORMATION
recommended feedback and gain resistor values
The THS300x is fabricated using Texas Instruments 30-V complementary bipolar process, HVBiCOM. This
process provides the excellent isolation and extremely high slew rates that result in superior distortion
characteristics.
As with all current-feedback amplifiers, the bandwidth of the THS300x is an inversely proportional function of
thevalueofthefeedbackresistor(seeFigures26to34). Therecommendedresistorsfortheoptimumfrequency
response are shown in Table 1. These should be used as a starting point and once optimum values are found,
1% tolerance resistors should be used to maintain frequency response characteristics. For most applications,
a feedback resistor value of 1 kΩ is recommended – a good compromise between bandwidth and phase margin
that yields a very stable amplifier.
Consistent with current-feedback amplifiers, increasing the gain is best accomplished by changing the gain
resistor, not the feedback resistor. This is because the bandwidth of the amplifier is dominated by the feedback
resistor value and internal dominant-pole capacitor. The ability to control the amplifier gain independent of the
bandwidth constitutes a major advantage of current-feedback amplifiers over conventional voltage-feedback
amplifiers. Therefore, once a frequency response is found suitable to a particular application, adjust the value
of the gain resistor to increase or decrease the overall amplifier gain.
Finally, it is important to realize the effects of the feedback resistance on distortion. Increasing the resistance
decreases the loop gain and increases the distortion. It is also important to know that decreasing load
impedance increases total harmonic distortion (THD). Typically, the third-order harmonic distortion increases
more than the second-order harmonic distortion.
Table 1. Recommended Resistor Values for Optimum Frequency Response
GAIN
1
R
for V
= ±15 V
R
F
for V = ±5 V
CC
F
CC
1 kΩ
1 kΩ
750 Ω
620 Ω
620 Ω
2, –1
–2
680 Ω
620 Ω
560 Ω
5
offset voltage
Theoutputoffsetvoltage,(V )isthesumoftheinputoffsetvoltage(V )andbothinputbiascurrents(I )times
OO
IO
IB
the corresponding gains. The following schematic and formula can be used to calculate the output offset
voltage:
R
F
I
IB–
R
G
+
–
+
V
IO
V
O
R
S
I
IB+
R
R
R
R
F
F
V
V
1
I
R
1
I
R
OO
IO
IB
S
IB–
F
G
G
Figure 48. Output Offset Voltage Model
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APPLICATION INFORMATION
noise calculations and noise figure
Noise can cause errors on very small signals. This is especially true for amplifying small signals coming over
a transmission line or an antenna. The noise model for current-feedback amplifiers (CFB) is the same as for
voltage feedback amplifiers (VFB). The only difference between the two is that CFB amplifiers generally specify
different current-noise parameters for each input, while VFB amplifiers usually only specify one noise-current
parameter. The noise model is shown in Figure 49. This model includes all of the noise sources as follows:
•
•
•
•
e = amplifier internal voltage noise (nV/√Hz)
n
IN+ = noninverting current noise (pA/√Hz)
IN– = inverting current noise (pA/√Hz)
e
= thermal voltage noise associated with each resistor (e = 4 kTR )
Rx x
Rx
e
Rs
e
n
R
Noiseless
S
+
_
e
ni
e
no
IN+
IN–
e
Rf
R
F
e
Rg
R
G
Figure 49. Noise Model
The total equivalent input noise density (e ) is calculated by using the following equation:
ni
2
2
2
e
e
IN
R
IN–
R
R
4 kTR
4 kT R
R
n
s
ni
S
F
G
F
G
Where:
–23
k = Boltzmann’s constant = 1.380658 × 10
T = temperature in degrees Kelvin (273 +°C)
R || R = parallel resistance of R and R
F
G
F
G
To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (e ) by the
ni
overall amplifier gain (A ).
V
R
R
F
e
e
A
e
1
(Noninverting Case)
no
ni
ni
V
G
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APPLICATION INFORMATION
noise calculations and noise figure (continued)
As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the
closed-loop gain is increased (by reducing R ), the input noise is reduced considerably because of the parallel
G
resistance term. This leads to the general conclusion that the most dominant noise sources are the source
resistor (R ) and the internal amplifier noise voltage (e ). Because noise is summed in a root-mean-squares
S
n
method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly
simplify the formula and make noise calculations much easier.
This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise
figure is a measure of noise degradation caused by the amplifier. The value of the source resistance must be
defined and is typically 50 Ω in RF applications.
2
e
ni
NF
10log
e
2
Rs
Because the dominant noise components are generally the source resistance and the internal amplifier noise
voltage, we can approximate noise figure as:
2
2
e
IN
R
n
S
NF
10log 1
4 kTR
S
The Figure 50 shows the noise figure graph for the THS300x.
NOISE FIGURE
vs
SOURCE RESISTANCE
20
f = 10 kHz
18
T
A
= 25°C
16
14
12
10
8
6
4
2
0
10
100
1k
10k
R
– Source Resistance – Ω
S
Figure 50. Noise Figure vs Source Resistance
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APPLICATION INFORMATION
slew rate
The slew rate performance of a current-feedback amplifier, like the THS300x, is affected by many different
factors. Some of these factors are external to the device, such as amplifier configuration and PCB parasitics,
and others are internal to the device, such as available currents and node capacitance. Understanding some
of these factors should help the PCB designer arrive at a more optimum circuit with fewer problems.
Whether the THS300x is used in an inverting amplifier configuration or a noninverting configuration can impact
theoutputslewrate. Ascanbeseenfromthespecificationtablesaswellassomeofthefiguresinthisdatasheet,
slew-rate performance in the inverting configuration is faster than in the noninverting configuration. This is
because in the inverting configuration the input terminals of the amplifier are at a virtual ground and do not
significantly change voltage as the input changes. Consequently, the time to charge any capacitance on these
input nodes is less than for the noninverting configuration, where the input nodes actually do change in voltage
an amount equal to the size of the input step. In addition, any PCB parasitic capacitance on the input nodes
degrades the slew rate further simply because there is more capacitance to charge. Also, if the supply voltage
(V )totheamplifierisreduced, slewratedecreasesbecausethereislesscurrentavailablewithintheamplifier
CC
to charge the capacitance on the input nodes as well as other internal nodes.
Internally, the THS300x has other factors that impact the slew rate. The amplifier’s behavior during the slew-rate
transition varies slightly depending upon the rise time of the input. This is because of the way the input stage
handles faster and faster input edges. Slew rates (as measured at the amplifier output) of less than about
1500 V/µs are processed by the input stage in a very linear fashion. Consequently, the output waveform
smoothly transitions between initial and final voltage levels. This is shown in Figure 51. For slew rates greater
than 1500 V/µs, additional slew-enhancing transistors present in the input stage begin to turn on to support
these faster signals. The result is an amplifier with extremely fast slew-rate capabilities. Figures 41 and 52 show
waveforms for these faster slew rates. The additional aberrations present in the output waveform with these
faster-slewing input signals are due to the brief saturation of the internal current mirrors. This phenomenon,
which typically lasts less than 20 ns, is considered normal operation and is not detrimental to the device in any
way. If for any reason this type of response is not desired, then increasing the feedback resistor or slowing down
the input-signal slew rate reduces the effect.
SLEW RATE
SLEW RATE
4
2
4
2
0
0
10
5
–2
5
SR = 2400 V/µs
Gain = 5
SR = 1500 V/µs
Gain = 5
0
0
V
R
R
= ±15 V
= 150 Ω
= 1 kΩ
V
R
R
= ±15 V
= 150 Ω
= 1 kΩ
CC
L
F
CC
L
F
–5
–5
t /t = 5 ns
r f
t /t = 10 ns
r f
–10
–15
–10
–15
0
20 40 60 80 100 120 140 160 180 200
0
20 40 60 80 100 120 140 160 180 200
t – Time – ns
t – Time – ns
Figure 51
Figure 52
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THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
APPLICATION INFORMATION
driving a capacitive load
Driving capacitive loads with high-performance amplifiers is not a problem as long as certain precautions are
taken. The first is to realize that the THS300x has been internally compensated to maximize its bandwidth and
slew-rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the
output will decrease the device’s phase margin leading to high-frequency ringing or oscillations. Therefore, for
capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of
the amplifier, as shown in Figure 53. A minimum value of 20 Ω should work well for most applications. For
example, in 75-Ω transmission systems, setting the series resistor value to 75 Ω both isolates any capacitance
loading and provides the proper line impedance matching at the source end.
1 kΩ
1 kΩ
_
Input
20 Ω
Output
LOAD
THS300x
+
C
Figure 53. Driving a Capacitive Load
PCB design considerations
Proper PCB design techniques in two areas are important to assure proper operation of the THS300x. These
areas are high-speed layout techniques and thermal-management techniques. Because the THS300x is a
high-speed part, the following guidelines are recommended.
Ground plane – It is essential that a ground plane be used on the board to provide all components with a
low inductive ground connection. Although a ground connection directly to a terminal of the THS300x is not
necessarily required, it is recommended that the thermal pad of the package be tied to ground. This serves
two functions: it provides a low inductive ground to the device substrate to minimize internal crosstalk, and
it provides the path for heat removal.
Input stray capacitance – To minimize potential problems with amplifier oscillation, the capacitance at the
inverting input of the amplifiers must be kept to a minimum. To do this, PCB trace runs to the inverting input
must be as short as possible, the ground plane must be removed under any etch runs connected to the
inverting input, and external components should be placed as close as possible to the inverting input. This
isespeciallytrueinthenoninvertingconfiguration. AnexampleofthiscanbeseeninFigure54, whichshows
what happens when a 1-pF capacitor is added to the inverting input terminal. The bandwidth increases at
the expense of peaking. This is because some of the error current is flowing through the stray capacitor
instead of the inverting node of the amplifier. Although, while the device is in the inverting mode, stray
capacitance at the inverting input has a minimal effect. This is because the inverting node is at a virtual
ground and the voltage does not fluctuate nearly as much as in the noninverting configuration. This can be
seen in Figure 55, where a 10-pF capacitor adds only 0.35 dB of peaking. In general, as the gain of the
system increases, the output peaking due to this capacitor decreases. While this can initially look like a
faster and better system, overshoot and ringing are more likely to occur under fast transient conditions. So
proper analysis of adding a capacitor to the inverting input node should be performed for stable operation.
23
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
APPLICATION INFORMATION
PCB design considerations (continued)
OUTPUT AMPLITUDE
vs
OUTPUT AMPLITUDE
vs
FREQUENCY
FREQUENCY
7
1
C = 10 pF
I
1 kΩ
C = 1 pF
I
6
0
–1
–2
–3
–4
C
in
in
V
out
–
+
5
4
3
C = Stray C Only
I
V
R
150 Ω
=
L
C
in
50 Ω
V
in
1 kΩ
50 Ω
1 kΩ
2
1
0
V
out
–
+
R
150 Ω
=
L
–5
–6
–1
–2
C = 0 pF
I
Gain = 1
Gain = –1
= ±15 V
CC
= 200 mV RMS
(Stray C Only)
V
V
= ±15 V
CC
V
–7
–8
–3
–4
= 200 mV RMS
O
V
O
100k
1M
10M 100M
1G
100k
1M
10M
100M
1G
f – Frequency – Hz
f – Frequency – Hz
Figure 54
Figure 55
Proper power-supply decoupling – Use a minimum 6.8-µF tantalum capacitor in parallel with a 0.1-µF
ceramic capacitor on each supply terminal. It may be possible to share the tantalum among several
amplifiers depending on the application, but a 0.1-µF ceramic capacitor should always be used on the
supply terminal of every amplifier. In addition, the 0.1-µF capacitor should be placed as close as possible
tothesupplyterminal. Asthisdistanceincreases, theinductanceintheconnectingetchmakesthecapacitor
less effective. The designer should strive for distances of less than 0.1 inches between the device power
terminal and the ceramic capacitors.
thermal information
The THS300x incorporates output-current-limiting protection. Should the output become shorted to ground, the
output current is automatically limited to the value given in the data sheet. While this protects the output against
excessivecurrent, the device internal power dissipation increases due to the high current and large voltage drop
across the output transistors. Continuous output shorts are not recommended and could damage the device.
Additionally, connection of the amplifier output to one of the supply rails (±V ) is not recommended. Failure
CC
of the device is possible under this condition and should be avoided. But, the THS300x does not incorporate
thermal-shutdown protection. Because of this, special attention must be paid to the device’s power dissipation
or failure may result.
24
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
APPLICATION INFORMATION
thermal information (continued)
The thermal coefficient θ is approximately 169°C/W for the SOIC 8-pin D package. For a given θ , the
JA
JA
maximum power dissipation, shown in Figure 56, is calculated by the following formula:
T
–T
MAX
A
P
D
JA
Where:
P
= Maximum power dissipation of THS300x (watts)
= Absolute maximum junction temperature (150°C)
= Free-ambient air temperature (°C)
D
T
MAX
T
A
θ
= Thermal coefficient from die junction to ambient air (°C/W)
JA
MAXIMUM POWER DISSIPATION
vs
FREE-AIR TEMPERATURE
1.5
SOIC-D Package:
θ
T
= 169°C/W
= 150°C
JA
J
No Airflow
1
0.5
0
–40 –20
0
20
40
60
80
100
T
A
– Free-Air Temperature – °C
Figure 56. Maximum Power Dissipation vs Free-Air Temperature
25
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
APPLICATION INFORMATION
general configurations
A common error for the first-time CFB user is the creation of a unity gain buffer amplifier by shorting the output
directly to the inverting input. A CFB amplifier in this configuration will oscillate and is not recommended. The
THS300x, like all CFB amplifiers, must have a feedback resistor for stable operation. Additionally, placing
capacitors directly from the output to the inverting input is not recommended. This is because, at high
frequencies, a capacitor has a very low impedance. This results in an unstable amplifier and should not be
considered when using a current-feedback amplifier. Because of this, integrators and simple low-pass filters,
which are easily implemented on a VFB amplifier, have to be designed slightly differently. If filtering is required,
simply place an RC-filter at the noninverting terminal of the operational-amplifier (see Figure 57).
R
R
F
G
1
f
–3dB
2 R1C1
V
R
F
–
O
1
1
V
O
V
R
1
sR1C1
I
G
+
V
I
R1
C1
Figure 57. Single-Pole Low-Pass Filter
If a multiple-pole filter is required, the use of a Sallen-Key filter can work very well with CFB amplifiers. This is
because the filtering elements are not in the negative feedback loop and stability is not compromised. Because
oftheirhighslew-ratesandhighbandwidths, CFBamplifierscancreateveryaccuratesignalsandhelpminimize
distortion. An example is shown in Figure 58.
C1
R1 = R2 = R
C1 = C2 = C
Q = Peaking Factor
(Butterworth Q = 0.707)
+
_
V
I
1
R1
R2
f
–3dB
2 RC
C2
R
F
1
R
=
G
R
F
2 –
)
(
R
Q
G
Figure 58. 2-Pole Low-Pass Sallen-Key Filter
There are two simple ways to create an integrator with a CFB amplifier. The first, shown in Figure 59, adds a
resistor in series with the capacitor. This is acceptable because at high frequencies, the resistor is dominant
and the feedback impedance never drops below the resistor value. The second, shown in Figure 60, uses
positive feedback to create the integration. Caution is advised because oscillations can occur due to the positive
feedback.
26
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
APPLICATION INFORMATION
general configurations (continued)
C1
R
F
1
S
R C1
V
R
R
R
G
F
O
F
–
+
V
I
V
S
I
G
V
O
THS300x
Figure 59. Inverting CFB Integrator
R
R
F
G
For Stable Operation:
R
R
R2
F
≥
R1 || R
–
+
G
A
THS300x
V
O
R
R
F
1 +
V
O
V
I
G
)
(
R1
R2
sR1C1
V
I
C1
R
A
Figure 60. Noninverting CFB Integrator
The THS300x may also be employed as a very good video distribution amplifier. One characteristic of
distribution amplifiers is the fact that the differential phase (DP) and the differential gain (DG) are compromised
as the number of lines increases and the closed-loop gain increases (see Figures 22 to 25 for more information).
Be sure to use termination resistors throughout the distribution system to minimize reflections and capacitive
loading.
750 Ω
750 Ω
75-Ω Transmission Line
75 Ω
–
+
V
O1
V
I
THS300x
75 Ω
75 Ω
N Lines
75 Ω
V
ON
75 Ω
Figure 61. Video Distribution Amplifier Application
27
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
APPLICATION INFORMATION
evaluation board
Evaluation boards are available for the THS3001 (literature #SLOP130) and the THS3002 (literature
#SLOP241). The boards have been configured for very low parasitic capacitance in order to realize the full
performance of the amplifier. Schematics of the evaluation boards are shown in Figures 62 and 63. The circuitry
hasbeendesignedsothattheamplifiermaybeusedineitheraninvertingornoninvertingconfiguration. Toorder
the evaluation board contact your local TI sales office or distributor. For more detailed information, refer to the
THS3001EVMUser’sManual(literature#SLOV021)ortheTHS3002EVMUser’sGuide(literature#SLOVxxx).
To order the evaluation board, contact your local TI sales office or distributor.
V
CC
+
+
C1
C2
6.8 µF
0.1 µF
R1
1 kΩ
R2
49.9 Ω
IN+
+
_
R3
49.9 Ω
OUT
THS3001
R5
C3
1 kΩ
6.8 µF
+
C4
0.1 µF
IN–
V
CC
–
R4
49.9 Ω
Figure 62. THS3001 Evaluation Board Schematic
28
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
APPLICATION INFORMATION
evaluation board (continued)
V
CC
+
C5
0.1 µF
C3
R6
301 Ω
R5
R4
100 Ω
THS3002
U1:A
8
R1
2
3
R7
–
+
100 Ω
49.9 Ω
R3
100 Ω
OUT1
1
C4
4
0.1 µF
R2
0 Ω
V
CC
–
C6
R8
R14
301 Ω
R13
R12
100 Ω
THS3002
U1:B
R9
6
R15
–
100 Ω
49.9 Ω
R11
100 Ω
7
OUT2
5
+
R10
0 Ω
Figure 63. THS3002 Evaluation Board Schematic
29
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
MECHANICAL INFORMATION
D (R-PDSO-G**)
PLASTIC SMALL-OUTLINE PACKAGE
14 PIN SHOWN
PINS **
0.050 (1,27)
8
14
16
DIM
0.020 (0,51)
0.014 (0,35)
0.010 (0,25)
0.197
(5,00)
0.344
(8,75)
0.394
(10,00)
M
A MAX
A MIN
14
8
0.189
(4,80)
0.337
(8,55)
0.386
(9,80)
0.244 (6,20)
0.228 (5,80)
0.008 (0,20) NOM
0.157 (4,00)
0.150 (3,81)
Gage Plane
1
7
A
0.010 (0,25)
0°–8°
0.044 (1,12)
0.016 (0,40)
Seating Plane
0.004 (0,10)
0.010 (0,25)
0.004 (0,10)
0.069 (1,75) MAX
4040047/D 10/96
NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15).
D. Falls within JEDEC MS-012
30
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THS3001, THS3002
420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS
SLOS217A – JULY 1998 – REVISED JUNE 1999
MECHANICAL INFORMATION
DGN (S-PDSO-G8)
PowerPAD PLASTIC SMALL-OUTLINE PACKAGE
0,38
0,25
0,65
M
0,25
8
5
Thermal Pad
(See Note D)
0,15 NOM
3,05
2,95
4,98
4,78
Gage Plane
0,25
0°–6°
1
4
0,69
0,41
3,05
2,95
Seating Plane
0,10
0,15
0,05
1,07 MAX
4073271/A 01/98
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Body dimensions include mold flash or protrusions.
D. The package thermal performance may be enhanced by attaching an external heat sink to the thermal pad. This pad is electrically
and thermally connected to the backside of the die and possibly selected leads.
E. Falls within JEDEC MO-187
PowerPAD is a trademark of Texas Instruments Incorporated.
31
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
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TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
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CERTAIN APPLICATIONS USING SEMICONDUCTOR PRODUCTS MAY INVOLVE POTENTIAL RISKS OF
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In order to minimize risks associated with the customer’s applications, adequate design and operating
safeguards must be provided by the customer to minimize inherent or procedural hazards.
TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent
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Copyright 1999, Texas Instruments Incorporated
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