LM2833XSD/NOPB [TI]
1.5MHz/3MHz 3.0A 降压直流/直流开关稳压器 | DSC | 10 | -40 to 125;型号: | LM2833XSD/NOPB |
厂家: | TEXAS INSTRUMENTS |
描述: | 1.5MHz/3MHz 3.0A 降压直流/直流开关稳压器 | DSC | 10 | -40 to 125 开关 信息通信管理 光电二极管 稳压器 |
文件: | 总32页 (文件大小:1399K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LM2833
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LM2833 1.5MHz/3MHz 3.0A Step-Down DC-DC Switching Regulator
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1
FEATURES
DESCRIPTION
The LM2833 regulator is a monolithic, high frequency,
PWM step-down DC/DC converter available in a 10-
pin WSON or MSOP-PowerPAD package. It contains
all the active functions to provide local DC/DC
conversion with fast transient response and accurate
regulation in the smallest possible PCB area. With a
minimum of external components, the LM2833 is
easy to use. The ability to drive 3.0A loads with an
internal 56 mΩ PMOS switch using state-of-the-art
0.5µm BiCMOS technology results in the best power
density available. The world-class control circuitry
allows on-times as low as 30ns, thus supporting
exceptionally high frequency conversion over the
entire 3V to 5.5V input operating range down to the
minimum output voltage of 0.6V. Switching frequency
is internally set to 1.5MHz or 3.0MHz, allowing the
use of extremely small surface mount inductors and
capacitors. Even though the operating frequency is
high, efficiencies up to 93% are easy to achieve.
External shutdown is included, featuring an ultra-low
stand-by current of 300nA. The LM2833 utilizes peak
current-mode control and internal compensation to
provide high-performance regulation over a wide
range of operating conditions. Additional features
include internal soft-start circuitry to reduce inrush
current, cycle-by-cycle current limit, frequency
foldback, thermal shutdown, and output over-voltage
protection.
2
•
•
•
Input Voltage Range of 3.0V to 5.5V
Output Voltage Range of 0.6V to 4.5V
Tiny MSOP-PowerPAD 10 or WSON-10
Package
•
•
3.0A Steady-State Output Current
High Switching Frequencies
–
–
1.5MHz (LM2833X)
3.0MHz (LM2833Z)
•
•
•
Enable Pin
56mΩ PMOS Switch
0.6V, 2% Internal Voltage Reference Over Line
and Temperature
•
•
Internal Soft-Start
Internally Compensated Peak Current-Mode
Control
•
Cycle-by-Cycle Current Limit and Thermal
Shutdown
•
•
•
Frequency Foldback Protection
Input Voltage UVLO (Under-Voltage Lockout)
Output Over-Voltage Protection
APPLICATIONS
•
•
•
•
•
•
Multimedia Set Top Box
Broadband Communications
Core Power in HDDs
Data Acquisition/Telemetry
USB Powered Devices
DSL Modems
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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LM2833
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Typical Application Circuit
L1
V
OUT
7,8
D1
9,10
SW
V
IN
VIND
R1
R3
C3
2
EN
LM2833
5
4
FB
C2
1
C1
VINC
R2
NC
PGND
SGND
3
6
Connection Diagrams
10 VIND
VINC
EN
1
2
3
4
5
1
2
3
4
5
VIND
VIND
SW
10
9
VINC
EN
VIND
9
8
7
6
SGND
SW
DAP
SGND
NC
DAP
8
SW
SW
NC
FB
7
PGND
FB
6
PGND
Figure 1. 10-Pin WSON
See Package DSC
Figure 2. 10-pin MSOP-PowerPAD
See Package DGQ
PIN DESCRIPTIONS
Pin(s)
Name
Description
1
VINC
Input supply for internal bias and control circuitry. Need to locally bypass this pin to GND.
Enable control input. Logic high enables operation. Do not allow this pin to float or subject to
voltages greater than VIN + 0.3V.
2
EN
Signal (analog) ground. Place the bottom resistor of the feedback network as close as possible to
this pin for good load regulation.
3
SGND
4
5
NC
FB
No user function, connect this pin to GND.
Feedback pin. Connect this pin to the external resistor divider to set output voltage.
Power ground pin. Provides ground return path for the internal driver.
Switch pins. Connect these pins to the inductor and catch diode.
Input supply voltage. Connect a bypass capacitor locally from these pins to PGND.
6
PGND
SW
7, 8
9, 10
VIND
Connect to system ground for low thermal impedance, but it cannot be used as a primary GND
connection.
DAP
Die Attach Pad
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings(1)(2)
VINC, VIND
-0.5V to 7V
-0.5V to 3V
-0.5V to 7V
-0.5V to 7V
2kV
FB Voltage
EN Voltage
SW Voltage
ESD Susceptibility(3)
Junction Temperature(4)
Storage Temperature
Soldering Information
150°C
−65°C to +150°C
220°C
Infrared/Convection Reflow (15sec)
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of
device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or
other conditions beyond those indicated in the recommended Operating Ratings is not implied. The recommended Operating Ratings
indicate conditions at which the device is functional and should not be operated beyond such conditions.
(2) If Military/Aerospace specified devices are required, please contact Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) Human body model, 1.5kΩ in series with 100pF.
(4) Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device.
Operating Ratings
VINC, VIND
3V to 5.5V
−40°C to +125°C
Junction Temperature
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Electrical Characteristics
Unless otherwise specified under the Conditions column, VIN = 5V. Limits in standard type are for TJ = 25°C only; limits in
boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are specified
through test, design, or statistical correlation. Typical values represent the most likely parametric norm, and are provided for
reference purposes only.
Symbol
Parameter
Conditions
Min
Typ
Max
0.612
0.616
Units
WSON-10 Package
0.588
0.584
0.600
0.600
VFB
Feedback Voltage
V
MSOP-PowerPAD-10
Package
ΔVFB/(ΔVINxVFB
)
Feedback Voltage Line Regulation
Feedback Input Bias Current
VIN = 3V to 5.5V
0.08
0.1
2.70
2.35
0.35
1.5
3.0
95
%/V
nA
IB
100
VIN Rising
VIN Falling
2.90
Undervoltage Lockout
UVLO Hysteresis
V
V
UVLO
1.85
LM2833X
1.1
2.25
86
1.95
3.75
fSW
Switching Frequency
MHz
LM2833Z
LM2833X
DMAX
Maximum Duty Cycle
Minimum Duty Cycle
%
%
LM2833Z
80
90
LM2833X
5
DMIN
LM2833Z
7
WSON-10 Package
58
90
RDS(ON)
Switch On Resistance
mΩ
MSOP-PowerPAD 10
Package
56
ICL
Switch Current Limit
Enable Threshold Voltage
Shutdown Threshold Voltage
Switch Leakage
3.4
1.8
4.4
A
V
VEN_TH
0.4
ISW
IEN
100
100
3.2
nA
nA
Enable Pin Current
Sink/Source
LM2833X, VFB = 0.55
LM2833Z, VFB = 0.55
All Options VEN = 0V
All Options
5
Quiescent Current (switching)
mA
IQ
4.3
6.5
Quiescent Current (shutdown)
300
0.32
400
800
53
nA
V
VFB_F
fFB
FB Frequency Foldback Threshold
LM2833X, VFB = 0V
LM2833Z, VFB = 0V
WSON-10 Package
Foldback Frequency
kHz
Junction to Ambient
0 LFPM Air Flow(1)
θJA
°C/W
MSOP-PowerPAD-10
Package
50
12
11
WSON-10 Package
θJC
Junction to Case(1)
°C/W
MSOP-PowerPAD-10
Package
Junction Temperature
Rising
TSD
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis
165
15
°C
°C
Junction Temperature
Falling
TSD_HYS
(1) Applies for packages soldered directly onto a 4” x 3” 4-layer standard JEDEC board in still air.
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Typical Performance Characteristics
Unless otherwise specified, VIN = 5V and TA = 25°C.
Efficiency vs Load Current - "LM2833X" and "LM2833Z"
Efficiency vs Load Current - "LM2833X"
Figure 3.
Figure 4.
Efficiency vs Load Current - "LM2833Z"
Oscillator Frequency vs Temperature - "LM2833X"
Figure 5.
Figure 6.
Oscillator Frequency vs Temperature - "LM2833Z"
Current Limit vs Temperature
Figure 7.
Figure 8.
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Typical Performance Characteristics (continued)
Unless otherwise specified, VIN = 5V and TA = 25°C.
RDS(ON) vs Temperature (WSON-10 Package)
RDS(ON) vs Temperature (MSOP-PowerPAD-10 Package)
Figure 9.
Figure 10.
LM2833X IQ (Quiescent Current)
LM2833Z IQ (Quiescent Current)
Figure 11.
Figure 12.
VFB vs Temperature
Frequency Foldback
Figure 13.
Figure 14.
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Typical Performance Characteristics (continued)
Unless otherwise specified, VIN = 5V and TA = 25°C.
Loop Gain and Phase - "LM2833X"
Loop Gain and Phase - "LM2833Z"
Figure 15.
Figure 16.
Load Step Response - "LM2833X"
Line Transient Response - "LM2833X"
Figure 17.
Figure 18.
Startup by EN - "LM2833X"
Shutdown by EN - "LM2833X"
Figure 19.
Figure 20.
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Typical Performance Characteristics (continued)
Unless otherwise specified, VIN = 5V and TA = 25°C.
Startup with EN tied to VIN - "LM2833X"
Short-circuit Triggering - "LM2833X"
Figure 21.
Figure 22.
Recovery from Thermal Shutdown - "LM2833X"
Short-circuit Release - "LM2833X"
Figure 23.
Figure 24.
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Block Diagram
Figure 25. Simplified Block Diagram
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APPLICATION INFORMATION
THEORY OF OPERATION
The LM2833 is a constant frequency PWM buck regulator IC that delivers a 3.0A load current. The regulator is
available in preset switching frequencies of 1.5MHz or 3.0MHz. This high frequency allows the LM2833 to
operate with small surface mount capacitors and inductors, resulting in a DC/DC converter that requires a
minimum amount of board space. The LM2833 is internally compensated, therefore it is simple to use and
requires few external components. The LM2833 uses peak current-mode control to regulate the output voltage.
The following description of operation of the LM2833 will refer to the Typical Application Circuit, to the waveforms
in Figure 26 and simplified block diagram in Figure 25. The LM2833 supplies a regulated output voltage by
switching the internal PMOS power switch at a constant frequency and variable duty cycle. A switching cycle
begins at the falling edge of the reset pulse generated by the internal oscillator. When this pulse goes low, the
output control logic turns on the internal PMOS power switch. During this on-time, the SW pin voltage (VSW
)
swings up to approximately VIN, and the inductor current (IL) increases with a linear slope. IL is measured by the
current sense amplifier, which generates an output proportional to the switch current. The sense signal is
summed with the regulator’s corrective ramp and compared to the error amplifier’s output, which is proportional
to the difference between the feedback voltage and VREF. When the PWM comparator output goes high, the
internal power switch turns off until the next switching cycle begins. During the switch off-time, the inductor
current discharges through the catch diode D1, which forces the SW pin to swing below ground by the forward
voltage (VD) of the catch diode. The regulator loop adjusts the duty cycle (D) to maintain a constant output
voltage.
V
SW
D = T /T
ON SW
V
IN
SW
Voltage
T
OFF
T
ON
0
D
t
-V
T
SW
I
L
I
I
LPK
OUT
Di
L
Inductor
Current
0
t
Figure 26. SW Pin Voltage and Inductor Current Waveforms
SOFT-START/SHUTDOWN
The LM2833 has both enable and shutdown modes that are controlled by the EN pin. Connecting a voltage
source greater than 1.8V to the EN pin enables the operation of the LM2833, while reducing this voltage below
0.4V places the part in a low quiescent current (300nA typical) shutdown mode. There is no internal pull-up on
EN pin, therefore an external signal is required to initiate switching. Do not allow this pin to float or rise to 0.3V
above VIN. It should be noted that when the EN pin voltage rises above 1.8V while the input voltage is greater
than UVLO, there is 15µs delay before switching starts. During this delay the LM2833 will go through a power on
reset state after which the internal soft-start process commences. During soft-start, the error amplifier’s reference
voltage ramps from 0V to its nominal value of 0.6V in approximately 600µs. This forces the regulator output to
ramp up in a controlled fashion, which helps reduce inrush current seen at the input and minimizes output
voltage overshoot.
The simplest way to enable the operation of the LM2833 is to connect the EN pin to VIN which allows self start-up
of the LM2833 whenever the input voltage is applied. However, when an input voltage of slow rise time is used to
power the application and if both the input voltage and the output voltage are not fully established before the soft-
start time elapses, the control circuit will command maximum duty cycle operation of the internal power switch to
bring up the output voltage rapidly. When the feedback pin voltage exceeds 0.6V, the duty cycle will have to
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reduce from the maximum value accordingly, to maintain regulation. It takes a finite amount of time for this
reduction of duty cycle and this can result in a transient in output voltage for a short duration, as shown in
Figure 27. In applications where this output voltage overshoot is undesirable, one simple solution is to add a
feed-forward capacitor (CFF) across the top feedback resistor R1 to speed Gm Amplifier recovery. In practice, a
27nF to 100nF ceramic capacitor is usually a good choice to remove the overshoot completely or limit the
overshoot to an insignificant level during startup, as shown in Figure 28. Another more effective solution is to
control EN pin voltage by a separate logic signal, and pull the signal high only after VIN is fully established. In this
way, the chip can execute a normal, complete soft start process, minimizing any output voltage overshoot. Under
some circumstances at cold temperature, this approach may also be required to minimize any unwanted output
voltage transients that may occur when the input voltage rises slowly. For a fast rising input voltage (100µs for
example), there is no need to control EN separately or add a feed-forward capacitor since the soft-start can bring
up output voltage smoothly as shown in Figure 29.
During startup, the LM2833 gradually increases the switching frequency from 400kHz (LM2833X) or 800kHz
(LM2833Z) to the nominal fixed value, as the feedback voltage increases (see FREQUENCY FOLDBACK section
for more information). Since the internal corrective ramp signal adjusts its slope dynamically, and is proportional
to the switching frequency during startup, a larger output capacitance may be required to insure a smooth output
voltage rise, at low programmed output voltage and high output load current.
Figure 27. Startup Response to VIN
Figure 28. Startup Response to VIN with CFF
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Figure 29. Startup Response to VIN with 100µs rise time
FREQUENCY FOLDBACK
The LM2833 uses frequency foldback to help limit switch current and power dissipation during start-up, short-
circuit and over load conditions by sensing if the feedback voltage is below 0.32V (typical). The LM2833 will
reduce the switching frequency from the nominal fixed value (1.5MHz or 3.0MHz) down to 400kHz (LM2833X) or
800kHz (LM2833Z) when the feedback voltage drops to 0V. See Figure 14 in the Typical Performance
Characteristics section.
LOAD STEP RESPONSE
The LM2833 has a fixed internal loop compensation, which results in a small-signal loop bandwidth highly related
to the output voltage level. In general, the loop bandwidth at low voltage is larger than at high voltage due to the
increased overall loop gain. The limited bandwidth at high output voltage may pose a challenge when loop step
response is concerned. In this case, one effective approach to improving loop step response is to add a feed-
forward capacitor (CFF) in the range of 27nF to 100nF in parallel with the upper feedback resistor (assuming the
lower feedback resistor is 2kΩ), as shown in Figure 30. The feed-forward capacitor introduces a zero-pole pair
which helps compensate the loop. The position of the zero-pole pair is a function of the feedback resistors and
capacitor:
(1)
(2)
Note the factor in parenthesis is the ratio of the output voltage to the feedback voltage. As the output voltage
gets close to 0.6V, the pole moves towards the zero, tending to cancel it out. Consequently, adding CFF will have
less effect on the step response at lower output voltages.
As an example, Figure 32 shows that at the output voltage of 3.3V, a 47nF of CFF can boost the loop bandwidth
to 117kHz, from the original 23kHz as shown in Figure 31. Correspondingly, the responses to a load step
between 0.3A and 3A without and with CFF are shown in Figure 33 and Figure 34 respectively. The higher loop
bandwidth as a result of CFF reduces the total output excursion by more than half.
Aside from the above approach, increasing the output capacitance is generally also effective to reduce the
excursion in output voltage caused by a load step. This approach remains valid for applications where the
desired output voltages are close to the feedback voltage.
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Figure 30. Adding a CFF Capacitor
Figure 31. Loop Gain and Phase without CFF
Figure 32. Loop Gain and Phase with CFF
Figure 33. Load Step Response without CFF
Figure 34. Load Step Response with CFF
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OUTPUT OVER-VOLTAGE PROTECTION
The LM2833 has a built in output over-voltage comparator that compares the FB pin voltage to a threshold
voltage that is 15% higher than the internal reference VREF. Once the FB pin voltage exceeds this threshold level
(typically 0.69V), the internal PMOS power switch is turned off, which allows the output voltage to decrease
towards regulation.
UNDER-VOLTAGE LOCKOUT
Under-voltage lockout (UVLO) prevents the LM2833 from operating until the input voltage exceeds 2.70V
(typical). The UVLO threshold has approximately 350mV of hysteresis, so the part will operate until VIN drops
below 2.35V (typical). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic.
CURRENT LIMIT
The LM2833 uses cycle-by-cycle current limiting to protect the internal power switch. During each switching
cycle, a current limit comparator detects if the power switch current exceeds 4.4A (typical), and turns off the
switch until the next switching cycle begins.
THERMAL SHUTDOWN
Thermal shutdown limits total power dissipation by turning off the internal power switch when the IC junction
temperature typically exceeds 165°C. After thermal shutdown occurs, the power switch does not turn on again
until the junction temperature drops below approximately 150°C.
Design Guide
INDUCTOR SELECTION
The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VOUT) to input voltage (VIN):
VOUT
D =
VIN
(3)
The catch diode (D1) forward voltage drop and the voltage drop across the internal PMOS must be included to
calculate a more accurate duty cycle. Calculate D by using the following formula:
VOUT + VD
D =
VIN + VD - VSW
(4)
VSW can be approximated by:
VSW = IOUT x RDS(ON)
where
•
IOUT is output load current.
(5)
The diode forward drop (VD) can range from 0.3V to 0.7V depending on the quality of the diode. The lower the
VD, the higher the operating efficiency of the converter.
The inductor value determines the output ripple current (ΔiL, as defined in Figure 26). Lower inductor values
decrease the size of the inductor, but increase the output ripple current. An increase in the inductor value will
decrease the output ripple current. In general, the ratio of ripple current to the output current is optimized when it
is set between 0.2 and 0.4 for output currents above 2A. This ratio r is defined as:
DiL
r =
lOUT
(6)
One must ensure that the minimum current limit (3.4A) is not exceeded, so the peak current in the inductor must
be calculated. The peak current (ILPK) in the inductor is calculated by:
ILPK = IOUT + ΔiL/2
(7)
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When the designed maximum output current is reduced, the ratio r can be increased. At a current of 0.1A, r can
be made as high as 0.9. The ripple ratio can be increased at lighter loads because the net ripple is actually quite
low, and if r remains constant the inductor value can be made quite large. An equation empirically developed for
the maximum ripple ratio at any current below 2A is:
-0.3667
r = 0.387 x IOUT
(8)
Note that this is just a guideline, and it needs to be combined with two important factors for proper selection of
inductance values at any operating condition. The first consideration is at output voltage above 2.5V, one needs
to ensure that the inductance given by the above guideline should not be less than 1µH for the LM2833X or
0.5µH for the LM2833Z. Since the LM2833 has a fixed internal corrective ramp signal, a very low inductance
value at high output voltage will generate a very steep down slope of inductor current, which will result in an
insufficient slope compensation, and cause instability known as sub-harmonic oscillation. Another consideration
is at low load current, one needs to ensure that the inductance value given by the guideline should not exceed
10µH for the LM2833X and 4.7µH for the LM2833Z, since too much inductance effectively flattens the down
slope of the inductor current, and may significantly limit the system bandwidth and phase margin resulting in
instability.
The LM2833 operates at frequencies allowing the use of ceramic output capacitors without compromising
transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing output ripple.
See the OUTPUT CAPACITOR section for more details on calculating output voltage ripple.
Now that the ripple current is determined, the inductance is calculated by:
VOUT + VD
x (1-D)
L =
IOUT x r x fSW
where
•
fSW is the switching frequency.
(9)
When selecting an inductor, make sure that it is capable of supporting the peak output current without saturating.
Inductor saturation will result in a sudden reduction in inductance and prevent the regulator from operating
properly. Because of the operating frequency of the LM2833, ferrite based inductors are preferred to minimize
core losses. This presents little restriction since the variety and availability of ferrite-based inductors is large.
Lastly, inductors with lower series resistance (DCR) will provide better operating efficiency. For recommended
inductor selection, refer to Design Examples.
INPUT CAPACITOR
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage rating, RMS current rating, and ESL
(Equivalent Series Inductance). The input voltage rating is specifically stated by the capacitor manufacturer.
Make sure to check any recommended deratings and also verify if there is any significant change in capacitance
at the operating input voltage and the operating temperature. The input capacitor maximum RMS input current
rating (IRMS-IN) must be greater than:
r2
IRMS-IN = IOUT
x
D x
1 - D +
12
Neglecting inductor ripple simplifies the above equation to:
IRMS-IN = IOUT
(10)
x
D x
1 - D
(11)
It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always
calculate the RMS at the point where the duty cycle D is closest to 0.5. The ESL of an input capacitor is usually
determined by the effective cross sectional area of the current path. As a rule of thumb, a large leaded capacitor
will have high ESL and a 1206 ceramic chip capacitor will have very low ESL. At the operating frequencies of the
LM2833, leaded capacitors may have an ESL so large that the resulting impedance (2πfL) will be higher than
that required to provide stable operation. It is strongly recommended to use ceramic capacitors due to their low
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ESR and low ESL. A 22µF multilayer ceramic capacitor (MLCC) is a good choice for most applications. In cases
where large capacitance is required, use surface mount capacitors such as Tantalum capacitors and place at
least a 4.7µF ceramic capacitor close to the VIN pin. For MLCCs it is recommended to use X7R or X5R
dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance varies over operating
conditions.
OUTPUT CAPACITOR
The output capacitor is selected based upon the desired output ripple and transient response. The initial current
of a load transient is provided mainly by the output capacitor. The output ripple of the converter is:
1
RESR
+
DVOUT = DIL
8 x fSW x COUT
(12)
When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the
output ripple will be approximately sinusoidal and 90° phase shifted from the switching action. Given the
availability and quality of MLCCs and the expected output voltage of designs using the LM2833, there is really no
need to review any other capacitor technologies. Another benefit of ceramic capacitors is their ability to bypass
high frequency noise. A certain amount of switching edge noise will couple through parasitic capacitances in the
inductor to the output. A ceramic capacitor will bypass this noise while a tantalum will not. Since the output
capacitor is one of the two external components that control the stability of the regulator control loop, most
applications will require a minimum of 22µF output capacitance. In the case of low output voltage, a larger output
capacitance is required to ensure sufficient phase margin. Capacitance can often, but not always, be increased
significantly with little detriment to the regulator stability. Like the input capacitor, recommended multilayer
ceramic capacitors are X7R or X5R types. Again, verify actual capacitance at the desired operating voltage and
temperature. Check the RMS current rating of the capacitor. The maximum RMS current rating of the capacitor
is:
(13)
One may select a 1206 size MLCC for output capacitor, since its current rating is typically above 1A, more than
enough for the requirement.
CATCH DIODE
The catch diode conducts during the switch off-time. A Schottky diode is recommended for its fast switching time
and low forward voltage drop. The catch diode should be chosen such that its current rating is greater than:
ID = IOUT x (1-D)
(14)
The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.
To improve efficiency, choose a Schottky diode with a low forward voltage drop.
OUTPUT VOLTAGE
The output voltage is set using the following equation where R2 is connected between the FB pin and GND, and
R1 is connected between VOUT and the FB pin. A good value for R2 is 2kΩ.
VOUT
x R2
- 1
R1 =
VREF
VREF = 0.60V
(15)
(16)
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EFFICIENCY ESTIMATION
The complete LM2833 DC/DC converter efficiency can be calculated in the following manner:
POUT
h =
PIN
(17)
(18)
Or
POUT
h =
POUT + PLOSS
Calculations for determining the most significant power losses are shown below. Other losses totaling less than
2% are not discussed.
The main power loss (PLOSS) in the converter includes two basic types of losses: switching loss and conduction
loss. In addition, there is loss associated with the power required for the internal circuitry of IC. Conduction
losses usually dominate at higher output loads, whereas switching losses dominate at lower output loads. The
first step in determining the losses is to calculate the duty cycle (D):
VOUT + VD
D =
VIN + VD - VSW
(19)
VSW is the voltage drop across the internal power switch when it is on, and is equal to:
VSW = IOUT x RDS(ON)
(20)
VD is the forward voltage drop across the catch diode. It can be obtained from the diode manufactures Electrical
Characteristics section. If the DC voltage drop across the inductor (VDCR) is accounted for, the equation
becomes:
VOUT + VD + VDCR
D =
VIN + VD - VSW
(21)
The conduction losses in the catch diode are calculated as follows:
PDIODE = VD x IOUT x (1-D)
(22)
Often this is the single most significant power loss in the circuit. Care should be taken to choose a Schottky
diode with a low forward voltage drop.
Another significant external power loss is the conduction loss in the output inductor. The equation can be
simplified to:
PIND = IOUT2 x RDCR
(23)
The LM2833 conduction loss is mainly associated with the internal power switch:
2
DiL
1
3
PCOND = (IOUT2 x D) x
x
x RDS (ON)
1 +
IOUT
(24)
(25)
If the inductor ripple current is fairly small, the conduction losses can be simplified to:
PCOND = IOUT2 x RDS(ON) x D
Switching losses are also associated with the internal power switch. They occur during the switch on and off
transition periods, where voltages and currents overlap resulting in power loss. The simplest means to determine
this loss is to empirically measuring the rise and fall times (10% to 90%) of the switch at the switch node.
Switching Power Loss is calculated as follows:
PSWR = 0.5 x (VIN x IOUT x fSW x TRISE
)
(26)
(27)
(28)
PSWF = 0.5 x (VIN x IOUT x fSW x TFALL
)
PSW = PSWR + PSWF
The power loss required for operation of the internal circuitry is given by:
PQ = IQ x VIN
(29)
17
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IQ is the quiescent operating current, and is typically around 3.2mA for the LM2833X, and 4.3mA for the
LM2833Z.
An example of efficiency calculation for a typical application is shown in Table 1:
Table 1. Power Loss Tabulation
Conditions
Power loss
VIN
5V
3.3V
VOUT
IOUT
3.0A
POUT
PDIODE
PCOND
9.9W
VD
0.33V
56mΩ
1.5MHz
10ns
277mW
363mW
RDS(ON)
fSW
TRISE
PSW
225mW
TFALL
10ns
INDDCR
28mΩ
3.2mA
89.7%
PIND
PQ
252mW
16mW
IQ
η
D is calculated to be 0.72
PLOSS = Σ ( PCOND + PSW + PQ + PIND + PDIODE
)
(30)
(31)
PLOSS = 1.133W
PCB LAYOUT CONSIDERATIONS
When planning layout there are a few things to consider to achieve a clean, regulated output. The most important
consideration is the close coupling of the GND connections of the input capacitor C1 and the catch diode D1.
These ground ends should be close to one another and be connected to the GND plane with at least two
through-holes. Place these components as close to the IC as possible. The next consideration is the location of
the GND connection of the output capacitor C2, which should be near the GND connections of C1 and D1. There
should be a continuous ground plane on the bottom layer of a two-layer board except under the switching node
island. The signal ground SGND (pin 3) and power ground PGND (pin 6) should be tied together and connected
to ground plane through vias.
The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup
that causes inaccurate regulation. The feedback resistors should be placed as close as possible to the IC, with
the GND of R2 placed as close as possible to the SGND of the IC. The VOUT trace to R1 should be routed away
from the inductor and any other traces that are switching.
High AC currents flow through the VIN, SW and VOUT traces, so they should be as short and wide as possible.
Radiated noise can be decreased by choosing a shielded inductor.
The remaining components should also be placed as close as possible to the IC. Please see Application Note
AN-1229 SNVA054 for further considerations and the LM2833 demo board as an example of a four-layer layout.
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LM2833X Design Example 1
L1
V
OUT
7,8
V
IN
9,10
SW
VIND
D1
R1
R2
R3
C3
2
1
EN
LM2833X
5
4
C2
FB
C1
VINC
NC
PGND
6
SGND
3
Figure 35. LM2833X (1.5MHz): VIN = 3.3V, Output = 1.2V/3.0A
Table 2. Bill of Materials
Part ID
Part Value
3.0A Buck Regulator
22µF, 6.3V, X5R
47µF, 6.3V, X5R
0.22µF, 10V, X7R
Schottky, 0.33V at 3A, VR=30V
1.8µH, 3.6A
Manufacturer
TI
Part Number
LM2833X
U1
C1, Input Cap
TDK
C3216X5R0J226M
C3216X5R0J476M
GRM216R71A224KC01D
CMS01
C2, Output Cap
TDK
C3, Bypass Cap
Murata
Toshiba
TDK
D1, Catch Diode
L1
R1
R2
R3
LTF5022T-1R8N3R6
CRCW08052K00FKEA
CRCW08052K00FKEA
CRCW080510R0FKEA
2.0kΩ, 1%
Vishay
Vishay
Vishay
2.0kΩ, 1%
10Ω, 1%
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LM2833X Design Example 2
Figure 36. LM2833X (1.5MHz): VIN = 5V, Output = 3.3V/3.0A
Table 3. Bill of Materials
Part ID
Part Value
Manufacturer
TI
Part Number
U1
3.0A Buck Regulator
22µF, 6.3V, X5R
47µF, 6.3V, X5R
0.22µF, 10V, X7R
47nF, 10V, X7R
Schottky, 0.43V at 3A, VR=30V
1.2µH, 4.2A
LM2833X
C1, Input Cap
TDK
C3216X5R0J226M
C3216X5R0J476M
GRM216R71A224KC01D
0805ZC473JAZ2A
C2, Output Cap
TDK
C3, Bypass Cap
Murata
AVX
CFF, Feed-forward Cap
D1, Catch Diode
Vishay
TDK
SSA33L-E3/61T
L1
R1
R2
R3
LTF5022T-1R2N4R2
CRCW080510K2FKEA
CRCW08052K26FKEA
CRCW080510R0FKEA
10.2kΩ, 1%
Vishay
Vishay
Vishay
2.26kΩ, 1%
10Ω, 1%
20
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LM2833Z Design Example 3
L1
V
OUT
7,8
V
IN
9,10
SW
VIND
D1
R1
R2
R3
C3
2
1
EN
LM2833Z
5
4
C2
FB
C1
VINC
NC
PGND
6
SGND
3
Figure 37. LM2833Z (3MHz): VIN = 3.3V, Output = 1.2V/3.0A
Table 4. Bill of Materials
Part ID
Part Value
3.0A Buck Regulator
22µF, 6.3V, X5R
47µF, 6.3V, X5R
0.22µF, 10V, X7R
Schottky, 0.33V at 3A, VR=30V
1.0µH, 4.0A
Manufacturer
TI
Part Number
LM2833Z
U1
C1, Input Cap
TDK
C3216X5R0J226M
C3216X5R0J476M
GRM216R71A224KC01D
CMS01
C2, Output Cap
TDK
C3, Bypass Cap
Murata
Toshiba
Taiyo Yuden
Vishay
Vishay
Vishay
D1, Catch Diode
L1
R1
R2
R3
NP04SZB1R0N
2.0kΩ, 1%
CRCW08052K00FKEA
CRCW08052K00FKEA
CRCW080510R0FKEA
2.0kΩ, 1%
10Ω, 1%
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LM2833Z Design Example 4
Figure 38. LM2833Z (3MHz): VIN = 5V, Output = 3.3V/3.0A
Table 5. Bill of Materials
Part ID
Part Value
Manufacturer
TI
Part Number
U1
3.0A Buck Regulator
22µF, 6.3V, X5R
47µF, 6.3V, X5R
0.22µF, 10V, X7R
47nF, 10V, X7R
Schottky, 0.43V at 3A, VR=30V
1.0µH, 4.0A
LM2833Z
C1, Input Cap
TDK
C3216X5R0J226M
C3216X5R0J476M
GRM216R71A224KC01D
0805ZC473JAZ2A
SSA33L-E3/61T
C2, Output Cap
TDK
C3, Bypass Cap
Murata
AVX
CFF, Feed-forward Cap
D1, Catch Diode
Vishay
Taiyo Yuden
Vishay
Vishay
Vishay
L1
R1
R2
R3
NP04SZB1R0N
10.2kΩ, 1%
CRCW080510K2FKEA
CRCW08052K26FKEA
CRCW080510R0FKEA
2.26kΩ, 1%
10Ω, 1%
22
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SNVS505E –MAY 2008–REVISED APRIL 2013
REVISION HISTORY
Changes from Revision D (April 2013) to Revision E
Page
•
Changed layout of National Data Sheet to TI format .......................................................................................................... 22
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status Package Type Package Pins Package
Eco Plan
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
Samples
Drawing
Qty
(1)
(2)
(3)
(4/5)
(6)
LM2833XMY/NOPB
LM2833XSD/NOPB
LM2833ZMY/NOPB
LM2833ZMYX/NOPB
LM2833ZSD/NOPB
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
HVSSOP
WSON
DGQ
DSC
DGQ
DGQ
DSC
10
10
10
10
10
1000 RoHS & Green
1000 RoHS & Green
1000 RoHS & Green
3500 RoHS & Green
1000 RoHS & Green
SN
Level-3-260C-168 HR
Level-1-260C-UNLIM
Level-3-260C-168 HR
Level-3-260C-168 HR
Level-1-260C-UNLIM
-40 to 125
-40 to 125
-40 to 125
-40 to 125
-40 to 125
SPYB
SN
SN
SN
SN
2833X
SPZB
SPZB
2833Z
HVSSOP
HVSSOP
WSON
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
9-Aug-2022
TAPE AND REEL INFORMATION
REEL DIMENSIONS
TAPE DIMENSIONS
K0
P1
W
B0
Reel
Diameter
Cavity
A0
A0 Dimension designed to accommodate the component width
B0 Dimension designed to accommodate the component length
K0 Dimension designed to accommodate the component thickness
Overall width of the carrier tape
W
P1 Pitch between successive cavity centers
Reel Width (W1)
QUADRANT ASSIGNMENTS FOR PIN 1 ORIENTATION IN TAPE
Sprocket Holes
Q1 Q2
Q3 Q4
Q1 Q2
Q3 Q4
User Direction of Feed
Pocket Quadrants
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
LM2833XMY/NOPB
LM2833XSD/NOPB
LM2833ZMY/NOPB
HVSSOP DGQ
WSON DSC
HVSSOP DGQ
10
10
10
10
10
1000
1000
1000
3500
1000
178.0
178.0
178.0
330.0
178.0
12.4
12.4
12.4
12.4
12.4
5.3
3.3
5.3
5.3
3.3
3.4
3.3
3.4
3.4
3.3
1.4
1.0
1.4
1.4
1.0
8.0
8.0
8.0
8.0
8.0
12.0
12.0
12.0
12.0
12.0
Q1
Q1
Q1
Q1
Q1
LM2833ZMYX/NOPB HVSSOP DGQ
LM2833ZSD/NOPB WSON DSC
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
9-Aug-2022
TAPE AND REEL BOX DIMENSIONS
Width (mm)
H
W
L
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
LM2833XMY/NOPB
LM2833XSD/NOPB
LM2833ZMY/NOPB
LM2833ZMYX/NOPB
LM2833ZSD/NOPB
HVSSOP
WSON
DGQ
DSC
DGQ
DGQ
DSC
10
10
10
10
10
1000
1000
1000
3500
1000
208.0
210.0
208.0
356.0
210.0
191.0
185.0
191.0
356.0
185.0
35.0
35.0
35.0
35.0
35.0
HVSSOP
HVSSOP
WSON
Pack Materials-Page 2
PACKAGE OUTLINE
DGQ0010A
PowerPADTM - 1.1 mm max height
S
C
A
L
E
3
.
7
0
0
PLASTIC SMALL OUTLINE
C
5.05
4.75
TYP
SEATING PLANE
PIN 1 ID
AREA
A
0.1 C
8X 0.5
10
1
3.1
2.9
NOTE 3
2X
2
5
6
0.27
0.17
10X
3.1
2.9
1.1 MAX
0.08
C A B
B
NOTE 4
0.23
0.13
TYP
SEE DETAIL A
EXPOSED
THERMAL PAD
4
5
0.25
GAGE PLANE
2.05
1.75
0.15
0.05
0.7
0.4
8
0 - 8
1
DETAIL A
TYPICAL
1.88
1.58
4214864/A 05/2020
PowerPAD is a trademark of Texas Instruments.
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.15 mm per side.
4. This dimension does not include interlead flash. Interlead flash shall not exceed 0.25 mm per side.
5. Reference JEDEC registration MO-187, variation BA-T.
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EXAMPLE BOARD LAYOUT
DGQ0010A
PowerPADTM - 1.1 mm max height
PLASTIC SMALL OUTLINE
(2.2)
NOTE 9
(1.88)
SOLDER MASK
OPENING
SOLDER MASK
DEFINED PAD
SEE DETAILS
10X (1.45)
10X (0.3)
1
10
(1.3)
TYP
(2.05)
SOLDER MASK
OPENING
SYMM
(3.1)
NOTE 9
8X (0.5)
6
5
(R0.05) TYP
SYMM
METAL COVERED
BY SOLDER MASK
(
0.2) TYP
VIA
(1.3) TYP
(4.4)
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE:15X
SOLDER MASK
OPENING
METAL UNDER
SOLDER MASK
SOLDER MASK
OPENING
METAL
EXPOSED
METAL
EXPOSED
METAL
0.05 MAX
ALL AROUND
0.05 MIN
ALL AROUND
SOLDER MASK
DEFINED
NON SOLDER MASK
DEFINED
SOLDER MASK DETAILS
4214864/A 05/2020
NOTES: (continued)
6. Publication IPC-7351 may have alternate designs.
7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).
9. Size of metal pad may vary due to creepage requirement.
www.ti.com
EXAMPLE STENCIL DESIGN
DGQ0010A
PowerPADTM - 1.1 mm max height
PLASTIC SMALL OUTLINE
(1.88)
BASED ON
0.125 THICK
STENCIL
10X (1.45)
10X (0.3)
1
10
(2.05)
SYMM
BASED ON
0.125 THICK
STENCIL
8X (0.5)
5
6
(R0.05) TYP
SEE TABLE FOR
SYMM
(4.4)
DIFFERENT OPENINGS
FOR OTHER STENCIL
THICKNESSES
METAL COVERED
BY SOLDER MASK
SOLDER PASTE EXAMPLE
EXPOSED PAD
100% PRINTED SOLDER COVERAGE BY AREA
SCALE:15X
STENCIL
THICKNESS
SOLDER STENCIL
OPENING
0.1
2.10 X 2.29
1.88 X 2.05 (SHOWN)
1.72 X 1.87
0.125
0.150
0.175
1.59 X 1.73
4214864/A 05/2020
NOTES: (continued)
10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
11. Board assembly site may have different recommendations for stencil design.
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MECHANICAL DATA
DSC0010A
SDA10A (Rev A)
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Copyright © 2022, Texas Instruments Incorporated
相关型号:
LM2833XSDX
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