ISL6267HRZ-T7 [RENESAS]
DUAL SWITCHING CONTROLLER;型号: | ISL6267HRZ-T7 |
厂家: | RENESAS TECHNOLOGY CORP |
描述: | DUAL SWITCHING CONTROLLER 开关 |
文件: | 总33页 (文件大小:1091K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
Multiphase PWM Regulator for AMD Fusion™ Mobile
CPUs
ISL6267
Features
The ISL6267 is designed to be completely compliant with AMD
Fusion™ specifications. The ISL6267 controls two Voltage
Regulators (VRs), with three integrated gate drivers. The first
VR can be configured as 3-, 2-, or 1-phase VR, while the second
output can be configured as 2- or 1-phase VR, providing
maximum flexibility. The two VRs share the serial control bus
to communicate with the CPU and achieve lower cost and
smaller board area compared with two-chip solutions.
• Supports AMD SVI 1.0 serial data bus interface
• Dual output controller with integrated drivers
- Core VR configurable 3-, 2-, 1-phase with two integrated
drivers
- Northbridge VR configurable 2- or 1-phase with one
integrated driver
• Precision voltage regulation
- 0.5% system accuracy over-temperature
- 0V to 1.55V in 12.5mV steps
- Enhanced load line accuracy
The PWM modulator of the ISL6267 is based on Intersil’s R3
(Robust Ripple Regulator) Technology™. Compared with the
traditional multi-phase buck regulator, the R3 modulator
commands variable switching frequency during load
transients, achieving faster transient response. With the same
modulator, it naturally goes into pulse frequency modulation in
light load conditions, which achieves higher light load
efficiency and extends battery life.
• Supports multiple current sensing methods
- Lossless inductor DCR current sensing
- Precision resistor current sensing
• Programmable 1-, 2- or 3-phase for the core output and 1- or
2-phase for the northbridge output
The ISL6267 has several other key features. Both outputs
support DCR current sensing with a single NTC thermostat for
DCR temperature compensation or accurate resistor current
sensing. Both of the outputs utilize remote voltage sense,
adjustable switching frequency, current monitor, OC
protection, independent power-good indicators, temperature
monitors, and a common thermal alert.
• Adaptive body diode conduction time reduction
• Superior noise immunity and transient response
• Output current monitor and thermal monitor
• Differential remote voltage sensing
• High efficiency across entire load range
• Programmable +VID offset for both core and NB
• Programmable switching frequency for both outputs
• Excellent dynamic current balance between phases
• OCP/WOC, OVP, PGOOD, and thermal monitor
• Small footprint 48 Ld 6x6 QFN package
• Pb-free (RoHS compliant)
Applications
• AMD fusion CPU/GPU core power
• Notebook computer
Core Performance on ISL6267EVAL1Z
100
1.12
1.10
1.08
1.06
90
80
V
= 8V
IN
70
60
50
40
30
20
10
0
V
= 12V
IN
V
= 8V
IN
V
= 19V
IN
1.04
1.02
1.00
0.98
0.96
V
= 12V
IN
V
= 19V
IN
V
CORE = 1.1V
OUT
V
CORE = 1.1V
OUT
5
0
5
10 15 20 25 30 35 40 45 50 55
(A)
0
10 15 20 25 30 35 40 45 50 55
I
OUT
I
(A)
OUT
FIGURE 2. V
vs LOAD
FIGURE 1. EFFICIENCY vs LOAD
OUT
FN7801.1
January 8, 2013
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 |Copyright Intersil Americas Inc. 2011, 2013. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
1
ISL6267
Simplified Application Circuit For High Power CPU Core
BOOT_NB
VIN
UG1_NB
VNB
ISEN1_NB
PH1_NB
ISEN2_NB
LG1_NB
NB_PH1
VNB1
VNB1
VNB2
ISUMN_NB
ISUMP_NB
VW_NB
PWM2_NB
NTC_NB
NB_PH2
VNB2
COMP_NB
FB_NB
FB2_NB
PROG1
PROG2
VSEN_NB
RTN_NB
THERMAL INDICATOR
VR_HOT
NTC
VNB_SENSE
ISL6267
VIN
PWROK
SVD
µP
SVC
PWM3
VW
VO3
PH3
COMP
VIN
BOOT2
FB
UG2
PH2
VSEN
VCORE
VCORE_SENSE
RTN
LG2
PH2
VO2
VIN
ISEN3
ISEN2
BOOT1
UG1
ISEN1
ISUMN
VO1
VO2
VO3
PH1
LG1
PH1
VO1
ISUMP
FIGURE 3. TYPICAL APPLICATION CIRCUIT USING INDUCTOR DCR SENSING
January 8, 2013
FN7801.1
2
ISL6267
Simplified Application Circuit For AMD Torpedo Platform
NTC_NB
ISEN1_NB
VIN
BOOT_NB
ISEN2_NB
UG1_NB
ISUMN_NB
ISUMP_NB
VNB1
VNB2
VNB
PH1_NB
LG1_NB
NB_PH1
VIN
VNB1
VW_NB
COMP_NB
PWM2_NB
FB_NB
NB_PH2
VNB2
FB2_NB
VSEN_NB
RTN_NB
NTC
VNB_SENSE
THERMAL INDICATOR
VR_HOT
ISL6267
PROG1
PROG2
PWROK
SVD
µP
SVC
PWM3
+5V
VW
VIN
BOOT2
COMP
UG2
PH2
VCORE
FB
ISEN3/FB2
LG2
PH2
VO2
VSEN
RTN
VCORE_SENSE
VIN
BOOT1
UG1
PH1
LG1
ISEN2
ISEN1
PH1
VO1
VO1
ISUMN
ISUMP
VO2
FIGURE 4. TYPICAL APPLICATION CIRCUIT USING INDUCTOR DCR SENSING
January 8, 2013
FN7801.1
3
ISL6267
Simplified Application Circuit For Low Power CPU Core And NB
+5V
ISEN2_NB
OPEN
ISEN1_NB
ISUMN_NB
VIN
BOOT_NB
UG1_NB
PH1_NB
VNB1
VNB
ISUMP_NB
LG1_NB
NB_PH1
VNB1
NTC_NB
VW_NB
optional
OPEN
PWM2_NB
COMP_NB
FB_NB
PROG1
PROG2
VSEN_NB
VNB_SENSE
RTN_NB
NTC
THERMAL INDICATOR
VR_HOT
PWROK
SVD
µP
ISL6267
SVC
PWM3
+5V
BOOT2
OPEN
OPEN
OPEN
ISEN1
ISEN2
ISEN3
VW
UG2
PH2
OPEN
OPEN
+5V
LG2
OPEN
OPEN
optional
COMP
FB
PGND2
BOOT1
UG1
VIN
VCORE
VSEN
RTN
PH1
LG1
VCORE_SENSE
VO1
PH1
VO1
ISUMN
ISUMP
FIGURE 5. TYPICAL APPLICATION CIRCUIT USING INDUCTOR DCR SENSING
January 8, 2013
FN7801.1
4
ISL6267
Simplified Application Circuit Showing Resistor Sensing
+5V
ISEN2_NB
OPEN
ISEN1_NB
ISUMN_NB
VIN
BOOT_NB
UG1_NB
PH1_NB
LG1_NB
NB_N
VNB
ISUMP_NB
VW_NB
NB_N
NB_P
NTC_NB
optional
OPEN
PWM2_NB
COMP_NB
FB_NB
PROG1
PROG2
VSEN_NB
RTN_NB
VNB_SENSE
NTC
THERMAL INDICATOR
VR_HOT
PWROK
SVD
µP
ISL6267
SVC
PWM3
+5V
OPEN
BOOT2
UG2
OPEN
OPEN
ISEN1
ISEN2
ISEN3
VW
OPEN
OPEN
+5V
PH2
OPEN
OPEN
LG2
optional
PGND2
BOOT1
UG1
COMP
FB
VIN
VCORE
VSEN
RTN
PH1
LG1
VCORE_SENSE
VON
VON
VOP
ISUMN
ISUMP
FIGURE 6. TYPICAL APPLICATION CIRCUIT USING INDUCTOR DCR SENSING
January 8, 2013
FN7801.1
5
ISL6267
Block Diagram
SVD
SVC
SERIAL
VID
PROG1
PROG2
VCORE
VNB
A/D
INTERFACE
BOOT_NB
UG_NB
PWROK
NORTHBRIDGE
CONTROLLER AND DRIVER
DAC1
DAC2
(SIMILAR ARCHITECTURE TO
CORE SECTION)
PH_NB
SLEEP
MODE
(PSI_L)
LG_NB
D/A
PWM2_NB
VDD
ISEN1
PHASE
IBAL
PROG1
PROG2
OFFSET
VOLTAGE
CURRENT
ISEN2
BALANCE
ISEN3/FB2
PWM3
NTC_NB
NTC
T_MONITOR
BOOT2
UG2
TEMP
MONITOR
DRIVER
VR_HOT
PH2
VW
DAC1
DRIVER
DRIVER
+
LG2
RTN
Σ
+
+
MODULATOR
BOOT1
UG1
E/A
-
FB
PH1
COMP
VCCP
LG1
DRIVER
+
-
ISUMP
ISUMN
OC AND WOC
CURRENT
SENSE
PROTECTION
PGOOD
OV
PROTECTION
VSEN
January 8, 2013
FN7801.1
6
ISL6267
Pin Configuration
ISL6267
(48 LD QFN)
TOP VIEW
43
48 47 46 45 44
42 41 40 39 38 37
PWM2_NB
BOOT2
UG2
1
36
35
34
33
32
31
30
29
28
27
26
25
FB2_NB
FB_NB
COMP_NB
VW_NB
2
3
4
5
6
7
8
9
PH2
LG2
PGOOD_NB
SVD
GND PAD
(BOTTOM)
VCCP
PWM3
LG1
PWROK
SVC
PH1
ENABLE
PGOOD 10
UG1
BOOT1
PROG1
11
12
VR_HOT
NTC
3 7
Pin Descriptions
PIN NUMBER
SYMBOL
DESCRIPTION
1
FB2_NB
The components connecting to FB2_NB are used to adjust the compensation in 1-phase mode to achieve
optimum performance.
2
3
4
FB_NB
COMP_NB
VW_NB
Output voltage feedback to the inverting input of the Northbridge controller error amplifier.
Northbridge VR error amplifier output.
Window voltage set pin used to set the switching frequency for the Northbridge controller. A resistor from
this pin to COMP_NB programs the switching frequency (8kΩ gives approximately 300kHz).
5
PGOOD_NB
Open-drain output to indicate the Northbridge portion of the IC is ready to supply regulated voltage.
Pull-up externally to VCCP or 3.3V.
6
7
SVD
Serial VID data bi-directional signal from the CPU processor master device to the VR.
2
PWROK
System power good input. When this pin is high, the SVI interface is active and the I C protocol is running.
While this pin is low, the SVC and SVD input states determine the pre-PWROK metal VID. This pin must
be low prior to the ISL6267 PGOOD output going high per the AMD SVI Controller Guidelines.
8
9
SVC
Serial VID clock input from the CPU processor master device.
Enable input. A high level logic on this pin enables both VRs.
ENABLE
PGOOD
10
Open-drain output to indicate the Core portion of the IC is ready to supply regulated voltage. Pull-up
externally to VCCP or 3.3V.
11
12
13
VR_HOT
NTC
Thermal overload open drain output indicator active LOW.
Thermistor input to VR_HOT circuit to monitor Core VR temperature.
VW
Window voltage set pin used to set the switching frequency for the Core controller. A resistor from this
pin to COMP programs the switching frequency (8kΩ gives approximately 300kHz).
14
15
COMP
FB
Error amplifier output.
Output voltage feedback to the inverting input of the Core controller error amplifier.
January 8, 2013
FN7801.1
7
ISL6267
Pin Descriptions(Continued)
PIN NUMBER
SYMBOL
DESCRIPTION
16
ISEN3/FB2
When the Core VR of ISL6267 is configured in 3-phase mode, this pin is ISEN3. ISEN3 is the individual
current sensing for Channel 3. When the Core VR of ISL6267 is configured in 2-phase mode, this pin is
FB2. There is a switch between the FB2 pin and the FB pin. The switch is on in 2-phase mode and is off
in 1-phase mode. The components connecting to FB2 are used to adjust the compensation in 1-phase
mode to achieve optimum performance.
17
ISEN2
Individual current sensing for Channel 2 of the Core VR. When ISEN2 is pulled to 5V VDD, the controller
disables Channel 2, and the Core VR runs in single-phase mode.
18
19
20
ISEN1
VSEN
RTN
Individual current sensing for Channel 1 of the Core output.
Output voltage sense pin for the Core controller. Connect to the +sense pin of the microprocessor die.
Output voltage sense return pin for the Core controller. Connect to the -sense pin of the microprocessor
die.
21
22
23
24
25
26
ISUMN
ISUMP
VDD
Inverting input of the transconductance amplifier for current monitor and load line of Core output.
Non-inverting input of the transconductance amplifier for current monitor and load line of Core output.
5V bias power.
VIN
Battery supply voltage, used for feed-forward.
PROG1
BOOT1
Program pin for setting output voltage offset for Core VR.
Connect an MLCC capacitor across the BOOT1 and the phase (PH1) pin. The boot capacitor is charged
through an internal boot diode connected from the VCCP pin to the BOOT1 pin, each time the PH1 pin
drops below VCCP minus the voltage dropped across the internal boot diode.
27
28
UG1
PH1
Output of the Phase 1 high-side MOSFET gate driver of the Core VR. Connect the UG1 pin to the gate of
the Phase 1 high-side MOSFET.
Current return path for the Phase 1 high-side MOSFET gate driver of VR1. Connect the PH1 pin to the node
consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor of
Phase 1.
29
30
31
32
33
LG1
PWM3
VCCP
LG2
Output of the Phase 1 low-side MOSFET gate driver of the Core VR. Connect the LG1 pin to the gate of the
Phase 1 low-side MOSFET.
PWM output for Channel 3 of the Core VR. When PWM3 is pulled to 5V VDD, the controller disables Phase
3 and runs in 2-phase mode.
Input voltage bias for the internal gate drivers. Connect +5V to the VCCP pin. Decouple with at least 1µF
of capacitance to GND. A high quality, X7R dielectric MLCC capacitor is recommended.
Output of the Phase 2 low-side MOSFET gate driver of VR1. Connect the LG2 pin to the gate of the
Phase 2 low-side MOSFET.
PH2
Current return path for the Phase 2 high-side MOSFET gate driver of the Core VR. Connect the PH2 pin to
the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor
of Phase 2.
34
35
UG2
Output of the Phase 2 high-side MOSFET gate driver of the Core VR. Connect the UG2 pin to the gate of
the Phase 2 high-side MOSFET.
BOOT2
Connect an MLCC capacitor across the BOOT2 and PH2 pins. The boot capacitor is charged through an
internal boot diode connected from the VCCP pin to the BOOT2 pin, each time the PH2 pin drops below
VCCP minus the voltage dropped across the internal boot diode.
36
37
PWM2_NB
LG1_NB
PWM output for Channel 2 of the Northbridge VR.
Output of the low-side MOSFET gate driver of the Northbridge VR. Connect the LG1_NB pin to the gate of
the low-side MOSFET of VR2.
38
39
PH1_NB
UG1_NB
Current return path for the high-side MOSFET gate driver of the Northbridge VR. Connect the PH1_NB pin
to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output
inductor of the Northbridge VR.
Output of the high-side MOSFET gate driver of the Northbridge VR. Connect the UG1_NB pin to the gate
of the high-side MOSFET.
January 8, 2013
FN7801.1
8
ISL6267
Pin Descriptions(Continued)
PIN NUMBER
SYMBOL
DESCRIPTION
40
BOOT1_NB
Connect an MLCC capacitor across the BOOT1_NB and the PH1_NB pins. The boot capacitor is charged
through an internal boot diode connected from the VCCP pin to the BOOT1_NB pin, each time the
PH1_NB pin drops below VCCP minus the voltage dropped across the internal boot diode.
41
42
43
44
PROG2
NTC_NB
Program pin for setting output voltage offset for Northbridge VR.
Thermistor input to VR_HOT circuit to monitor Northbridge VR temperature.
Inverting input of the transconductance amplifier for current monitor and load line of the Northbridge VR.
ISUMN_NB
ISUMP_NB
Non-inverting input of the transconductance amplifier for current monitor and load line of the
Northbridge VR.
45
46
47
48
RTN_NB
VSEN_NB
ISEN2_NB
Output voltage sense return pin for the Northbridge controller. Connect to the -sense pin of the
microprocessor die.
Output voltage sense pin for the Northbridge controller. Connect to the +sense pin of the microprocessor
die.
Individual current sensing for Channel 2 of the Northbridge VR. When ISEN2 is pulled to 5V VDD, the
controller will disable Channel 2 and the Northbridge VR will run single-phase.
ISEN1_NB
Individual current sensing for Channel 1 of the Northbridge VR.
GND (Bottom Pad)
Signal common of the IC. Unless otherwise stated, signals are referenced to the GND pin.
Ordering Information
PART NUMBER
PART
TEMP.
RANGE (°C)
PACKAGE
(Pb-free)
PKG.
DWG. #
(Notes 1, 2, 3)
MARKING
ISL6267HRZ
ISL6267 HRZ
-10 to +100
-40 to +100
48 Ld 6x6 QFN
48 Ld 6x6 QFN
L48.6x6B
L48.6x6B
ISL6267IRZ
ISL6267EVAL1Z
NOTES:
ISL6267 IRZ
Evaluation Board
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL6267. For more information on MSL please see tech brief TB363.
January 8, 2013
9
FN7801.1
ISL6267
Table of Contents
Absolute Maximum Ratings. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Thermal Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Recommended Operating Conditions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Electrical Specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Gate Driver Timing Diagram. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Theory of Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Multiphase R3™ Modulator. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Diode Emulation and Period Stretching . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Start-up Timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Power-On Reset . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Serial VID Interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Pre-PWROK Metal VID . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16
VFIX Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
SVI Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .17
VID-on-the-Fly Transition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
SVI WIRE Protocol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
SVI Bus Protocol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .19
VR Offset Programming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .19
Voltage Regulation and Load Line Implementation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Differential Sensing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .20
Phase Current Balancing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .20
CCM Switching Frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Modes of Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Dynamic Operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Protections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
FB2 Function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .23
Adaptive Body Diode Conduction Time Reduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Key Component Selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Inductor DCR Current-Sensing Network. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Resistor Current-Sensing Network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Overcurrent Protection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Load Line Slope . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27
Compensator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27
Current Balancing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27
NTC Thermal Monitors and VR_HOT Function. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Layout Guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Revision History . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Products . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Package Outline Drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
January 8, 2013
10
FN7801.1
ISL6267
Absolute Maximum Ratings
Thermal Information
Supply Voltage, V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V
Battery Voltage, V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +28V
IN
Thermal Resistance (Typical)
48 Ld QFN Package (Notes 4, 5) . . . . . . . .
θ
JA (°C/W)
28
θ
JC (°C/W)
DD
1
Boot Voltage (BOOT). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V
Boot to Phase Voltage (BOOT-PHASE) . . . . . . . . . . . . . . . . -0.3V to +7V(DC)
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +9V(<10ns)
Phase Voltage (PHASE) . . . . . . . . . . . . . . . . -7V (<20ns Pulse Width, 10µJ)
UGATE Voltage (UGATE) . . . . . . . . . .PHASE - 0.3V (DC) to BOOTPHASE - 5V
. . . . . . . . . . . . . . . . . (<20ns Pulse Width, 10µJ) to BOOT LGATE Voltage
. . . . . . . . . . . . . . . . . . . . . -2.5V (<20ns Pulse Width, 5µJ) to VDD + 0.3V
All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to (VDD + 0.3V)
Open Drain Outputs, PGOOD, VR_HOT. . . . . . . . . . . . . . . . . . . -0.3V to +7V
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . .+150°C
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . .+150°C
Storage Temperature Range. . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Supply Voltage, V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+5V ±5%
DD
Battery Voltage, V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5V to 25V
IN
Ambient Temperature
HRZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-10°C to +100°C
IRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +100°C
Junction Temperature
HRZ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-10°C to +125°C
IRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. θ is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
JA
Brief TB379.
5. For θ , the “case temp” location is the center of the exposed metal pad on the package underside.
JC
Electrical Specifications Operating Conditions: V = 5V, T = -10°C to +100°C (HRZ), T = -40°C to +100°C (IRZ), f = 300kHz,
DD
A
A
SW
unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C.
MIN
MAX
PARAMETER
INPUT POWER SUPPLY
+5V Supply Current
SYMBOL
TEST CONDITIONS
(Note 6)
TYP
9
(Note 6)
UNITS
I
ENABLE = 1V
ENABLE = 0V
ENABLE = 0V
ENABLE = 1V
10.5
1
mA
µA
µA
kΩ
VDD
Battery Supply Current
I
1
VIN
V
Input Resistance
R
550
IN
VIN
POWER-ON-RESET THRESHOLDS
VDD POR Threshold
VDD_POR
VDD_POR
V
V
V
V
rising
4.35
4.15
4.00
3.30
4.5
V
V
V
V
r
DD
DD
falling
4.00
2.8
f
VIN POR Threshold
VIN_POR
VIN_POR
rising
4.35
r
IN
IN
falling
f
SYSTEM AND REFERENCES
System Accuracy
HRZ% Error
(V
No load; closed loop, active mode range
VID = 0.75V to 1.55V
)
-0.5
-8
+0.5
+8
%
CC_CORE
VID = 0.50V to 0.7375V
VID = 0.25V to 0.4875V
mV
mV
-15
+15
IRZ% Error
(V
No load; closed loop, active mode range
VID = 0.75V to 1.55V
)
-0.8
-10
-18
+0.8
+10
+18
%
mV
mV
V
CC_CORE
VID = 0.50V to 0.7375V
VID = 0.25V to 0.4875V
VID = [0000000]
Maximum Output Voltage
Minimum Output Voltage
V
1.55
0.0
CC_CORE(max)
V
VID = [1111111]
V
CC_CORE(min)
January 8, 2013
FN7801.1
11
ISL6267
Electrical Specifications Operating Conditions: V = 5V, T = -10°C to +100°C (HRZ), T = -40°C to +100°C (IRZ), f = 300kHz,
A
A
DD
SW
unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. (Continued)
MIN
MAX
PARAMETER
CHANNEL FREQUENCY
Nominal Channel Frequency
SYMBOL
TEST CONDITIONS
(Note 6)
TYP
(Note 6)
UNITS
HRZ, f
280
275
200
300
300
320
325
500
kHz
kHz
kHz
SW(nom)
IRZ, f
SW(nom)
Adjustment Range
AMPLIFIERS
Current-Sense Amplifier Input Offset
Error Amp DC Gain
I
= 0A
-0.15
+0.15
mV
dB
FB
A
90
18
v0
GBW
Error Amp Gain-Bandwidth Product
ISEN
C = 20pF
MHz
L
Imbalance Voltage
Maximum of ISENs - Minimum of ISENs
1
mV
nA
Input Bias Current
20
POWER-GOOD AND PROTECTION MONITORS
PGOOD Low Voltage
PGOOD Leakage Current
PGOOD Delay
V
I
= 4mA
PGOOD
0.26
460
0.4
1
V
OL
I
PGOOD = 3.3V
-1
µA
µs
OH
t
PGD
GATE DRIVER
UGATE Pull-Up Resistance
UGATE Source Current
UGATE Sink Resistance
UGATE Sink Current
R
200mA Source Current
UGATE - PHASE = 2.5V
1.0
2.0
1.0
2.0
1.0
2.0
0.5
4.0
23
1.5
1.5
1.5
0.9
Ω
A
UGPU
I
UGSRC
R
250mA Sink Current
Ω
A
UGPD
I
UGATE - PHASE = 2.5V
UGSNK
LGATE Pull-Up Resistance
LGATE Source Current
LGATE Sink Resistance
LGATE Sink Current
R
250mA Source Current
LGATE - VSSP = 2.5V
Ω
A
LGPU
I
LGSRC
R
250mA Sink Current
Ω
A
LGPD
I
LGATE - VSSP = 2.5V
LGSNK
UGATE to LGATE Deadtime
LGATE to UGATE Deadtime
BOOTSTRAP DIODE
t
UGATE falling to LGATE rising, no load
LGATE falling to UGATE rising, no load
ns
ns
UGFLGR
LGFUGR
t
28
Forward Voltage
V
PVCC = 5V, I = 2mA
F
0.58
0.2
V
F
Reverse Leakage
I
V
= 25V
R
µA
R
PROTECTION
Overvoltage Threshold
Severe Overvoltage Threshold
Undervoltage Threshold
Current Imbalance Threshold
Core OCP Current Threshold
OV
VSEN rising above setpoint for > 1ms
V rising above threshold > 0.5 µs
O
200
260
270
1.800
330
9
330
400
mV
V
H
OV
HS
OV
VSEN falls below setpoint for > 1ms
One ISEN above another ISEN for > 1.2ms
3-Phase CCM, 2-Phase CCM, 1-Phase
3-Phase DE
mV
mV
μA
μA
μA
μA
μA
μA
μA
μA
μA
μA
H
HRZ
IRZ
50
16
60
70
24
36
70
24
36
70
36
70
36
20
2-Phase DE
24
30
3-Phase CCM, 2-Phase CCM, 1-Phase
3-Phase DE
49.5
16
60
20
2-Phase DE
23.5
50
30
Northbridge OCP Current Threshold
HRZ
IRZ
2-Phase CCM, 1-Phase
2-Phase DE
60
24
30
2-Phase CCM, 1-Phase
2-Phase DE
49.5
23.5
60
30
January 8, 2013
FN7801.1
12
ISL6267
Electrical Specifications Operating Conditions: V = 5V, T = -10°C to +100°C (HRZ), T = -40°C to +100°C (IRZ), f = 300kHz,
A
A
DD
SW
unless otherwise noted. Boldface limits apply over the operating temperature range, -40°C to +100°C. (Continued)
MIN
MAX
PARAMETER
LOGIC THRESHOLDS
SYMBOL
TEST CONDITIONS
(Note 6)
0.7
TYP
(Note 6)
UNITS
ENABLE Input Low
ENABLE Input High
PWM
V
0.3
V
V
IL
V
IH
PWM Output Low
V
Sinking 5mA
1.0
V
V
0L
PWM Output High
V
Sourcing 5mA
PWM = 2.5V
3.5
0H
PWM Tri-State Leakage
THERMAL MONITOR
NTC Source Current
NTC_NB Source Current
Thermal Monitor Trip Voltage
Thermal Monitor Reset Voltage
INPUTS
2
µA
NTC = 1.3V
57
57
67
67
µA
µA
V
NTC_NB = 1.3V
Falling Threshold
Rising Threshold
0.87
0.91
0.88
0.92
0.89
0.93
V
ENABLE Leakage Current
I
EN = 0V
EN = 1V
-1
0
18
µA
ENABLE
36
10
µA
Slew Rate (for VID Change)
Soft-Start Slew Rate
SR
5
7.5
mV/µs
mV/µs
SSR
1.25
1.875
2.5
SVI INTERFACE
PWROK, SVC, SVD Input Logic High
PWROK, SVC, SVD Input Logic Low
SVC, SVD Leakage
V
0.798
V
V
IH
V
0.57
1
IL
EN = 0V, SVC and SVD = 0V
EN = 5V, SVC and SVD = 1.8V
µA
µA
1
NOTE:
6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
Gate Driver Timing Diagram
PWM
t
LGFUGR
t
FU
t
RU
1V
UGATE
LGATE
1V
t
RL
t
FL
t
UGFLGR
January 8, 2013
FN7801.1
13
ISL6267
out-of-phase. If VR1 is in 1-phase mode, the master clock signal
will be distributed to Phase 1 only and be the Clock1 signal.
Theory of Operation
Multiphase R ™ Modulator
3
VW
The ISL6267 is a multiphase regulator implementing two voltage
HYSTERETIC
VCRM
regulators, V and VDDNB, on one chip controlled by AMD’s™
WINDOW
DD
SVI1™ protocol. V can be programmed for 1-, 2- or 3-phase
DD
COMP
operation. VDDNB can be configured for 1-phase or 2-phase
3
operation. Both regulators use the Intersil patented R ™ (Robust
MASTER
CLOCK
3
Ripple Regulator) modulator. The R ™ modulator combines the
best features of fixed frequency PWM and hysteretic PWM while
eliminating many of their shortcomings. Figure 7 conceptually
shows the multiphase R ™ modulator circuit, and Figure 8 shows
CLOCK1
PWM1
3
the operation principles.
MASTER CLOCK CIRCUIT
MASTER
VW
CLOCK2
PWM2
CLOCK
CLOCK1
CLOCK2
CLOCK3
COMP
VCRM
MASTER
CLOCK
PHASE
SEQUENCER
GMVO
CRM
CLOCK3
SLAVE CIRCUIT 1
PWM3
L1
IL1
PHASE1
CLOCK1
PWM1
VO
S
R
VW
Q
VW
CO
VCRS1
GM
CRS1
VCRS2
VCRS3 VCRS1
SLAVE CIRCUIT 2
L2
IL2
3
PHASE2
CLOCK2
PWM2
S
FIGURE 8. R ™ MODULATOR OPERATION PRINCIPLES IN
STEADY STATE
VW
Q
R
Each slave circuit has its own ripple capacitor C , whose voltage
rs
VCRS2
GM
mimics the inductor ripple current. A g amplifier converts the
m
CRS2
inductor voltage into a current source to charge and discharge
C . The slave circuit turns on its PWM pulse upon receiving the
SLAVE CIRCUIT 3
rs
L3
IL3
PHASE3
clock signal, and the current source charges C . When C
CLOCK3
PWM3
S
rs
rs
VW
Q
R
voltage V hits VW, the slave circuit turns off the PWM pulse,
Crs
and the current source discharges C .
rs
VCRS3
GM
Since the controller works with V , which are large amplitude
crs
CRS3
and noise-free synthesized signals, it achieves lower phase jitter
than conventional hysteretic mode and fixed PWM mode
controllers. Unlike conventional hysteretic mode converters, the
error amplifier allows the ISL6267 to maintain a 0.5% output
voltage accuracy.
3
FIGURE 7. R ™ MODULATOR CIRCUIT
A current source flows from the VW pin to the COMP pin, creating
a voltage window set by the resistor between the two pins. This
voltage window is called “VW window” in the following
discussion.
Figure 9 shows the operation principles during load insertion
response. The COMP voltage rises during load insertion,
generating the master clock signal more quickly, so the PWM
pulses turn on earlier, increasing the effective switching
frequency. This allows for higher control loop bandwidth than
conventional fixed frequency PWM controllers. The VW voltage
rises as the COMP voltage rises, making the PWM pulses wider.
During load release response, the COMP voltage falls. It takes
the master clock circuit longer to generate the next master clock
signal so the PWM pulse is held off until needed. The VW voltage
falls as the COMP voltage falls, reducing the current PWM pulse
width. This kind of behavior gives the ISL6267 excellent response
speed.
Inside the IC, the modulator uses the master clock circuit to
generate the clocks for the slave circuits. The modulator
discharges the ripple capacitor C with a current source equal
rm
to g V , where g is a gain factor. C voltage V
m o rm CRM
is a
m
sawtooth waveform traversing between the VW and COMP
voltages. It resets to VW when it hits COMP, and generates a
one-shot master clock signal. A phase sequencer distributes the
master clock signal to the slave circuits. If V is in 3-phase
DD
mode, the master clock signal is distributed to the three phases,
and the Clock 1~3 signals will be 120° out-of-phase. If VR1 is in
2-phase mode, the master clock signal is distributed to Phases 1
and 2, and the Clock1 and Clock2 signals will be 180°
The fact that all the phases share the same VW window voltage
also ensures excellent dynamic current balance among phases.
January 8, 2013
FN7801.1
14
ISL6267
VW
P H A S E
COMP
VCRM
U G A TE
LG A TE
MASTER
CLOCK
CLOCK1
PWM1
IL
CLOCK2
PWM2
FIGURE 10. DIODE EMULATION
CLOCK3
PWM3
CCM/DCM
BOUNDARY
VW
VCRS
VW
IL
VCRS1
VCRS3
VCRS2
LIGHT DCM
VW
VCRS
3
FIGURE 9. R ™ MODULATOR OPERATION PRINCIPLES IN LOAD
INSERTION RESPONSE
IL
Diode Emulation and Period Stretching
DEEP DCM
VW
The ISL6267 can operate in diode emulation (DE) mode to
improve light-load efficiency. In DE mode, the low-side MOSFET
conducts when the current is flowing from source to drain and
does not allow reverse current, thus emulating a diode. As
Figure 10 shows, when LGATE is on, the low-side MOSFET carries
current, creating negative voltage on the phase node due to the
voltage drop across the ON-resistance. The ISL6267 monitors the
current by monitoring the phase node voltage. It turns off LGATE
when the phase node voltage reaches zero to prevent the inductor
current from reversing the direction and creating unnecessary
power loss. If the load current is light enough, (see Figure 10). The
inductor current reaches and stays at zero before the next phase
node pulse, and the regulator is in discontinuous conduction
mode (DCM). If the load current is heavy enough, the inductor
current will never reaches 0A, and the regulator is in CCM,
although the controller is in DE mode.
VCRS
IL
FIGURE 11. PERIOD STRETCHING
Start-up Timing
With the controller's V and V voltages above their POR
DD IN
threshold, the start-up sequence begins when ENABLE exceeds the
logic high threshold. Figure 12 shows the typical start-up timing of
VR1 and VR2. The ISL6267 uses digital soft-start to rampup DAC
to the voltage programmed by the Metal VID. PGOOD is asserted
high and low at the end of the rampup. Similar results occur if
ENABLE is tied to V , with the soft-start sequence starting
800µs after V crosses the POR threshold.
DD
DD
Figure 11 shows the operation principle in diode emulation mode
at light load. The load gets incrementally lighter in the three cases
from top to bottom. The PWM on-time is determined by the VW
window size and therefore is the same, making the inductor
current triangle the same in the three cases. The ISL6267 clamps
the ripple capacitor voltage V
in DE mode to make it mimic the
CRS
inductor current. It takes the COMP voltage longer to hit V
,
CRS
naturally stretching the switching period. The inductor current
triangles move farther apart such that the inductor current
average value is equal to the load current. The reduced switching
frequency helps increase light-load efficiency.
January 8, 2013
FN7801.1
15
ISL6267
Hysteresis between the rising and the falling thresholds assure
VDD
the ISL6267 does not inadvertently turn off unless the bias
voltage drops substantially (see “Electrical Specifications” on
page 11).
SLEW RATE
1.875mV/µs
ENABLE
MetalVID
VID COMMAND
VOLTAGE
800µs
Serial VID Interface
DAC
The on-board Serial VID Interface (SVI) circuitry allows the
processor to directly control the Core and Northbridge voltage
reference levels within the ISL6267. The SVC and SVD states are
decoded according to the PWROK inputs as described in the
following sections. The ISL6267 uses a digital-to-analog
converter (DAC) to generate a reference voltage based on the
decoded SVI value. See Figure 13 for a simple SVI interface
timing diagram.
PGOOD
PWROK
VIN
FIGURE 12. TYPICAL SOFT-START WAVEFORMS
Power-On Reset
Before the controller has sufficient bias to guarantee proper
operation, the ISL6267 requires both a +5V input supply tied to
Pre-PWROK Metal VID
Typical motherboard start-up begins with the controller decoding
the SVC and SVD inputs to determine the pre-PWROK Metal VID
setting (see Table 1). Once the ENABLE input exceeds the rising
threshold, the ISL6267 decodes and locks the decoded value in
an on-board hold register.
V
V
and PV , as well as a battery or other input supply tied to
CC
, to exceed their respective rising power-on reset (POR)
CC
IN
thresholds. Once these thresholds are reached or exceeded, the
ISL6267 has enough bias to begin checking SVI inputs.
1
4
5
6
2
3
7
8
9
10
11
12
VCC
SVC
SVD
ENABLE
PWROK
V_SVI
V_SVI
METAL_VID
METAL_VID
V
/ V
CORE NB
PGOOD
Interval 1 to 2: ISL6267 waits to POR.
Interval 2 to 3: SVC and SVD are externally set to pre-Metal VID code.
Interval 3 to 4: ENABLE locks pre-Metal VID code. All outputs soft-start to this level.
Interval 4 to 5: PGOOD signal goes HIGH, indicating proper operation.
Interval 5 to 6: CPU detects PGOOD high, and drives PWROK high, to allow ISL6267 to prepare for SVI commands.
Interval 6 to 7: SVC and SVD data lines communicate change in VID code.
Interval 7 to 8: ISL6267 responds to VID-ON-THE-FLY code change.
Interval 8 to 9: PWROK is driven low, and ISL6267 returns all outputs to pre-PWROK Metal VID level.
Interval 9 to 10: PWROK driven high once again by CPU, and ISL6267 prepares for SVI commands.
Interval 10 to 11: SVC and SVD data lines communicate new VID code.
Interval 11 to 12: ISL6267 drives outputs to new VID code level.
Post 12: Enable falls, all internal drivers are tri-stated, and PGOOD is driven low.
FIGURE 13. SVI INTERFACE TIMING DIAGRAM: TYPICAL PRE-PWROK METAL VID START-UP
January 8, 2013
FN7801.1
16
ISL6267
If the PWROK input is de-asserted, then the controller steps both
TABLE 1. PRE-PWROK METAL VID CODES
OUTPUT VOLTAGE
the Core and the Northbridge VRs back to the stored pre-PWROK
metal VID level in the holding register from initial soft-start. No
attempt is made to read the SVC and SVD inputs during this time.
If PWROK is re-asserted, then the on-board SVI interface waits for
a set VID command.
SVC
0
SVD
(V)
0
1.1
1.0
0.9
0.8
0
1
If ENABLE goes low during normal operation, all internal drivers
are tri-stated and PGOOD is pulled low. This event clears the
pre-PWROK metal VID code and forces the controller to check
SVC and SVD upon restart.
1
0
1
1
The internal DAC circuitry begins to ramp Core and Northbridge
VRs to the decoded pre-PWROK Metal VID output level. The
digital soft-start circuitry ramps the internal reference to the
target gradually at a fixed rate of approximately 2mV/µs. The
controlled ramp of all output voltage planes reduces in-rush
current during the soft-start interval. At the end of the soft-start
interval, the PGOOD output transitions high, indicating all output
planes are within regulation limits.
A POR event on either VCC or VIN during normal operation shuts
down both regulators, and both PGOOD outputs are pulled low.
The pre-PWROK metal VID code is not retained.
VID-on-the-Fly Transition
Once PWROK is high, the ISL6267 detects this flag and begins
monitoring the SVC and SVD pins for SVI instructions. The
microprocessor follows the protocol outlined in the following
sections to send instructions for the VID-on-the-fly transitions.
The ISL6267 decodes the instruction and acknowledges the new
VID code. For the VID codes higher than the current VID level, the
ISL6267 begins stepping the commanded VR outputs to the new
VID target with a typical slew rate of 7.5mV/µs, which meets the
AMD requirements.
If the ENABLE input falls below the enable falling threshold, the
ISL6267 tri-states both outputs. PGOOD is pulled low with the
loss of ENABLE. The Core and Northbridge planes decay, based
on output capacitance and load leakage resistance. If bias to V
CC
falls below the POR level, the ISL6267 responds in the manner
previously described. Once V and ENABLE rise above their
CC
respective rising thresholds, the internal DAC circuitry reacquires
a pre-PWROK metal VID code, and the controller soft-starts.
When the VID codes are lower than the current VID level, the
ISL6267 checks the state of PSI_L. If PSI_L is high, the controller
begins stepping the regulator output to the new VID target with a
typical slew rate of -7.5mV/µs. If PSI_L is low, the controller
allows the output voltage to decay and slowly steps the DAC
down with the natural decay of the output. This allows the
controller to quickly recover and move to a high VID code if
commanded. AMD requirements under these conditions do not
require the regulator to meet the minimum slew rate
specification of -5mV/µs. In either case, the slew rate is not
allowed to exceed 10mV/µs. The ISL6267 does not change the
state of PGOOD (VCCPWRGD in AMD specifications), when a
VID-on-the-fly transition occurs.
VFIX Mode
The ISL6267 does not support VFIX Mode. In the event a CPU is
not present on a motherboard and the ISL6267 is powered on,
the state of SVC and SVD sets the pre-PWROK metal VID as the
“Pre-PWROK Metal VID” on page 16 and begins soft-starting.
SVI Mode
Once the controller has successfully soft-starts and PGOOD and
PGOOD_NB transition high, the processor can assert PWROK to
signal the ISL6267 to prepare for SVI commands. The controller
actively monitors the SVI interface for set VID commands to
move the plane voltages to start-up VID values. Details of the SVI
Bus protocol are provided in the “AMD Design Guide for Voltage
Regulator Controllers Accepting Serial VID Codes” specification.
SVI WIRE Protocol
The SVI WIRE protocol is based on the I C bus concept. Two wires
2
[serial clock (SVC) and serial data (SVD)], carry information
between the AMD processor (master) and the VR controller
(slave) on the bus. The master initiates and terminates SVI
transactions and drives the clock, SVC, during a transaction. The
AMD processor is always the master, and the voltage regulators
are the slaves. The slave receives the SVI transactions and acts
accordingly. Mobile SVI WIRE protocol timing is based on
Once a set VID command is received, the ISL6267 decodes the
information to determine which VR is affected and which VID
target is required (see Table 2). The internal DAC circuitry steps
the output voltage of the VR commanded to the new VID level.
During this time, one or more of the VR outputs could be
targeted. In the event either VR is commanded to power-off by
serial VID commands, the PGOOD signal remains asserted.
2
high-speed mode I C. See AMD publication #40182 for
additional details.
January 8, 2013
FN7801.1
17
ISL6267
.
TABLE 2. SERIAL VID CODES
SVID
[6:0]
VOLTAGE
(V)
SVID
[6:0]
VOLTAGE
(V)
SVID
[6:0]
VOLTAGE
(V)
SVID
[6:0]
VOLTAGE
(V)
000_0000b
000_0001b
000_0010b
000_0011b
000_0100b
000_0101b
000_0110b
000_0111b
000_1000b
000_1001b
000_1010b
000_1011b
000_1100b
000_1101b
000_1110b
000_1111b
001_0000b
001_0001b
001_0010b
001_0011b
001_0100b
001_0101b
001_0110b
001_0111b
001_1000b
001_1001b
001_1010b
001_1011b
001_1100b
001_1101b
001_1110b
001_1111b
1.5500
1.5375
1.5250
1.5125
1.5000
1.4875
1.4750
1.4625
1.4500
1.4375
1.4250
1.4125
1.4000
1.3875
1.3750
1.3625
1.3500
1.3375
1.3250
1.3125
1.3000
1.2875
1.2750
1.2625
1.2500
1.2375
1.2250
1.2125
1.2000
1.1875
1.1750
1.1625
010_0000b
010_0001b
010_0010b
010_0011b
010_0100b
010_0101b
010_0110b
010_0111b
010_1000b
010_1001b
010_1010b
010_1011b
010_1100b
010_1101b
010_1110b
010_1111b
011_0000b
011_0001b
011_0010b
011_0011b
011_0100b
011_0101b
011_0110b
011_0111b
011_1000b
011_1001b
011_1010b
011_1011b
011_1100b
011_1101b
011_1110b
011_1111b
1.1500
1.1375
1.1250
1.1125
1.1000
1.0875
1.0750
1.0625
1.0500
1.0375
1.0250
1.0125
1.0000
0.9875
0.9750
0.9625
0.9500
0.9375
0.9250
0.9125
0.9000
0.8875
0.8750
0.8625
0.8500
0.8375
0.8250
0.8125
0.8000
0.7875
0.7750
0.7625
100_0000b
100_0001b
100_0010b
100_0011b
100_0100b
100_0101b
100_0110b
100_0111b
100_1000b
100_1001b
100_1010b
100_1011b
100_1100b
100_1101b
100_1110b
100_1111b
101_0000b
101_0001b
101_0010b
101_0011b
101_0100b
101_0101b
101_0110b
101_0111b
101_1000b
101_1001b
101_1010b
101_1011b
101_1100b
101_1101b
101_1110b
101_1111b
0.7500
0.7375
0.7250
0.7125
0.7000
0.6875
0.6750
0.6625
0.6500
0.6375
0.6250
0.6125
0.6000
0.5875
0.5750
0.5625
0.5500
0.5375
0.5250
0.5125
0.5000
0.4875*
0.4750*
0.4625*
0.4500*
0.4375*
0.4250*
0.4125*
0.4000*
0.3875*
0.3750*
0.3625*
110_0000b
110_0001b
110_0010b
110_0011b
110_0100b
110_0101b
110_0110b
110_0111b
110_1000b
110_1001b
110_1010b
110_1011b
110_1100b
110_1101b
110_1110b
110_1111b
111_0000b
111_0001b
111_0010b
111_0011b
111_0100b
111_0101b
111_0110b
111_0111b
111_1000b
111_1001b
111_1010b
111_1011b
111_1100b
111_1101b
111_1110b
111_1111b
0.3500*
0.3375*
0.3250*
0.3125*
0.3000*
0.2875*
0.2750*
0.2625*
0.2500*
0.2375*
0.2250*
0.2125*
0.2000*
0.1875*
0.1750*
0.1625*
0.1500*
0.1375*
0.1250*
0.1125*
0.1000*
0.0875*
0.0750*
0.0625*
0.0500*
0.0375*
0.0250*
0.0125*
OFF
OFF
OFF
OFF
NOTE: *Indicates a VID not required for AMD Family 10h processors.
January 8, 2013
FN7801.1
18
ISL6267
See Table 3
SVID
5
4
3
2
1
0
7
5
4
2
1
6
6
3
0
SVC
SVD
SLAVE ADDRESS PHASE
DATA PHASE
FIGURE 14. SEND BYTE EXAMPLE
SVI Bus Protocol
VR Offset Programming
The AMD processor bus protocol is compliant with SMBus send
byte protocol for VID transactions (see Figure 14). During a send
byte transaction, the processor sends the start sequence
followed by the slave address of the VR for which the VID
command applies. The address byte must be configured
according to Table 3. The processor then sends the write bit. After
the write bit, if the ISL6267 receives a valid address byte, it
sends the acknowledge bit. The processor then sends the PSI-L
bit and VID bits during the data phase. The Serial VID 8-bit data
field encoding is outlined in Table 4. If the ISL6267 receives a
valid 8-bit code during the data phase, it sends the acknowledge
bit. Finally, the processor sends the stop sequence. After the
ISL6267 has detected the stop, it can then proceed with the
VID-on-the-fly transition.
A positive or negative offset is programmed for the Core VR using
a resistor to ground from the PROG1 pin and the Northbridge in a
similar manner from the PROG2 pin. Table 5 provides the resistor
value to select the desired output voltage offset
TABLE 5. PROGx PIN RESISTOR VALUE
RESISTOR
VALUE
[Ω]
PROG1
PROG1
VNBOFFSET
[mV]
V
CORE OFFSET
[mV]
0
50
50
590
43.75
37.50
31.25
25.00
18.75
12.50
6.25
43.75
37.50
31.25
25.00
18.75
12.50
6.25
1100
1690
2260
3160
4320
5620
6650
7870
9530
11500
14000
16500
18700
OPEN
TABLE 3. SVI SEND BYTE ADDRESS DESCRIPTION
BITS
DESCRIPTION
6:4 Always 110b
3
2
Reserved by AMD for future use
VDD1; if set, then the following data byte contains the VID for
VDD1 [Note: The ISL6267 does not support VDD1]
0.00
0.00
-6.25
-6.25
1
0
VDD0; if set, then the following data byte contains the VID for
VID0
-12.50
-18.75
-25.00
-31.25
-37.50
-43.75
-12.50
-18.75
-25.00
-31.25
-37.50
-43.75
VDDNB; if set then the following data byte contains the VID for
VIDNB
TABLE 4. SERIAL VID 8-BIT DATA FIELD ENCODING
DESCRIPTION
BITS
7
PSI_L:
=0 means the processor is at an optimal load for the regulators
to enter power-saving mode
=1 means the processor is not at an optimal load for the
regulators to enter power-saving mode
Voltage Regulation and Load Line
Implementation
After the start sequence, the ISL6267 regulates the output voltage
to the value set by the VID information, per Table 2. The ISL6267
controls the no-load output voltage to an accuracy of ±0.5% over
the range of 0.75V to 1.55V. A differential amplifier allows
voltage sensing for precise voltage regulation at the
microprocessor die.
6:0 SVID[6:0] as defined in Table 2.
Operation
After the start-up sequence, the ISL6267 begins regulating the
Core and Northbridge output voltages to the pre-PWROK metal
VID programmed. The controller monitors SVI commands to
determine when to enter power-saving mode, implement
dynamic VID changes, and shut down individual outputs.
January 8, 2013
FN7801.1
19
ISL6267
amplifier regulates the inverting and non-inverting input voltages
to be equal as shown in Equation 3:
Rdroop
VCCSENSE
+
-
VCC
+ V
= V
+ VSS
SENSE
(EQ. 3)
SENSE
DAC
Vdroop
droop
FB
VR LOCAL VO
“CATCH” RESISTOR
Idroop
Rewriting Equation 3 and substituting Equation 2 gives
Equation 4 is the exact equation required for load-line
implementation.
+
-
E/A
VIDs
COMP
VID<0:7>
VSSSENSE
DAC
X 1
Σ
VDAC
+
RTN
VSS
VCC
– VSS
= V
– R
× I
droop droop
(EQ. 4)
+
-
SENSE
SENSE
DAC
INTERNAL TO IC
The VCC
SENSE
and VSS
signals come from the processor die.
“CATCH” RESISTOR
SENSE
The feedback is an open circuit in the absence of the processor. As
Figure 15 shows, it is recommended to add a “catch” resistor to feed
the VR local output voltage back to the compensator, and to add
another “catch” resistor to connect the VR local output ground to the
RTN pin. These resistors, typically 10Ω~100Ω, provide voltage
feedback if the system is powered up without a processor installed.
FIGURE 15. DIFFERENTIAL SENSING AND LOAD LINE
IMPLEMENTATION
As the load current increases from zero, the output voltage
droops from the VID table value by an amount proportional to the
load current, to achieve the load line. The ISL6267 can sense the
inductor current through the intrinsic DC Resistance (DCR) of the
inductors, as shown in Figures 15 and 16, or through resistors in
series with the inductors as shown in Figure 17. In both methods,
Phase Current Balancing
Rdcr3
Rdcr2
Rdcr1
L3
L2
L1
Rpcb3
capacitor C voltage represents the inductor total currents. A
n
PHASE3
Risen
droop amplifier converts C voltage into an internal current
n
IL3
IL2
IL1
ISEN3
INTERNAL
source with the gain set by resistor R . The current source is used
i
Cisen
for load line implementation, current monitoring and overcurrent
protection.
Rpcb2
VO
INTERNAL
TO IC
PHASE2
Risen
ISEN2
Figure 15 shows the load-line implementation. The ISL6267
Cisen
drives a current source (I
Equation 1.
) out of the FB pin, as described by
droop
Rpcb1
PHASE1
Risen
2xV
Cn
R
i
ISEN1
----------------
I
=
(EQ. 1)
droop
Cisen
FIGURE 16. CURRENT BALANCING CIRCUIT
When using inductor DCR current sensing, a single NTC element
is used to compensate the positive temperature coefficient of the
copper winding, thus sustaining the load-line accuracy with
reduced cost.
The ISL6267 monitors individual phase average current by
monitoring the ISEN1, ISEN2, and ISEN3 voltages. Figure 16
shows the current balancing circuit recommended for the
ISL6267. Each phase node voltage is averaged by a low-pass
I
flows through resistor R
droop
and creates a voltage drop as
filter consisting of R
corresponding ISEN pin. R
and C
, and is presented to the
droop
shown in Equation 2.
isen
isen
should be routed to the inductor
isen
phase-node pad in order to eliminate the effect of phase node
parasitic PCB DCR. Equations 5 through 7 give the ISEN pin
voltages:
V
= R
× I
droop droop
(EQ. 2)
droop
V
= (R
+ R
) × I
V
is the droop voltage required to implement load line.
(EQ. 5)
droop
ISEN1
dcr1
pcb1
L1
Changing R
or scaling I
can change the load line slope.
droop
sets the overcurrent protection level, it is
droop
Since I
droop
V
= (R
+ R
) × I
recommended to first scale I
then select an appropriate R
load line slope.
based on OCP requirement,
value to obtain the desired
(EQ. 6)
(EQ. 7)
ISEN2
dcr2
pcb2
L2
droop
droop
V
= (R
+ R
) × I
ISEN3
dcr3
pcb3
L3
Differential Sensing
Figure 15 shows the differential voltage sensing scheme.
VCC and VSS are the remote voltage sensing signals
from the processor die. A unity gain differential amplifier senses
where R
, R
and R
are inductor DCR; R
, R
dcr1 dcr2
dcr3
pcb1 pcb2
SENSE SENSE
and R
are parasitic PCB DCR between the inductor output
pcb3
side pad and the output voltage rail; and I , I and I are
inductor average currents.
L1 L2
L3
the VSS
voltage and adds it to the DAC output. The error
SENSE
January 8, 2013
FN7801.1
20
ISL6267
The ISL6267 adjusts the phase pulse-width relative to the other
phases to make V = V = V , thus to achieve
Rewriting Equation 11 gives Equation 13:
(EQ. 13)
(EQ. 14)
(EQ. 15)
ISEN1 ISEN2 ISEN3
V
– V
= V – V
1n 2p 2n
1p
I
= I = I , when R = R = R
and
L1 L2 L3 dcr1 dcr2 dcr3
R
= R
= R
.
pcb1
pcb2
pcb3
Rewriting Equation 12 gives Equation 14:
Using the same components for L1, L2 and L3 provides a good
match of R , R and R . Board layout determines R
V
– V
= V – V
2n 3p 3n
2p
,
dcr1 dcr2 dcr3 pcb1
R
and R
. It is recommended to have symmetrical layout
Combining Equations 13 and 14 gives:
pcb2
pcb3
for the power delivery path between each inductor and the output
V
– V
= V – V
= V – V
3n
1p
1n
2p
2n
3p
voltage rail, such that R
= R
pcb2
= R
.
pcb1
pcb3
Rdcr3
L3
Therefore:
Rpcb3
V3p
PHASE3
(EQ. 16)
R
× I
= R
× I
= R
× I
dcr3 L3
Risen
Risen
Risen
dcr1
L1
dcr2
L2
IL3
IL2
IL1
ISEN3
Cisen
V3n
Current balancing (I = I = I ) is achieved when
L1 L2 L3
INTERNAL
TO IC
R
= R
= R
. R , R
and R do not have any
dcr1
effect.
dcr2 dcr3 pcb1 pcb2
pcb3
Rdcr2
L2
L1
Rpcb2
V
o
V2p
Risen
PHASE2
Since the slave ripple capacitor voltages mimic the inductor
currents, the R ™ modulator can naturally achieve excellent
ISEN2
V2n
3
Risen
Risen
Cisen
current balancing during steady state and dynamic operations.
Figure 18 shows the current balancing performance of the
evaluation board with load transient of 12A/51A at different rep
rates. The inductor currents follow the load current dynamic
change with the output capacitors supplying the difference. The
inductor currents can track the load current well at a low
repetition rate, but cannot keep up when the repetition rate gets
into the hundred-kHz range, where it is out of the control loop
bandwidth. The controller achieves excellent current balancing in
all cases installed.
Rdcr1
Rpcb1
V1p
PHASE1
Risen
Risen
Risen
ISEN1
Cisen
V1n
FIGURE 17. DIFFERENTIAL-SENSING CURRENT BALANCING
CIRCUIT
Sometimes, it is difficult to implement symmetrical layout. For
the circuit shown in Figure 16, asymmetric layout causes
CCM Switching Frequency
The R
resistor between the COMP and the VW pins sets the
fset
different R
, R
and R values, thus creating a current
pcb1 pcb2
pcb3
VW windows size and therefore sets the switching frequency.
When the ISL6267 is in continuous conduction mode (CCM), the
switching frequency is not absolutely constant due to the nature
imbalance. Figure 17 shows a differential sensing current
balancing circuit recommended for the ISL6267. The current
sensing traces should be routed to the inductor pads so they only
pick up the inductor DCR voltage. Each ISEN pin sees the average
voltage of three sources: its own, phase inductor phase-node
pad, and the other two phases inductor output side pads.
Equations 8 through 10 give the ISEN pin voltages:
3
of the R ™ modulator. As explained in the “Multiphase R3™
Modulator” on page 14, the effective switching frequency
increases during load insertion and decreases during load
release to achieve fast response. Thus, the switching frequency is
relatively constant at steady state. Variation is expected when
the power stage condition, such as input voltage, output voltage,
load, etc. changes. The variation is usually less than 15% and
does not have any significant effect on output voltage ripple
magnitude. Equation 17 gives an estimate of the
V
= V + V + V
1p 2n 3n
(EQ. 8)
ISEN1
V
V
= V + V + V
1n 2p
(EQ. 9)
ISEN2
ISEN3
3n
3p
frequency-setting resistor (R ) value. A value of 8kΩ R
gives
fset
fset
= V + V + V
(EQ. 10)
1n
2n
approximately 300kHz switching frequency. Lower resistance
gives higher switching frequency.
The ISL6267 will make V
Equations 11 and 12:
= V
= V as shown in
ISEN3
ISEN1
ISEN2
(EQ. 17)
R
(kΩ) = (Period(μs) – 0.29) × 2.65
fset
V
+ V + V
= V + V + V
1n 2p
(EQ. 11)
(EQ. 12)
1p
2n
3n
3n
3p
V
+ V + V
= V + V + V
1n 2n
1n
2p
3n
January 8, 2013
FN7801.1
21
ISL6267
Modes of Operation
REP RATE = 10kHz
TABLE 6. CORE VR MODES OF OPERATION
OCP
THRESHOLD
(µA)
PWM3
To
External Stage
Driver
ISEN2
CONFIG. PSL_L
MODE
3-phase CCM
1-phase DE
To Power 3-phase
1
0
60
20
CPU VR
Config.
Tied to 5V
2-phase
CPU VR
Config.
1
0
2-phase CCM
1-phase DE
60
30
Tied to
5V
1-phase
CPU VR
Config.
X
1-phase DE
60
REP RATE = 25kHz
The Core VR can be configured for 3, 2- or 1-phase operation.
Table 6 shows Core VR configurations and operational modes,
programmed by the PWM3 and ISEN2 pin status and the PS
command. For 2-phase configuration, tie the PWM3 pin to 5V. In
this configuration, phases 1 and 2 are active. For 1-phase
configuration, tie the PWM3 pin and the ISEN2 pin to 5V. In this
configuration, only phase-1 is an active, the controller operates in
DE mode and the PSI_L input been ignored.
In 3-phase configuration, Core VR operates in 3-phase CCM, with
PSI_L high. It enters 1-phase DE mode when PSI_L is low,
dropping phases 3 and 2, and reduces the overcurrent and the
way-overcurrent protection levels to one-third of the initial values.
REP RATE = 50kHz
In 2-phase configuration, Core VR operates in 2-phase CCM with
PSI_L high. It enters 1-phase DE mode with PSI_L low, by
dropping phase 2 and reduces the overcurrent and the way
overcurrent protection levels to one-half of the initial values.
In 1-phase configuration, the Core VR operates in 1-phase DE and
ignores the PSI_L input. If a resistor is placed from COMP pin to
GND with a value less than 150kΩ, then the Core VR operates in
1-phase CCM with PSI_L high and enters 1-phase DE mode when
PSI_L is low. A resistor value of 100kΩ is recommended.
REP RATE = 100kHz
TABLE 7. NORTHBRIDGE VR MODES OF OPERATION
OCP
THRESHOLD
ISEN2_NB
To Power
CONFIG.
PSL_L
MODE
2-phase CCM
1-phase DE
1-phase DE
(µA)
2-phase NB
VR Config.
1
0
X
60
Stage
30
Tied to 5V
1-phase NB
VR Config.
60
REP RATE = 200kHz
ISL6267 Northbridge (NB) VR can be configured for 2- or 1-phase
operation. Table 7 shows the Northbridge VR configurations and
operational modes, which are programmed by the ISEN2 pin
status and the PSI_L command. For 1-phase configuration, tie
the ISEN2_NB pin to 5V. In 1-phase configuration, the
Northbridge VR operates in 1-phase DE and ignores the PSI_L
input. If a resistor is placed from COMP_NB pin to GND with a
value less than 150kΩ, then the Northbridge VR operates in 1-
phase CCM with PSI_L high and enters 1-phase DE mode when
PSI_L is low. A resistor value of 100kΩ is recommended.
FIGURE 18. CURRENT BALANCING DURING DYNAMIC
OPERATION. CH1: I , CH2: I
, CH3: I , CH4:
L1
LOAD
L2
The Northbridge VR can be disabled completely by tying
ISUMN_NB to 5V.
I
L3
January 8, 2013
FN7801.1
22
ISL6267
threshold. When ENABLE and V return to their high operating
DD
levels, a soft-start occurs.
Dynamic Operation
Core VR and Northbridge VR behave the same during dynamic
operation. The controller responds to VID-on-the-fly changes by
slewing to the new voltage at the fixed 7.5mV/µs slew rate.
During negative VID transitions, the output voltage decays to the
lower VID value at the slew rate determined by the load.
Table 8 summarizes the fault protections.
TABLE 8. FAULT PROTECTION SUMMARY
FAULT DURATION
BEFORE
PROTECTION
PROTECTION
ACTION
FAULT
RESET
SVI_L low command prompts the controller to enter DE mode.
Overvoltage protection is blanked during VID down transition in
DE mode until the output voltage is within 60mV of the VID value.
FAULT TYPE
Overcurrent
120µs
1ms
PWM tri-state,
PGOOD latched toggle or
low
ENABLE
Phase Current
Unbalance
During load insertion response, the Fast Clock function increases
the PWM pulse response speed. The controller monitors the
VSEN pin voltage and compares it to 100ns-filtered version.
When the unfiltered version is 20mV below the filtered version,
the controller knows there is a fast voltage dip due to load
insertion, and it issues an additional master clock signal to
deliver a PWM pulse immediately.
V
toggle
DD
Way-Overcurrent
(1.5xOC)
Immediately
Overvoltage +200mV
PGOOD latched
low. Actively pulls
the output
voltage to below
VID value, then
tri-state.
3
The R ™ modulator intrinsically has voltage feed-forward. The
output voltage is insensitive to a fast slew rate input voltage
change.
Over-Temperature
400µs
N/A
Protections
Core VR and Northbridge VR both provide overcurrent,
current-balance and overvoltage fault protections. The controller
also provides over-temperature protection. The following
discussion is based on Core VR and also applies to Northbridge VR.
FB2 Function
The FB2 function is only available for Core VR or Northbridge VR
in 2-phase configuration.
C1
C1
R2
C3.1
C3.2
R2
C3.1
C3.2
CONTROLLER
IN
2-PHASE MODE
CONTROLLER
IN
1-PHASE MODE
The controller determines overcurrent protection (OCP) by
comparing the average value of the droop current (I
C2
R3
C2
R3
FB2
FB2
) with an
droop
internal current source threshold as Table 6 shows. It declares OCP
when I is above the threshold for 120µs.
R1
R1
VSEN
droop
VSEN
COMP
E/A
E/A
FB
FB
COMP
For overcurrent conditions above 1.5x the OCP level, the PWM
outputs immediately shuts off and PGOOD goes low to maximize
protection. This protection is also referred to as way-overcurrent
protection or fast overcurrent protection for short-circuit
protections.
VREF
VREF
FIGURE 19. FB2 FUNCTION
Figure 19 shows the FB2 function. A switch (called FB2 switch)
turns on to short the FB and the FB2 pins when the controller is in
2-phase mode. Capacitors C3.1 and C3.2 are in parallel, serving
as part of the compensator. When the controller enters 1-phase
mode, the FB2 switch turns off, removing C3.2 and leaving only
C3.1 in the compensator. The compensator gain increases with
the removal of C3.2. By properly sizing C3.1 and C3.2, the
compensator can be optimal for both 2-phase mode and 1-phase
mode.
The controller monitors the ISEN pin voltages to determine
current-balance protection. If the ISEN pin voltage difference is
greater than 9mV for 1ms, the controller will declare a fault and
latch off.
The controller takes the same actions for all of the previously
describe fault protections: de-assertion of PGOOD and turn-off of
the high-side and low-side power MOSFETs. Any residual inductor
current decays through the MOSFET body diodes.
When the FB2 switch is off, C3.2 is disconnected from the FB pin.
However, the controller still actively drives the FB2 pin voltage to
follow the FB pin voltage such that C3.2 voltage always follows
C3.1 voltage. When the controller turns on the FB2 switch, C3.2
is reconnected to the compensator smoothly.
The controller declares an overvoltage fault and de-asserts PGOOD if
the output voltage exceeds the VID set value by +250mV. The
ISL6267 immediately declares an OV fault, de-asserts PGOOD,
and turn on the low-side power MOSFETs. The low-side power
MOSFETs remain on until the output voltage is pulled down below
the VID set value when all power MOSFETs are turned off. If the
output voltage rises above the VID set value +250mV again, the
protection process is repeated. This behavior provides the
maximum amount of protection against shorted high-side power
MOSFETs while preventing output ringing below ground.
The FB2 function ensures excellent transient response in both
2-phase and 1-phase mode. If the FB2 function is not used,
populate C3.1 only.
Adaptive Body Diode Conduction Time
Reduction
In DCM, the controller turns off the low-side MOSFET when the
inductor current approaches zero. During on-time of the low-side
MOSFET, phase voltage is negative, and the amount is the
All of the previously described fault conditions can be reset by
bringing ENABLE low or by bringing V below the POR
DD
January 8, 2013
FN7801.1
23
ISL6267
MOSFET r
voltage drop, which is proportional to the
domain relationship between inductor total current I (s) and C
o
DS(ON)
n
inductor current. A phase comparator inside the controller
monitors the phase voltage during on-time of the low-side
MOSFET and compares it with a threshold to determine the zero
crossing point of the inductor current. If the inductor current has
not reached zero when the low-side MOSFET turns off, it will flow
through the low-side MOSFET body diode, causing the phase
node to have a larger voltage drop until it decays to zero. If the
inductor current has crossed zero and reversed the direction
when the low-side MOSFET turns off, it will flow through the
high-side MOSFET body diode, causing the phase node to have a
spike until it decays to zero. The controller continues monitoring
the phase voltage after turning off the low-side MOSFET. To
minimize the body diode-related loss, the controller also adjusts
the phase comparator threshold voltage accordingly in iterative
steps such that the low-side MOSFET body diode conducts for
approximately 40ns.
voltage V (s):
Cn
⎛
⎞
⎟
⎟
⎟
⎠
R
⎜
⎜
⎜
⎝
DCR
----------------------------------------- -----------
ntcnet
V
(s) =
×
× I (s) × A (s)
(EQ. 18)
Cn
o
cs
R
N
sum
--------------
+
R
ntcnet
N
(R
+ R ) × R
ntc p
ntcs
--------------------------------------------------
R
A
=
(EQ. 19)
(EQ. 20)
ntcnet
R
+ R
+ R
ntc p
ntcs
s
------
1 +
ω
L
s
----------------------
1 +
(s) =
cs
L
------------
ω
sns
DCR
-----------
=
ω
ω
(EQ. 21)
(EQ. 22)
L
Key Component Selection
Inductor DCR Current-Sensing Network
1
------------------------------------------------------
=
sns
R
sum
N
PHASE1 PHASE2 PHASE3
--------------
R
×
ntcnet
-----------------------------------------
× C
RSUM
n
R
sum
N
--------------
R
+
RSUM
ntcnet
where N is the number of phases.
ISUM+
RSUM
Transfer function A (s) always has unity gain at DC. The inductor
cs
DCR value increases as the winding temperature increases,
RNTCS
L
L
L
+
giving higher reading of the inductor DC current. The NTC R
ntc
CNVCN
RP
value decrease as its temperature decreases. Proper selection of
R , R , R and R parameters ensures that V
sum ntcs ntc Cn
-
RNTC
RO
p
DCR
DCR
DCR
represents the inductor total DC current over the temperature
range of interest.
ISUM-
RI
RO
RO
There are many sets of parameters that can properly temperature
compensate the DCR change. Since the NTC network and the R
sum
resistors form a voltage divider, V is always a fraction of the
cn
inductor DCR voltage. It is recommended to have a higher ratio of
to the inductor DCR voltage so the droop circuit has a higher
signal level to work with.
IO
V
cn
FIGURE 20. DCR CURRENT-SENSING NETWORK
A typical set of parameters that provide good temperature
Figure 20 shows the inductor DCR current-sensing network for a
3-phase solution. An inductor current flows through the DCR and
compensation are: R
= 3.65kΩ, R = 11kΩ, R = 2.61kΩ
sum
p
ntcs
and R = 10kΩ (ERT-J1VR103J). The NTC network parameters
creates a voltage drop. Each inductor has two resistors in R
ntc
sum
may need to be fine tuned on actual boards. One can apply full
load DC current and record the output voltage reading
immediately; then record the output voltage reading again when
the board has reached the thermal steady state. A good NTC
network can limit the output voltage drift to within 2mV. It is
recommended to follow the Intersil evaluation board layout and
current sensing network parameters to minimize engineering
time.
and R connected to the pads to accurately sense the inductor
o
current by sensing the DCR voltage drop. The R
and R
sum
o
resistors are connected in a summing network as shown, and feed
the total current information to the NTC network (consisting of
R
, R and R ) and capacitor C . R is a negative
ntcs ntc ntc
p
n
temperature coefficient (NTC) thermistor, used to temperature
compensate the inductor DCR change.
The inductor output side pads are electrically shorted in the
schematic but have some parasitic impedance in actual board
layout, which is why one cannot simply short them together for the
current-sensing summing network. It is recommended to use
1Ω~10Ω R to create quality signals. Since R value is much
V
(s) also needs to represent real-time I (s) for the controller to
Cn
o
achieve good transient response. Transfer function A (s) has a
cs
pole w
sns
and a zero w . One needs to match w and w so
sns
L
L
A
(s) is unity gain at all frequencies. By forcing w equal to w
cs sns
L
o
o
and solving for the solution. Equation 23 gives Cn value.
smaller than the rest of the current sensing circuit, the following
analysis ignores it.
The summed inductor current information is presented to the
capacitor C . Equations 18 thru 22 describe the frequency
n
January 8, 2013
FN7801.1
24
ISL6267
L
-----------------------------------------------------------
(EQ. 23)
C
=
i
o
n
R
i
sum
L
--------------
R
×
ntcnet
N
-----------------------------------------
× DCR
R
sum
--------------
R
+
ntcnet
N
For example, given N = 3, R
sum
= 3.65kΩ, R = 11kΩ,
p
V
o
R
= 2.61kΩ, R = 10kΩ, DCR = 0.88mΩ and L = 0.36µH,
ntcs
ntc
RING
BACK
Equation 23 gives C = 0.406µF.
n
Assuming the compensator design is correct, Figure 21 shows the
expected load transient response waveforms if C is correctly
n
FIGURE 24. OUTPUT VOLTAGE RING-BACK PROBLEM
selected. When the load current I
has a square change, the
also has a square response.
core
output voltage V
core
ISUM+
If C value is too large or too small, V (s) does not accurately
Cn
n
represent real-time I (s) and worsens the transient response.
o
Figure 22 shows the load transient response when C is too
n
small. V
sags excessively upon load insertion and may create
Rntcs
Cn.1
core
a system failure. Figure 23 shows the transient response when
C is too large. V is sluggish in drooping to its final value.
Vcn
Cn.2
Rp
n
core
There is excessive overshoot if load insertion occurs during this
time, which may negatively affect the CPU reliability.
Rn
Rntc
ISUM-
Ri
OPTIONAL
i
o
Cip
Rip
OPTIONAL
V
o
FIGURE 25. OPTIONAL CIRCUITS FOR RING-BACK REDUCTION
FIGURE 21. DESIRED LOAD TRANSIENT RESPONSE
WAVEFORMS
Figure 24 shows the output voltage ring-back problem during
load transient response. The load current i has a fast step
o
change, but the inductor current i cannot accurately follow.
L
i
o
Instead, i responds in first-order system fashion due to the
L
nature of the current loop. The ESR and ESL effect of the output
capacitors makes the output voltage V dip quickly upon load
o
current change. However, the controller regulates V according to
o
the droop current i
, which is a real-time representation of i ;
droop
L
V
o
therefore, it pulls V back to the level dictated by i , causing the
o
L
ring-back problem. This phenomenon is not observed when the
output capacitor has very low ESR and ESL, as is the case with all
ceramic capacitors.
FIGURE 22. LOAD TRANSIENT RESPONSE WHEN C IS TOO
n
SMALL
Figure 25 shows two optional circuits for reduction of the
ring-back. C is the capacitor used to match the inductor time
n
constant. It usually takes the parallel of two (or more) capacitors
to get the desired value. Figure 25 shows that two capacitors
i
o
(C and C ) are in parallel. Resistor R is an optional
n.1 n.2
n
component to reduce the V ring-back. At steady state,
o
C
+ C provides the desired C capacitance. At the beginning
n.1
n.2
n
of i change, the effective capacitance is less because R
o
n
increases the impedance of the C branch. As Figure 22 shows,
n.1
V
o
V tends to dip when C is too small, and this effect reduces the
o
n
V ring-back. This effect is more pronounced when C is much
o
n.1
larger than C . It is also more pronounced when R is bigger.
FIGURE 23. LOAD TRANSIENT RESPONSE WHEN C IS TOO
n
n.2
n
LARGE
However, the presence of R increases the ripple of the V signal
n n
if C is too small. It is recommended to keep C greater than
n.2 n.2
2200pF. R value usually is a few ohms. C , C and R values
n
n.1 n.2
n
should be determined through tuning the load transient response
January 8, 2013
FN7801.1
25
ISL6267
waveforms on an actual board. R and C form an R-C branch in
ip ip
Overcurrent Protection
Refer to Equation 1 on page 20 and Figures 20, 24 and 26;
parallel with R , providing a lower impedance path than R at the
i
i
beginning of i change. R and C do not have any effect at
o
ip ip
resistor R sets the droop current, I
. Tables 6 and 7 show the
i
droop
steady state. Through proper selection of R and C values,
ip ip
internal OCP threshold. It is recommended to design I
droop
i
can resemble i rather than i , and V will not ring back.
o L o
droop
without using the R
resistor.
comp
For example, the OCP threshold is 60µA for 3-phase solution.
is designed to be 40.9µA at full load. Therefore the OCP
The recommended value for R is 100Ω. C should be
ip ip
determined through tuning the load transient response
waveforms on an actual board. The recommended range for C
is 100pF~2000pF. However, it should be noted that the R -C
I
ip
ip ip
droop
trip level is 1.5x of the full load current.
branch may distort the i
triangular as the real inductor current, i
droop
spikes, which may adversely affect i
waveform. Instead of being
may have sharp
average value
droop
For inductor DCR sensing, Equation 27 gives the DC relationship
of V (s) and I (s):
cn
o
droop
⎛
⎜
⎜
⎜
⎝
⎞
⎟
⎟
⎟
⎠
detection and therefore may affect OCP accuracy. User discretion
is advised.
R
DCR
N
ntcnet
----------------------------------------- -----------
V
=
×
× I
Cn
o
(EQ. 27)
R
sum
--------------
+
R
ntcnet
N
Resistor Current-Sensing Network
PHASE1 PHASE2 PHASE3
Substitution of Equation 27 into Equation 1 gives Equation 28:
R
2
DCR
N
ntcnet
---- ----------------------------------------- -----------
× I
o
I
=
×
×
(EQ. 28)
droop
R
R
i
sum
L
L
L
--------------
+
R
ntcnet
N
Therefore:
DCR
DCR
DCR
2R
× DCR × I
o
RSUM
RSUM
RSUM
ntcnet
-------------------------------------------------------------------------------
R
=
(EQ. 29)
i
R
sum
N
⎛
⎞
⎠
--------------
N × R
+
× I
ntcnet
droop
⎝
ISUM+
Substitution of Equation 19 and application of the OCP condition
in Equation 29 gives Equation 30:
+
RSEN
RSEN
RSEN
VCN
CN
RI
ISUM-
RO
RO
RO
(R
+ R ) × R
ntc p
-
ntcs
--------------------------------------------------
2 ×
× DCR × I
omax
R
+ R
+ R
ntc p
ntcs
-------------------------------------------------------------------------------------------------------------------------
=
R
(EQ. 30)
i
(R
+ R ) × R
R
⎛
⎜
⎝
⎞
⎟
⎠
ntcs
ntc
p
sum
N
-------------------------------------------------- --------------
N ×
+
× I
droopmax
R
+ R
+ R
p
ntcs
ntc
IO
where I
is the full load current and I
is the
omax
droopmax
corresponding droop current. For example, given N = 3,
FIGURE 26. RESISTOR CURRENT-SENSING NETWORK
R
= 3.65kΩ, R = 11kΩ, R
= 2.61kΩ, R = 10kΩ,
sum
DCR = 0.88mΩ, I
p
ntcs ntc
= 51A and I
= 40.9µA.
Figure 26 shows the resistor current-sensing network for a
2-phase solution. Each inductor has a series current sensing
omax
droopmax
Equation 30 gives R = 606Ω.
i
resistor, R . R
accurately capture the inductor current information. The R
and R are connected to the R
pads to
sen sum
o
sen
For resistor sensing, Equation 31 gives the DC relationship of
sum
and C
n
V
(s) and I (s).
cn
o
and R resistors are connected to capacitor C . R
o
n
sum
R
sen
------------
form a filter for noise attenuation. Equations 24 thru 26 give the
(EQ. 31)
V
=
× I
Cn
o
N
V
(s) expression.
Cn
R
Substitution of Equation 31 into Equation 1 gives Equation 32:
R
sen
N
(EQ. 24)
(EQ. 25)
------------
V
(s) =
× I (s) × A
(s)
Rsen
Cn
o
2
sen
---- ------------
× I
o
I
=
×
(EQ. 32)
droop
R
N
1
i
----------------------
1 +
A
(s) =
Rsen
s
------------
Therefore:
2R
ω
sns
× I
sen
o
1
(EQ. 33)
---------------------------
=
R
---------------------------
ω
=
i
(EQ. 26)
N × I
Rsen
R
droop
sum
--------------
× C
n
N
Substitution of Equation 33 and application of the OCP condition
in Equation 29 gives Equation 34:
Transfer function A
(s) always has unity gain at DC.
Rsen
2R
× I
Current-sensing resistor R
value does not have significant
sen
omax
sen
variation over-temperature, so there is no need for the NTC
network. The recommended values are R = 1kΩ and
--------------------------------------
=
R
(EQ. 34)
i
N × I
droopmax
sum
where I
is the full load current and I
is the
omax
droopmax
C = 5600pF.
n
corresponding droop current. For example, given N = 3,
R
= 1mΩ, I
omax
i
= 51A and I
= 40.9µA, Equation 34
droopmax
sen
gives R = 831Ω.
January 8, 2013
FN7801.1
26
ISL6267
of the sensed output voltage, and then feeds it to the
Load Line Slope
See Figure 15 for load-line implementation.
compensator. T1 is measured after the summing node, and T2 is
measured in the voltage loop before the summing node. The
spreadsheet gives both T1(s) and T2(s) plots. However, only T2(s)
can actually be measured on an ISL6267 regulator.
For inductor DCR sensing, substitution of Equation 28 into
Equation 2 gives the load-line slope expression in Equation 35:
V
2R
R
DCR
---------------------- ----------------------------------------- -----------
droop
droop
R
ntcnet
-----------------
LL =
=
×
×
(EQ. 35)
VO
L
I
R
sum
N
o
i
--------------
+
R
ntcnet
Q1
N
iO
VIN
GATE
DRIVER
Q2
COUT
For resistor sensing, substitution of Equation 32 into Equation 2
gives the load line slope expression in Equation 36:
V
2R
× R
droop
sen droop
(EQ. 36)
-----------------
-----------------------------------------
=
LL =
I
N × R
o
i
LOAD LINE SLOPE
+
Substitution of Equation 29 and rewriting Equation 35, or
substitution of Equation 33 and rewriting Equation 36, gives the
same result as in Equation 37:
2Ω
+
-
EA
MOD.
+
COMP
VID
I
ISOLATION
TRANSFORMER
o
---------------
R
=
× LL
(EQ. 37)
droop
I
CHANNEL B
CHANNEL A
droop
LOOP GAIN =
One can use the full-load condition to calculate R
droop
. For
CHANNEL A
CHANNEL B
example, given I
= 51A, I
= 40.9µA and
omax
LL = 1.9mΩ, Equation 37 gives R
droopmax
NETWORK
ANALYZER
= 2.37kΩ.
droop
EXCITATION OUTPUT
It is recommended to start with the R
value calculated by
droop
FIGURE 28. LOOP GAIN T1(s) MEASUREMENT SET-UP
Equation 37 and fine-tune it on the actual board to get accurate
load-line slope. One should record the output voltage readings at
no load and at full load for load-line slope calculation. Reading
the output voltage at lighter load instead of full load will increase
the measurement error.
T1(s) is the total loop gain of the voltage loop and the droop loop.
It always has a higher crossover frequency than T2(s), therefore
has a higher impact on system stability.
T2(s) is the voltage loop gain with closed droop loop, thus having
a higher impact on output voltage response.
Compensator
Figure 21 shows the desired load transient response waveforms.
Figure 27 shows the equivalent circuit of a voltage regulator (VR)
with the droop function. A VR is equivalent to a voltage source
Design the compensator to get stable T1(s) and T2(s) with sufficient
phase margin and an output impedance equal to or smaller than
the load-line slope.
(= VID) and output impedance Z (s). If Z (s) is equal to the
out out
load-line slope LL, i.e., a constant output impedance, then in the
L
VO
entire frequency range, V will have a square response when I
has a square change.
o
o
Q1
IO
V
IN
GATE
Q2
CO
DRIVER
i
Zout(s) = LL
o
LOAD LINE SLOPE
Ω
+
20
VR
V
+
VID
LOAD
o
-
EA
MOD.
+
COMP
VID
ISOLATION
TRANSFORMER
CHANNEL B
CHANNEL A
LOOP GAIN =
FIGURE 27. VOLTAGE REGULATOR EQUIVALENT CIRCUIT
CHANNEL A
CHANNEL B
Intersil provides a Microsoft Excel-based spreadsheet to help
design the compensator and the current sensing network so that
VR achieves constant output impedance as a stable system.
NETWORK
ANALYZER
EXCITATION OUTPUT
FIGURE 29. LOOP GAIN T2(s) MEASUREMENT SET-UP
A VR with active droop function is a dual-loop system consisting of
a voltage loop and a droop loop, which is a current loop. However,
neither loop alone is sufficient to describe the entire system. The
spreadsheet shows two loop gain transfer functions, T1(s) and
T2(s), that describe the entire system. Figure 28 conceptually
shows T1(s) measurement set-up, and Figure 29 conceptually
shows T2(s) measurement set-up. The VR senses the inductor
current, multiplies it by a gain of the load-line slope, adds it on top
Current Balancing
Refer to Figures 16 through 20 for information on current
balancing. The ISL6267 achieves current balancing through
matching the ISEN pin voltages. R
and C
form filters to
isen
isen
remove the switching ripple of the phase node voltages. It is
recommended to use a rather long R
C
time constant such
isen isen
January 8, 2013
FN7801.1
27
ISL6267
that the ISEN voltages have minimal ripple and represent the DC
current flowing through the inductors. Recommended values are
R = 10kΩ and C = 0.22µF.
the state of VR_HOT, is defined in Equation 38.
0.88V
(EQ. 38)
---------------
= 14.7k
s
s
60μA
NTC Thermal Monitors and VR_HOT Function
The equivalent resistance at the typical reset threshold voltage of
0.92V required to change the state of VR_HOT back low, is
defined in Equation 39
The ISL6267 features three pins (NTC, NTC_NB, and VR_HOT)
which are allow the IC to monitor board temperature and alert
the AMD CPU of a thermal issue. Figure 30 shows the thermal
monitor feature of the ISL6267. An NTC network is connected
between the NTC and NTC_NB pins and GND. The controller
drives a 60µA current source out of the NTC pin and the NTC_NB
pin alternatively at 1kHz frequency with 50% duty cycle. The
pulsed current flows through the respective NTC resistor network
on the pins and creates a voltage that is compared to an
over-temperature trip threshold. If the voltage on both NTC pins is
higher than the over-temperature trip threshold, then VR_HOT is
pulled up by an external resistor on the pin.
0.92V
60μA
(EQ. 39)
---------------
= 15.3k
The NTC thermistor value correlates this resistance change to the
required temperature hysteresis. A standard 1% resistor is
typically needed to meet the NTC pin threshold voltage.
For example, a Panasonic NTC thermistor with B = 4700 has a
resistance ratio of 0.03322 of its nominal value at +105°C and
0.03956 of its nominal value at +100°C. The required resistance
of the NTC is defined in Equation 40.
(15.3kΩ – 14.7kΩ)
(0.03956 – 0.03322)
(EQ. 40)
-------------------------------------------------------
= 94.6kΩ
NTC
+V
The closest, larger thermistor value for B = 4700 is 100kΩ. The
R
p
+
60µA
NTC thermistor part number is ERTJ1VV104.
V
R
R
NTC
NTC
VR_HOT
-
At +105°C, a 100kΩ NTC resistance drops to
SW1
SW2
(0.03322 x 100kΩ) = 3.322kΩ. With a 60µA current flowing out
of the NTC pin, the voltage drop across the resistor is only
(3.322kΩ x 60µA) = 0.199V. This value is much lower than the
threshold voltage of 0.88V. A standard resistor, 1% tolerance,
added in series with the thermistor is required to raise the
voltage on the pin. The resistance required to meet the trip
threshold is calculated in Equation 41.
R
s
MONITOR
SW1 SW2
NTC_NB
+
R
p
INTERNAL TO
ISL6267
V
R
NTC
NTC
0.88V
60μA
-
(EQ. 41)
---------------
– 3.322kΩ = 11.34kΩ
R
s
The closest, standard 1% tolerance resistor is 11.3kΩ.
The NTC thermistor is placed in a hot spot on the board, typically
near the upper MOSFET of channel 1 of the respective output.
The standard resistor is placed next to the controller.
FIGURE 30. CIRCUITRY ASSOCIATED WITH THE THERMAL
MONITOR FEATURE OF THE ISL6267
As the board temperature rises, the NTC thermistor resistance
decreases and the voltage at the NTC pin drops. When the
voltage on the NTC pin drops below the over-temperature trip
threshold, then VR_HOT is pulled low. The VR_HOT signal is used
to change the CPU operation and decrease power consumption.
With the reduction in power consumption by the CPU, the board
temperature decreases and the NTC thermistor voltage rises.
Once the over-temperature threshold is tripped and VR_HOT is
taken low, the over-temperature threshold changes to the reset
level. The addition of hysteresis to the over-temperature
threshold prevents nuisance trips. Once both pin voltages exceed
the over-temperature reset threshold, the pull-down on VR_HOT
is released. The signal changes state and the CPU resumes
normal operation. The over-temperature threshold returns to the
trip level.
Selection of the NTC components can vary depending on how the
resistor network is configured. The equivalent resistance at the
typical over-temperature threshold voltage of 0.88V, to change
January 8, 2013
FN7801.1
28
ISL6267
Layout Guidelines
Table 9 shows layout considerations for the ISL6267 controller. Refer to the reference designators shown in Figure 31.
TABLE 9. LAYOUT CONSIDERATIONS FOR THE ISL6267 CONTROLLER
ISL6267
SYMBOL
GND
LAYOUT GUIDELINES
BOTTOM PAD
Create analog ground plane underneath the controller and the analog signal processing components. Do not let
the power ground plane overlap with the analog ground plane. Avoid allowing noisy planes/traces (e.g., phase
node) to crossover/overlap the analog plane.
1
2
FB2_NB
FB_NB
Place the compensator components (R25, R9, R24, C88, C51, C86, and C153) close to the controller.
3
COMP_NB
VW_NB
4
Place the capacitor (C85) across VW, and place COMP close to the controller.
No special consideration.
5
PGOOD_NB
6, 7, 8
9
SVD, PWROK, SVC Use good signal integrity practices.
ENABLE
PGOOD
VR_HOT
NTC
No special consideration.
No special consideration.
No special consideration.
10
11
12
Place the NTC thermistor (R46) close to the thermal source that is monitored to determine CPU V
throttling. Usually it is placed close to Core VR phase-1 high-side MOSFET.
thermal
CORE
13
14
15
16
VW
COMP
FB
Place the capacitor (C4) across VW and COMP close to the controller.
Place the compensator components (R7, R10, R11, C3, C6, C11 and C5) in general proximity to the controller.
FB2
ISEN3
ISEN2
ISEN1
Each ISEN pin has a capacitor (C
GND. Place Cisen capacitors as close as possible to the controller and keep the following loops small:
1. Any ISEN pin to another ISEN pin
) decoupling it to VSUMN and then through another capacitor (C
) to
vsumn
isen
17
18
2. Any ISEN pin to GND
The red traces in the following drawing show the loops to be minimized.
Phase1
L3
Risen
Ro
Ro
Ro
ISEN3
ISEN2
Cisen
Cisen
Cisen
V
o
Phase2
Risen
L2
L1
Phase3
Risen
ISEN1
GND
Vsumn
Cvsumn
19
20
VSEN
RTN
Place the VSEN/RTN filter (C12, C13) close to the controller for good decoupling.
January 8, 2013
FN7801.1
29
ISL6267
TABLE 9. LAYOUT CONSIDERATIONS FOR THE ISL6267 CONTROLLER (Continued)
LAYOUT GUIDELINES
ISL6267
21
SYMBOL
ISUMN
ISUMP
Place the current sensing circuit in general proximity of the controller.
Place capacitor Cn very close to the controller.
Place the NTC thermistor next to VR1 phase-1 inductor (L1) so it senses the inductor temperature correctly.
Each phase of the power stage sends a pair of VSUMP and VSUMN signals to the controller. Run these two
signals traces in parallel fashion with decent width (>20mil).
22
IMPORTANT: Sense the inductor current by routing the sensing circuit to the inductor pads. Route R63 and R71
to Core VR phase-1 side pad of inductor L1. Route R88 to the output side pad of inductor L1. Route R65 and
R72 to Core VR phase-2 side pad of inductor L2. Route R90 to the output side pad of inductor L2. If possible,
route the traces on a different layer from the inductor pad layer and use vias to connect the traces to the center
of the pads. If no via is allowed on the pad, consider routing the traces into the pads from the inside of the
inductor. The following drawings show the two preferred ways of routing current sensing traces.
INDUCTOR
INDUCTOR
VIAS
CURRENT-SENSING TRACES
CURRENT-SENSING TRACES
23
24
25
26
27
28
VDD
VIN
A capacitor (C16) decouples it to GND. Place it in close proximity to the controller.
A capacitor (C17) decouples it to GND. Place it in close proximity to the controller.
No special consideration.
PROG1
BOOT1
UGATE1
PHASE1
Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close.
Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from
crossing over or getting close. Recommend routing PHASE1 trace to VR1 phase-1 high-side MOSFET (Q2 and
Q8) source pins instead of general copper.
29
30
31
32
33
34
LGATE1
PWM3
VCCP
Use decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close.
No special consideration.
A capacitor (C22) decouples it to GND. Place it in close proximity to the controller.
Use decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close.
LGATE2
PHASE2
UGATE2
Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from
crossing over or getting close. Recommend routing PHASE2 trace to VR1 phase-2 high-side MOSFET (Q4 and
Q10) source pins instead of general copper.
35
36
37
38
39
BOOT2
Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close.
No special consideration.
PWM2_NB
LGATE1_NB
PHASE1_NB
UGATE1_NB
Use decent width (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close.
Run these two traces in parallel fashion with decent width (>30mil). Avoid any sensitive analog signal trace from
crossing over or getting close. Recommend routing PHASE1G trace to VR2 phase-1 high-side MOSFET source
pins instead of general copper.
40
41
42
BOOT1_NB
PROG2
Use decent wide trace (>30mil). Avoid any sensitive analog signal trace from crossing over or getting close.
No special consideration.
NTC_NB
Place the NTC thermistor close to the thermal source that is monitored to determine GT V
thermal
CORE
throttling. Usually it is placed close to Northbridge VR phase-1 high-side MOSFET.
43
44
ISUMN_NB
ISUMP_NB
Place the current sensing circuit in general proximity to the controller.
Place capacitor Cn very close to the controller.
Place the NTC thermistor next to Northbridge VR phase-1 inductor (L1) so it senses the inductor temperature
correctly.
See ISUMN and ISUMP pins for layout guidelines of current-sensing trace routing.
January 8, 2013
FN7801.1
30
ISL6267
TABLE 9. LAYOUT CONSIDERATIONS FOR THE ISL6267 CONTROLLER (Continued)
LAYOUT GUIDELINES
ISL6267
45
SYMBOL
RTN_NB
Place the VSEN/RTN filter (C89, C90) in close proximity to the controller for good decoupling.
46
VSEN_NB
ISEN2_NB
ISEN1_NB
47
See ISEN1, ISEN2 and ISEN3 pins for layout guidelines of current-balancing circuit trace routing.
48
FIGURE 31. PORTION OF ISL6267EVAL1Z EVALUATION BOARD SCHEMATIC
January 8, 2013
FN7801.1
31
ISL6267
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make
sure you have the latest revision.
DATE
REVISION
FN7801.0
FN7801.1
CHANGE
1/31/11
Initial Release.
Page 9: Added IRZ parts to ordering information table.
November 5, 2012
About Intersil
Intersil Corporation is a leader in the design and manufacture of high-performance analog, mixed-signal and power management
semiconductors. The company's products address some of the fastest growing markets within the industrial and infrastructure,
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our winning team, visit our website and career page at www.intersil.com.
For a complete listing of Applications, Related Documentation and Related Parts, please see the respective product information page.
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Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time
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January 8, 2013
32
FN7801.1
ISL6267
Package Outline Drawing
L48.6x6B
48 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 0, 9/09
4X
4.4
6.00
0.40
44X
A
6
B
PIN #1 INDEX AREA
48
37
6
1
36
PIN 1
INDEX AREA
4 .40 ± 0.15
25
12
0.15
(4X)
13
24
0.10 M C A B
0.05 M C
TOP VIEW
48X 0.45 ± 0.10
BOTTOM VIEW
4
48X 0.20
SEE DETAIL "X"
C
0.10
C
MAX 1.00
BASE PLANE
SEATING PLANE
0.08
( 44 X 0 . 40 )
( 5. 75 TYP )
(
C
SIDE VIEW
4. 40 )
5
0 . 2 REF
C
( 48X 0 . 20 )
0 . 00 MIN.
0 . 05 MAX.
( 48X 0 . 65 )
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3.
Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
Tiebar shown (if present) is a non-functional feature.
5.
The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
6.
January 8, 2013
FN7801.1
33
相关型号:
ISL6269AIRZ-T
25A SWITCHING CONTROLLER, 600kHz SWITCHING FREQ-MAX, PQCC16, 4 X 4 MM,ROHS COMPLIANT, MO-220, PLASTIC, QFN-16
RENESAS
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