LMC6001 [NSC]
Ultra Ultra-Low Input Current Amplifier; 超超低输入电流放大器
| 型号： | LMC6001 |
| 厂家： | 
National Semiconductor |
| 描述： | Ultra Ultra-Low Input Current Amplifier |
| 文件： | 总14页 (文件大小：308K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
March 1995
LMC6001
Ultra Ultra-Low Input Current Amplifier
can achieve higher signal to noise ratio than JFET input type
electrometer amplifiers. Other applications of the LMC6001
include long interval integrators, ultra-high input impedance
instrumentation amplifiers, and sensitive electrical-field mea-
surement circuits.
General Description
Featuring 100% tested input currents of 25 fA max., low op-
erating power, and ESD protection of 2000V, the LMC6001
achieves a new industry benchmark for low input current op-
erational amplifiers. By tightly controlling the molding com-
pound, National is able to offer this ultra-low input current in
a lower cost molded package.
Features
(Max limit, 25˚C unless otherwise noted)
n Input current (100% tested): 25 fA
n Input current over temp.: 2 pA
n Low power: 750 µA
To avoid long turn-on settling times common in other low in-
put current opamps, the LMC6001A is tested 3 times in the
first minute of operation. Even units that meet the 25 fA limit
are rejected if they drift.
√
n Low VOS
:
350 µV
Because of the ultra-low input current noise of 0.13 fA/ Hz,
the LMC6001 can provide almost noiseless amplification of
high resistance signal sources. Adding only 1 dB at 100 kΩ,
0.1 dB at 1 MΩ and 0.01 dB or less from 10 MΩ to 2,000 MΩ,
the LMC6001 is an almost noiseless amplifier.
√
@
n Low noise: 22 nV/ Hz 1 kHz Typ.
Applications
n Electrometer amplifier
n Photodiode preamplifier
n Ion detector
The LMC6001 is ideally suited for electrometer applications
requiring ultra-low input leakage such as sensitive photode-
tection transimpedance amplifiers and sensor amplifiers.
n A.T.E. leakage testing
√
Since input referred noise is only 22 nV/ Hz, the LMC6001
Connection Diagrams
8-Pin DIP
8-Pin Metal Can
DS011887-1
Top View
DS011887-2
Top View
© 1999 National Semiconductor Corporation
DS011887
www.national.com
Ordering Information
Package
Industrial Temperature Range
−40˚C to +85˚C
NSC Package
Drawing
N08E
8-Pin
LMC6001AIN, LMC6001BIN,
LMC6001CIN
Molded DIP
8-Pin
LMC6001AIH, LMC6001BIH
H08C
Metal Can
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2
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Current at Power Supply Pin
Power Dissipation
40 mA
(Note 9)
2 kV
ESD Tolerance (Note 9)
Operating Ratings (Note 1)
±
Differential Input Voltage
Voltage at Input/Output Pin
Supply Voltage (V+ − V−)
Output Short Circuit to V+
Output Short Circuit to V−
Lead Temperature
Supply Voltage
(V+) + 0.3V, (V−) − 0.3V
−0.3V to +16V
(Notes 2, 10)
Temperature Range
LMC6001AI, LMC6001BI, LMC6001CI
−40˚C ≤ TJ ≤ +85˚C
Supply Voltage
4.5V ≤ V+ ≤ 15.5V
(Note 2)
Thermal Resistance (Note 11)
θJA, N Package
(Soldering, 10 Sec.)
260˚C
−65˚C to +150˚C
150˚C
100˚C/W
145˚C/W
45˚C/W
Storage Temperature
Junction Temperature
Current at Input Pin
θJA, H Package
θJC, H Package
±
±
10 mA
30 mA
Power Dissipation
(Note 8)
Current at Output Pin
DC Electrical Characteristics
=
Limits in standard typeface guaranteed for TJ 25˚C and limits in boldface type apply at the temperature extremes. Unless
−
otherwise specified, V+ 5V, V
=
=
=
>
0V, VCM 1.5V, and RL 1M.
Symbol
Parameter
Input Current
Conditions
Limits (Note 5)
Units
Typical
(Note 4)
LMC6001AI LMC6001BI LMC6001CI
=
IB
Either Input, VCM 0V,
10
5
25
2000
1000
0.35
1.0
100
4000
2000
1.0
1000
4000
2000
1.0
=
±
VS
5V
fA
IOS
Input Offset Current
Input Offset Voltage
VOS
1.7
2.0
mV
=
=
±
VS
5V, VCM 0V
0.7
1.35
2.0
1.35
1.35
10
TCVOS
Input Offset
2.5
10
µV/˚C
Voltage Drift
>
RIN
Input Resistance
Common Mode
Rejection Ratio
Positive Power Supply
Rejection Ratio
Negative Power
Supply Rejection Ratio
Large Signal
1
Tera Ω
CMRR
0V ≤ VCM ≤ 7.5V
83
75
72
72
68
66
63
V+ 10V
=
+PSRR
−PSRR
AV
5V ≤ V+ ≤ 15V
0V ≥ V− ≥ −10V
83
73
66
66
dB
min
70
63
63
94
80
74
74
77
71
71
=
Sourcing, RL 2 kΩ
1400
350
400
300
200
90
300
200
90
Voltage Gain
(Note 6)
300
V/mV
min
=
Sinking, RL 2 kΩ
180
(Note 6)
100
60
60
V+ 5V and 15V
−0.4
V+ − 1.9
4.87
0.10
14.63
0.26
−0.1
0
−0.1
0
−0.1
0
V
max
V
=
VCM
Input Common-Mode
Voltage
For CMRR ≥ 60 dB
V+ − 2.3
V+ − 2.5
4.80
4.73
0.14
0.17
14.50
14.34
0.35
0.45
V+ − 2.3
V+ − 2.5
4.75
4.67
0.20
0.24
14.37
14.25
0.44
0.56
V+ − 2.3
V+ − 2.5
4.75
4.67
0.20
0.24
14.37
14.25
0.44
0.56
min
V
VO
Output Swing
V+ 5V
=
=
RL 2 kΩ to 2.5V
min
V
max
V
V+ 15V
=
=
RL 2 kΩ to 7.5V
min
V
max
3
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DC Electrical Characteristics (Continued)
=
Limits in standard typeface guaranteed for TJ 25˚C and limits in boldface type apply at the temperature extremes. Unless
−
otherwise specified, V+ 5V, V
=
=
=
>
0V, VCM 1.5V, and RL 1M.
Symbol
Parameter
Conditions
Limits (Note 5)
Units
Typical
(Note 4)
LMC6001AI LMC6001BI LMC6001CI
IO
Output Current
Sourcing, V+ 5V,
=
22
21
16
10
13
8
13
8
=
VO 0V
Sinking, V+ 5V,
=
16
13
13
=
VO 5V
Sourcing, V+ 15V,
13
10
10
mA
min
=
30
28
23
23
=
VO 0V
Sinking, V+ 15V,
22
18
18
=
34
28
23
23
=
VO 13V (Note 10)
22
18
18
V+ 5V, VO 1.5V
450
550
750
900
850
950
750
900
850
950
750
900
850
950
=
=
IS
Supply Current
µA
max
V+ 15V, VO 7.5V
=
=
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4
AC Electrical Characteristics
=
Limits in standard typeface guaranteed for TJ 25˚C and limits in boldface type apply at the temperature extremes. Unless
−
otherwise specified, V+ 5V, V
0V, VCM 1.5V and RL 1M.
=
=
=
>
Symbol Parameter
Conditions
Typical
Limits (Note 5)
Units
(Note 4) LM6001AI LM6001BI LM6001CI
SR
Slew Rate
(Note 7)
1.5
0.8
0.8
0.8
V/µs
min
MHz
Deg
dB
0.6
0.6
0.6
GBW
φfm
GM
en
Gain-Bandwidth Product
Phase Margin
1.3
50
Gain Margin
17
=
√
nV/ Hz
Input-Referred Voltage Noise
Input-Referred Current Noise
Total Harmonic Distortion
F
F
F
1 kHz
1 kHz
22
=
=
√
in
0.13
0.01
fA/ Hz
=
10 kHz, AV −10,
THD
=
RL 100 kΩ,
%
=
VO 8 VPP
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is in-
tended to be functional but do not guarantee specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics. The
guaranteed specifications apply only for the test conditions listed.
Note 2: Applies to both single supply and split supply operation. Continuous short circuit operation at elevated ambient temperature can result in exceeding the maxi-
±
mum allowed junction temperature of 150˚C. Output currents in excess of 30 mA over long term may adversely affect reliability.
Note 3: The maximum power dissipation is function of and The maximum allowable power dissipation at any ambient temperature is
a
T
,
θ
,
JA
T .
A
J(max)
=
P
(T
− T )/θ .
J(max) A JA
D
Note 4: Typical values represent the most likely parametric norm.
Note 5: All limits are guaranteed by testing or statistical analysis.
+
=
=
=
7.5V and R connected to 7.5V. For Sourcing tests, 7.5V ≤ V ≤ 11.5V. For Sinking tests, 2.5V ≤ V ≤ 7.5V.
Note 6:
Note 7:
V
V
15V, V
CM
L
O
O
+
15V. Connected as Voltage Follower with 10V step input. Limit specified is the lower of the positive and negative slew rates.
=
Note 8: For operating at elevated temperatures the device must be derated based on the thermal resistance θ with P
(T − T )/θ .
J A JA
JA
D
Note 9: Human body model, 1.5 kΩ in series with 100 pF.
+
+
Note 10: Do not connect the output to V , when V is greater than 13V or reliability will be adversely affected.
Note 11: All numbers apply for packages soldered directly into a printed circuit board.
5
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=
=
±
Typical Performance Characteristics VS
7.5V, TA 25˚C, unless otherwise specified
Input Current
Input Current
Supply Current
=
vs Temperature
±
5V
vs VCM VS
vs Supply Voltage
DS011887-18
DS011887-21
DS011887-24
DS011887-16
DS011887-19
DS011887-22
DS011887-17
Input Voltage
vs Output Voltage
Common Mode Rejection
Ratio vs Frequency
Power Supply Rejection
Ratio vs Frequency
DS011887-20
Input Voltage Noise
vs Frequency
Noise Figure
vs Source Resistance
Output Characteristics
Sourcing Current
DS011887-23
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6
=
=
7.5V, TA 25˚C, unless otherwise
±
Typical Performance Characteristics VS
specified (Continued)
Output Characteristics
Sinking Current
Gain and Phase Response
vs Temperature
(−55˚C to +125˚C)
Gain and Phase
Response vs Capacitive Load
=
with RL 500 kΩ
DS011887-25
DS011887-26
DS011887-27
Open Loop
Frequency Response
Inverting Small Signal
Pulse Response
Inverting Large Signal
Pulse Response
DS011887-29
DS011887-30
DS011887-28
Non-Inverting Small
Signal Pulse Response
Non-Inverting Large
Signal Pulse Response
Stability vs
Capacitive Load
DS011887-31
DS011887-32
DS011887-33
Applications Hints
AMPLIFIER TOPOLOGY
op-amps. These features make the LMC6001 both easier to
design with, and provide higher speed than products typi-
cally found in this low power class.
The LMC6001 incorporates a novel op-amp design topology
that enables it to maintain rail-to-rail output swing even when
driving a large load. Instead of relying on a push-pull unity
gain output buffer stage, the output stage is taken directly
from the internal integrator, which provides both low output
impedance and large gain. Special feed-forward compensa-
tion design techniques are incorporated to maintain stability
over a wider range of operating conditions than traditional
COMPENSATING FOR INPUT CAPACITANCE
It is quite common to use large values of feedback resis-
tance for amplifiers with ultra-low input current, like the
LMC6001.
7
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Applications Hints (Continued)
Although the LMC6001 is highly stable over a wide range of
operating conditions, certain precautions must be met to
achieve the desired pulse response when a large feedback
resistor is used. Large feedback resistors with even small
values of input capacitance, due to transducers, photo-
diodes, and circuit board parasitics, reduce phase margins.
When high input impedances are demanded, guarding of the
LMC6001 is suggested. Guarding input lines will not only re-
duce leakage, but lowers stray input capacitance as well.
(See Printed-Circuit-Board Layout for High Impedance
Work).
The effect of input capacitance can be compensated for by
adding a capacitor, Cf, around the feedback resistors (as in
Figure 1 ) such that:
DS011887-6
FIGURE 2. LMC6001 Noninverting Gain of 10 Amplifier,
Compensated to Handle Capacitive Loads
or
R1 CIN ≤ R2 Cf
Since it is often difficult to know the exact value of CIN, Cf can
be experimentally adjusted so that the desired pulse re-
sponse is achieved. Refer to the LMC660 and LMC662 for a
more detailed discussion on compensating for input
capacitance.
In the circuit of Figure 2, R1 and C1 serve to counteract the
loss of phase margin by feeding the high frequency compo-
nent of the output signal back to the amplifier’s inverting in-
put, thereby preserving phase margin in the overall feedback
loop.
Capacitive load driving capability is enhanced by using a pul-
lup resistor to V+ (Figure 3). Typically a pullup resistor con-
ducting 500 µA or more will significantly improve capacitive
load responses. The value of the pullup resistor must be de-
termined based on the current sinking capability of the ampli-
fier with respect to the desired output swing. Open loop gain
of the amplifier can also be affected by the pullup resistor
(see Electrical Characteristics).
DS011887-5
FIGURE 1. Cancelling the Effect of Input Capacitance
CAPACITIVE LOAD TOLERANCE
All rail-to-rail output swing operational amplifiers have volt-
age gain in the output stage. A compensation capacitor is
normally included in this integrator stage. The frequency lo-
cation of the dominant pole is affected by the resistive load
on the amplifier. Capacitive load driving capability can be op-
timized by using an appropriate resistive load in parallel with
the capacitive load (see Typical Curves).
DS011887-7
FIGURE 3. Compensating for Large Capacitive
Loads with a Pullup Resistor
PRINTED-CIRCUIT-BOARD LAYOUT
FOR HIGH-IMPEDANCE WORK
Direct capacitive loading will reduce the phase margin of
many op-amps. A pole in the feedback loop is created by the
combination of the op-amp’s output impedance and the ca-
pacitive load. This pole induces phase lag at the unity-gain
crossover frequency of the amplifier resulting in either an os-
cillatory or underdamped pulse response. With a few exter-
nal components, op amps can easily indirectly drive capaci-
tive loads, as shown in Figure 2.
It is generally recognized that any circuit which must operate
with less than 1000 pA of leakage current requires special
layout of the PC board. When one wishes to take advantage
of the ultra-low bias current of the LMC6001, typically less
than 10 fA, it is essential to have an excellent layout. Fortu-
nately, the techniques of obtaining low leakages are quite
simple. First, the user must not ignore the surface leakage of
the PC board, even though it may sometimes appear accept-
ably low, because under conditions of high humidity or dust
or contamination, the surface leakage will be appreciable.
To minimize the effect of any surface leakage, lay out a ring
of foil completely surrounding the LMC6001’s inputs and the
terminals of capacitors, diodes, conductors, resistors, relay
terminals, etc., connected to the op-amp’s inputs, as in Fig-
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8
The designer should be aware that when it is inappropriate
to lay out a PC board for the sake of just a few circuits, there
is another technique which is even better than a guard ring
on a PC board: Don’t insert the amplifier’s input pin into the
board at all, but bend it up in the air and use only air as an in-
sulator. Air is an excellent insulator. In this case you may
have to forego some of the advantages of PC board con-
struction, but the advantages are sometimes well worth the
effort of using point-to-point up-in-the-air wiring. See Figure
6.
Applications Hints (Continued)
ure 4. To have a significant effect, guard rings should be
placed on both the top and bottom of the PC board. This PC
foil must then be connected to a voltage which is at the same
voltage as the amplifier inputs, since no leakage current can
flow between two points at the same potential. For example,
a PC board trace-to-pad resistance of 1012Ω, which is nor-
mally considered a very large resistance, could leak 5 pA if
the trace were a 5V bus adjacent to the pad of the input.
This would cause
a 500 times degradation from the
LMC6001’s actual performance. If a guard ring is used and
held within 1 mV of the inputs, then the same resistance of
1012Ω will only cause 10 fA of leakage current. Even this
small amount of leakage will degrade the extremely low input
current performance of the LMC6001. See Figure 5 for typi-
cal connections of guard rings for standard op-amp
configurations.
DS011887-12
(Input pins are lifted out of PC board and soldered directly to components.
All other pins connected to PC board).
FIGURE 6. Air Wiring
Another potential source of leakage that might be over-
looked is the device package. When the LMC6001 is manu-
factured, the device is always handled with conductive finger
cots. This is to assure that salts and skin oils do not cause
leakage paths on the surface of the package. We recom-
mend that these same precautions be adhered to, during all
phases of inspection, test and assembly.
DS011887-8
FIGURE 4. Examples of Guard
Ring in PC Board Layout
Latchup
CMOS devices tend to be susceptible to latchup due to their
internal parasitic SCR effects. The (I/O) input and output pins
look similar to the gate of the SCR. There is a minimum cur-
rent required to trigger the SCR gate lead. The LMC6001 is
designed to withstand 100 mA surge current on the I/O pins.
Some resistive method should be used to isolate any capaci-
tance from supplying excess current to the I/O pins. In addi-
tion, like an SCR, there is a minimum holding current for any
latchup mode. Limiting current to the supply pins will also in-
hibit latchup susceptibility.
Typical Applications
DS011887-9
The extremely high input resistance, and low power con-
sumption, of the LMC6001 make it ideal for applications that
require battery-powered instrumentation amplifiers. Ex-
amples of these types of applications are hand-held pH
probes, analytic medical instruments, electrostatic field de-
tectors and gas chromotographs.
Inverting Amplifier
Two Opamp, Temperature
DS011887-10
Compensated pH Probe Amplifier
Non-Inverting Amplifier
The signal from a pH probe has a typical resistance between
10 MΩ and 1000 MΩ. Because of this high value, it is very
important that the amplifier input currents be as small as
possible. The LMC6001 with less than 25 fA input current is
an ideal choice for this application.
The theoretical output of the standard Ag/AgCl pH probe is
59.16 mV/pH at 25˚C with 0V out at a pH of 7.00. This output
is proportional to absolute temperature. To compensate for
this, a temperature compensating resistor, R1, is placed in
DS011887-11
Follower
FIGURE 5. Typical Connections of Guard Rings
9
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1. The LMC6001A guarantees a 25 fA limit on input current
at 25˚C.
Two Opamp, Temperature
Compensated pH Probe Amplifier
2. The input ESD protection diodes in the LMC6042 are
only rated at 500V while the LMC6001 has much more
robust protection that is rated at 2000V.
(Continued)
the feedback loop. This cancels the temperature depen-
dence of the probe. This resistor must be mounted where it
will be at the same temperature as the liquid being mea-
sured.
The setup and calibration is simple with no interactions to
cause problems.
1. Disconnect the pH probe and with R3 set to about
mid-range and the noninverting input of the LMC6001
grounded, adjust R8 until the output is 700 mV.
The LMC6001 amplifies the probe output providing a scaled
±
voltage of 100 mV/pH from a pH of 7. The second opamp,
a micropower LMC6041 provides phase inversion and offset
so that the output is directly proportional to pH, over the full
range of the probe. The pH reading can now be directly dis-
played on a low cost, low power digital panel meter. Total
current consumption will be about 1 mA for the whole sys-
tem.
2. Apply −414.1 mV to the noninverting input of the
LMC6001. Adjust R3 for and output of 1400 mV. This
completes the calibration. As real pH probes may not
perform exactly to theory, minor gain and offset adjust-
ments should be made by trimming while measuring a
precision buffer solution.
The micropower dual operational amplifier, LMC6042, would
optimize power consumption but not offer these advantages:
DS011887-15
R1 100k + 3500 ppm/˚C (Note 12)
R2 68.1k
R3, 8 5k
R4, 9 100k
R5 36.5k
R6 619k
R7 97.6k
D1 LM4040D1Z-2.5
C1 2.2 µF
Note 12: (Micro-ohm style 144 or similar)
FIGURE 7. pH Probe Amplifier
Ultra-Low Input Current Instrumentation Amplifier
Figure 8 shows an instrumentation amplifier that features
R2 provides a simple means of adjusting gain over a wide
high differential and common mode input resistance
range without degrading CMRR. R7 is an initial trim used to
maximize CMRR without using super precision matched re-
sistors. For good CMRR over temperature, low drift resistors
should be used.
(
1014Ω), 0.01% gain accuracy at AV
=
1000, excellent
CMRR with 1 MΩ imbalance in source resistance. Input cur-
>
rent is less than 20 fA and offset drift is less than 2.5 µV/˚C.
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10
Ultra-Low Input Current Instrumentation Amplifier (Continued)
DS011887-13
=
=
=
R , and R R ; then
4
If R
1
R , R
5
3
6
7
=
9.85k).
A
V
≈ 100 for circuit shown (R
2
FIGURE 8. Instrumentation Amplifier
11
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12
Physical Dimensions inches (millimeters) unless otherwise noted
8-Pin Metal Can Package (H)
Order Number LMC6001AIH or LMC6001BIH
NS Package Number H08C
8-Pin Molded Dual-In-Line Package
Order Number LMC6001AIN, LMC6001BIN or LMC6001CIN
NS Package Number N08E
13
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Physical Dimensions inches (millimeters) unless otherwise noted (Continued)
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2. A critical component is any component of a life support
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