MAX16818ATI+ [MAXIM]
1.5MHz, 30A High-Efficiency, LED Driver with Rapid LED Current Pulsing; 为1.5MHz ， 30A高效的LED驱动器，可快速响应LED脉动电流
| 型号： | MAX16818ATI+ |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | 1.5MHz, 30A High-Efficiency, LED Driver with Rapid LED Current Pulsing |
| 文件： | 总26页 (文件大小：917K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
19-0666; Rev 0; 10/06
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
General Description
Features
o High-Current LED Driver Controller IC, Up to 30A
The MAX16818 pulse-width modulation (PWM) LED dri-
ver controller provides high-output-current capability in
a compact package with a minimum number of external
components. The MAX16818 is suitable for use in syn-
chronous and nonsynchronous step-down (buck)
topologies, as well as in boost, buck-boost, SEPIC, and
Cuk LED drivers. The MAX16818 is the first LED driver
controller that enables Maxim’s patent-pending technol-
ogy for fast LED current transients of up to 20A/µs and
30kHz dimming frequency.
Output Current
o Average-Current-Mode Control
o True-Differential Remote-Sense Input
o 4.75V to 5.5V or 7V to 28V Input Voltage Range
o Programmable Switching Frequency or External
Synchronization from 125kHz to 1.5MHz
o Clock Output for 180° Out-of-Phase Operation
o Integrated 4A Gate Drivers
This device utilizes average-current-mode control that
enables optimal use of MOSFETs with optimal charge
and on-resistance characteristics. This results in the
minimized need for external heatsinking even when
delivering up to 30A of LED current. True differential
sensing enables accurate control of the LED current. A
wide dimming range is easily implemented to accom-
modate an external PWM signal. An internal regulator
enables operation over a wide input voltage range:
4.75V to 5.5V or 7V to 28V and above with a simple
external biasing device. The wide switching frequency
range, up to 1.5MHz, allows for the use of small induc-
tors and capacitors.
o Output Overvoltage and Hiccup Mode
Overcurrent Protection
o Thermal Shutdown
o Thermally Enhanced 28-Pin Thin QFN Package
o -40°C to +125°C Operating Temperature Range
Ordering Information
PIN-
PACKAGE
PKG
CODE
PART
TEMP RANGE
The MAX16818 features a clock output with 180° phase
delay to control a second out-of-phase LED driver to
reduce input and output filter capacitors size or to mini-
mize ripple currents. The MAX16818 offers programma-
ble hiccup, overvoltage, and overtemperature protection.
MAX16818ATI+
MAX16818ETI+
-40°C to +125°C 28 TQFN-EP* T2855-3
-40°C to +85°C
28 TQFN-EP* T2855-3
+Denotes lead-free package.
*EP = Exposed paddle.
The MAX16818ETI+ is rated for the extended tempera-
ture range (-40°C to +85°C) and the MAX16818ATI+ is
rated for the automotive temperature range (-40°C to
+125°C). This LED driver controller is available in a
lead-free, 0.8mm high, 5mm x 5mm 28-pin TQFN pack-
age with exposed paddle.
Simplified Diagram
7V TO 28V
C1
L1
IN
Q1
EN
Applications
Front Projectors/Rear Projection TVs
DH
V
LED
ILIM
Portable and Pocket Projectors
MAX16818
Automotive, Bus/Truck Exterior Lighting
LCD TVs and Display Backlight
Q2
DL
C2
Q3
Automotive Emergency Lighting and Signage
OVI
CSP
R1
PGND
CLP
.
HIGH-FREQUENCY
PULSE TRAIN
Typical Operating Circuit and Pin Configuration located at
end of data sheet.
NOTE: MAXIM PATENT-PENDING TOPOLOGY
________________________________________________________________ Maxim Integrated Products
1
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
ABSOLUTE MAXIMUM RATINGS
IN to SGND.............................................................-0.3V to +30V
BST to SGND..........................................................-0.3V to +35V
BST to LX..................................................................-0.3V to +6V
Continuous Power Dissipation (T = +70°C)
A
28-Pin TQFN (derate 34.5mW/°C above +70°C) .......2758mW
Operating Temperature Range
DH to LX.......................................-0.3V to [(V
DL to PGND................................................-0.3V to (V
- V ) + 0.3V]
MAX16818ATI+..............................................-40°C to +125°C
MAX16818ETI+................................................-40°C to +85°C
Maximum Junction Temperature .....................................+150°C
Storage Temperature Range.............................-60°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
BST
LX_
+ 0.3V)
DD
V
V
to SGND............................................................-0.3V to +6V
CC
CC
, V
to PGND...................................................-0.3V to +6V
DD
SGND to PGND .....................................................-0.3V to +0.3V
All Other Pins to SGND...............................-0.3V to (V + 0.3V)
CC
MAX618
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V
= 5V, V
= V , T = T = T
to T
, unless otherwise noted. Typical specifications are at T = +25°C.) (Note 1)
MAX A
CC
DD
CC
A
J
MIN
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
SYSTEM SPECIFICATIONS
7
28
5.50
5.5
Input Voltage Range
V
V
IN
Short IN and V
operation
together for 5V input
CC
4.75
Quiescent Supply Current
I
EN = V
or SGND, not switching
2.7
mA
Q
CC
LED CURRENT REGULATOR
No load, V = 4.75V to 5.5V, f
= 500kHz
SW
0.594
0.594
0.6
0.6
0.606
0.606
IN
SENSE+ to SENSE- Accuracy
(Note 2)
V
No load, V = 7V to 28V, f
= 500kHz
IN
SW
Clock
Cycles
Soft-Start Time
t
SS
1024
STARTUP/INTERNAL REGULATOR
V
V
V
Undervoltage Lockout
Undervoltage Hysteresis
Output Voltage
UVLO
V
V
rising
CC
4.1
4.3
200
5.1
4.5
5.30
3.0
V
mV
V
CC
CC
CC
= 7V to 28V, I
= 0 to 60mA
4.85
IN
SOURCE
MOSFET DRIVERS
Output Driver Impedance
Output Driver Source/Sink Current
Nonoverlap Time
R
Low or high output, I
= 20mA
SOURCE/SINK
1.1
4
Ω
A
ON
I
,I
DH DL
t
C
= 5nF
DH/DL
35
ns
NO
OSCILLATOR
Switching Frequency Range
Switching Frequency
Switching Frequency
Switching Frequency
125
121
495
1515
-5
1500
129
547
1725
+5
kHz
kHz
R = 500kΩ
125
521
T
f
R = 120kΩ
T
SW
R = 39.9kΩ
T
1620
120kΩ ≤ R ≤ 500kΩ
T
Switching Frequency Accuracy
%
40kΩ ≤ R ≤ 120kΩ
-8
+8
T
2
_______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
ELECTRICAL CHARACTERISTICS (continued)
(V
= 5V, V
= V , T = T = T
to T
, unless otherwise noted. Typical specifications are at T = +25°C.) (Note 1)
MAX A
CC
DD
CC
A
J
MIN
PARAMETER
SYMBOL
CONDITIONS
With respect to DH, f = 125kHz
MIN
TYP
MAX
UNITS
CLKOUT Phase Shift
φ_
180
Degrees
CLKOUT
CLKOUTL
CLKOUTH
SW
CLKOUT Output Low Level
CLKOUT Output High Level
V
I
= 2mA
0.4
V
V
SINK
SOURCE
V
I
= 2mA
4.5
200
2.0
SYNC Input-High Pulse Width
SYNC Input Clock High Threshold
SYNC Input Clock Low Threshold
SYNC Pullup Current
t
ns
V
SYNC
V
SYNCH
V
0.4
750
0.4
V
SYNCL
SYNC_OUT
I
V
= 0V
250
µA
V
RT/SYNC
SYNC Power-Off Level
V
SYNC_OFF
INDUCTOR CURRENT LIMIT
Average Current-Limit Threshold
Reverse Current-Limit Threshold
Cycle-by-Cycle Current Limit
Cycle-by-Cycle Overload
V
CSP to CSN
CSP to CSN
CSP to CSN
24.0
-3.2
26.9
-2.3
60
28.2
-0.1
mV
mV
mV
ns
CL
V
CLR
V
to V
= 75mV
CSN
260
CSP
Hiccup Divider Ratio
Hiccup Reset Delay
LIM Input Impedance
CURRENT-SENSE AMPLIFIER
CSP or CSN Input Resistance
Common-Mode Range
Input Offset Voltage
Amplifier Gain
LIM to V , no switching
CM
0.547 0.558 0.565
V/V
ms
kΩ
200
LIM to SGND
55.9
R
4
kΩ
V
CS
V
V
= 7V to 28V
0
5.5
CMR(CS)
IN
V
0.1
34.5
4
mV
V/V
MHz
OS(CS)
A
V(CS)
3dB Bandwidth
f
3dB
CURRENT-ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)
Transconductance
Open-Loop Gain
g
550
50
µS
dB
m
A
No load
VOL(CE)
DIFFERENTIAL VOLTAGE AMPLIFIER FOR LED CURRENT (DIFF)
Common-Mode Voltage Range
DIFF Output Voltage
Input Offset Voltage
V
0
+1.0
V
V
CMR(DIFF)
V
V
= V = 0V
SENSE-
0.6
CM
OS(DIFF)
SENSE+
V
-1
+1
mV
V/V
MHz
mA
kΩ
Amplifier Gain
A
0.994
1
3
1.006
V(DIFF)
3dB Bandwidth
f
C
= 20pF
DIFF
3dB
OUT(DIFF)
Minimum Output-Current Drive
SENSE+ to SENSE- Input
I
4
R
VS
V
= 0V
50
100
SENSE-
V_IOUT AMPLIFIER
Gain-Bandwidth Product
3dB Bandwidth
V
V
= 2.0V
= 2.0V
4
1
MHz
MHz
µA
V_IOUT
V_IOUT
Output Sink Current
Output Source Current
30
90
µA
_______________________________________________________________________________________
3
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
ELECTRICAL CHARACTERISTICS (continued)
(V
= 5V, V
= V , T = T = T
to T
, unless otherwise noted. Typical specifications are at T = +25°C.) (Note 1)
MAX A
CC
DD
CC
A
J
MIN
PARAMETER
Maximum Load Capacitance
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
50
pF
V_IOUT Output to I
Function
Transfer
OUT
R
= 1mΩ, 100mV ≤ V_
≤ 5.5V
132.3
135
1
137.7
mV/A
mV
SENSE
IOUT
Offset Voltage
VOLTAGE-ERROR AMPLIFIER (EAOUT)
MAX618
Open-Loop Gain
A
70
3
dB
MHz
µA
VOLEA
Unity-Gain Bandwidth
EAN Input Bias Current
f
GBW
I
V
= 2.0V
EAN
-0.2
883
+0.03
+0.2
976
B(EA)
Error Amplifier Output Clamping
Voltage
V
With respect to V
930
90
mV
CLAMP(EA)
CM
POWER-GOOD AND OVERVOLTAGE PROTECTION
PGOOD goes low when V
threshold
is below this
OUT
PGOOD Trip Level
V
87.5
92.5
%V
OUT
UV
PGOOD Output Low Level
PGOOD Output Leakage Current
OVI Trip Threshold
V
I
= 4mA
0.4
1
V
PGLO
SINK
I
PGOOD = V
µA
V
PG
CC
OVP
With respect to SGND
1.244 1.276 1.308
0.2
TH
OVI
OVI Input Bias Current
ENABLE INPUT
I
µA
EN Input High Voltage
EN Input Hysteresis
V
EN rising
2.437
13.5
2.5
0.28
15
2.562
16.5
V
V
EN
EN Pullup Current
I
µA
EN
THERMAL SHUTDOWN
Thermal Shutdown
Temperature rising
150
30
°C
°C
Thermal Shutdown Hysteresis
Note 1: Specifications at T = +25° are 100% tested. Specifications over the temperature range are guaranteed by design.
A
Note 2: Does not include an error due to finite error amplifier gain. See the Voltage-Error Amplifier (EAOUT) section.
4
_______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
Typical Operating Characteristics
(T = +25°C, using Figure 5, unless otherwise noted.)
A
CURRENT-SENSE THRESHOLD
vs. OUTPUT VOLTAGE
SUPPLY CURRENT vs. TEMPERATURE
SUPPLY CURRENT (I ) vs. FREQUENCY
Q
70
68
66
64
62
60
29.0
28.5
28.0
27.5
27.0
26.5
26.0
60
50
40
30
20
10
0
EXTERNAL CLOCK
NO DRIVER LOAD
V
= 24V
IN
V
= 12V
IN
V
= 5V
IN
V
= 12V
IN
f
= 250kHz
SW
V
= 12V
= 250kHz
IN
C
/C = 22nF
DL DH
f
SW
-40
-15
10
35
60
85
0
1
2
3
4
5
100 300 500 700 900 1100 1300 1500
FREQUENCY (kHz)
TEMPERATURE (°C)
V
(V)
OUT
DRIVER RISE TIME
vs. DRIVER LOAD CAPACITANCE
V
CC
LOAD REGULATION
vs. INPUT VOLTAGE
HICCUP CURRENT LIMIT vs. R
EXT
26.0
25.5
25.0
24.5
24.0
23.5
23.0
100
80
60
40
20
0
5.25
5.15
5.05
4.95
4.85
4.75
V
= 12V
= 250kHz
IN
f
SW
V
= 24V
IN
V
V
= 12V
IN
DL
DH
= 5V
IN
V
= 12V
= 250kHz
IN
f
SW
R1 = 1mΩ
V
= 1.5V
OUT
0
4
8
12
(MΩ)
16
20
1
6
11
16
21
0
25
50
75
100
125
150
R
CAPACITANCE (nF)
V
LOAD CURRENT (mA)
EXT
CC
DRIVER FALL TIME
vs. DRIVER LOAD CAPACITANCE
LOW-SIDE DRIVER (DL) SINK
AND SOURCE CURRENT
HIGH-SIDE DRIVER (DH) SINK
AND SOURCE CURRENT
MAX16818 toc09
MAX16818 toc08
100
80
60
40
20
0
V
= 12V
= 250kHz
C
V
= 22nF
= 12V
IN
C
V
= 22nF
= 12V
LOAD
IN
LOAD
IN
f
SW
3A/div
2A/div
DL
DH
1
6
11
16
21
100ns/div
100ns/div
CAPACITANCE (nF)
_______________________________________________________________________________________
5
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Typical Operating Characteristics (continued)
(T = +25°C, using Figure 5, unless otherwise noted.)
A
HIGH-SIDE DRIVER (DH) FALL TIME
HIGH-SIDE DRIVER (DH) RISE TIME
LOW-SIDE DRIVER (DL) RISE TIME
MAX16818 toc11
MAX16818 toc10
MAX16818 toc12
C
V
= 22nF
= 12V
C
V
= 22nF
= 12V
C
V
= 22nF
= 12V
LOAD
IN
LOAD
IN
LOAD
IN
MAX618
2V/div
2V/div
2V/div
40ns/div
40ns/div
40ns/div
FREQUENCY vs. R
T
LOW-SIDE DRIVER (DL) FALL TIME
MAX16818 toc13
10,000
1000
100
V
= 12V
IN
C
= 22nF
= 12V
LOAD
V
IN
2V/div
30
110
190
270
350
430
510
40ns/div
70
150
230
310
390
470
R (kΩ)
T
FREQUENCY vs. TEMPERATURE
SYNC, CLKOUT, AND LX WAVEFORM
MAX16818 toc16
260
258
256
254
252
250
248
246
244
242
240
V
= 12V
IN
SYNC
5V/div
CLKOUT
5V/div
V
= 12V
= 250kHz
IN
f
SW
LX
10V/div
-40
-15
10
35
60
85
1μs/div
TEMPERATURE (°C)
6
_______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
Pin Description
PIN
1
NAME
FUNCTION
PGND
N.C.
DL
Power-Supply Ground
2, 7
3
No Connection. Not internally connected.
Low-Side Gate Driver Output
Boost Flying Capacitor Connection. Reservoir capacitor connection for the high-side MOSFET driver
supply. Connect a ceramic capacitor between BST and LX.
4
BST
5
6
LX
Source connection for the high-side MOSFET.
DH
High-Side Gate Driver Output. Drives the gate of the high-side MOSFET.
Signal Ground. Ground connection for the internal control circuitry. Connect SGND and PGND
together at one point near the IC.
8, 22, 25
SGND
9
CLKOUT
PGOOD
Oscillator Output. Rising edge of CLKOUT is phase-shifted from the rising edge of DH by 180°.
Power-Good Output
10
Output Enable. Drive high or leave unconnected for normal operation. Drive low to shut down the
power drivers. EN has an internal 15µA pullup current. Connect a capacitor from EN to SGND to
program the hiccup-mode duty cycle.
11
12
EN
Switching Frequency Programming and Chip-Enable Input. Connect a resistor from RT/SYNC to
SGND to set the internal oscillator frequency. Drive RT/SYNC to synchronize the switching frequency
with external clock.
RT/SYNC
13
14
V_IOUT
LIM
Voltage Source Output Proportional to the Inductor Current. The voltage at V_IOUT = 135 x I
x R .
LED S
Current-Limit Setting Input. Connect a resistor from LIM to SGND to set the hiccup current-limit
threshold. Connect a capacitor from LIM to SGND to ignore short output overcurrent pulses.
Overvoltage Protection. Connect OVI to DIFF. When OVI exceeds 12.7% above the programmed
output voltage, DH is latched low and DL is latched high. Toggle EN or recycle the input power to
reset the latch.
15
OVI
16
17
18
CLP
EAOUT
EAN
Current-Error Amplifier Output. Compensate the current loop by connecting an RC network to ground.
Voltage-Error Amplifier Output. Connect to the external compensation network.
Voltage-Error Amplifier Inverting Input
Differential Remote-Sense Amplifier Output. DIFF is the output of a precision unity-gain amplifier
whose inputs are SENSE+ and SENSE-.
19
20
DIFF
CSN
Current-Sense Differential Amplifier Negative Input. The differential voltage between CSN and CSP is
amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.
_______________________________________________________________________________________
7
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Pin Description (continued)
PIN
NAME
FUNCTION
Current-Sense Differential Amplifier Positive Input. The differential voltage between CSN and CSP is
amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.
21
CSP
Differential LED Current-Sensing Negative Input. SENSE- is used to sense the LED current. Connect
SENSE- to the negative side of the LED current-sense resistor.
23
SENSE-
Differential LED Current-Sensing Positive Input. SENSE+ is used to sense the LED current. Connect
SENSE+ to the positive side of the LED current-sense resistor.
24
26
27
SENSE+
IN
MAX618
Supply Voltage Connection. Connect IN to V
for a +5V system.
CC
Internal +5V Regulator Output. V
and 0.1µF ceramic capacitors.
is derived from the IN voltage. Bypass V
to SGND with 4.7µF
CC
CC
V
CC
Supply Voltage for Low-Side and High-Side Drivers. Connect a parallel combination of 0.1µF and 1µF
ceramic capacitors to PGND and a 1Ω resistor to V
from internal circuitry.
to filter out the high peak currents of the driver
28
—
V
CC
DD
Exposed Paddle. Connect the exposed paddle to a copper pad (SGND) to improve power
dissipation.
EP
8
_______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
Typical Application Circuits
ON/OFF
R6
C3
V
IN
V
CC
7V TO 28V
R3
R4
R5
13
V
LED
C2
L1
14
12
9
10
PGOOD CLKOUT SGND
N.C.
11
8
V
LED
LIM
V_IOUT RT/SYNC
EN
C10
D1
15
7
6
5
4
3
2
1
OVI
C9
C8
Q1
R12
DH
LX
16 CLP
R11
EAOUT
EAN
17
18
LED
STRING
C1
R7
C7
BST
DL
MAX16818
R10
19 DIFF
20 CSN
21 CSP
R2
N.C.
R1
PGND
SGND SENSE- SENSE+ SGND
24 25
IN
V
V
DD
CC
26
22
23
27
28
V
CC
V
IN
R8
C5
C4
C6
Figure 1. Typical Application Circuit for a Boost LED Driver (Nonsynchronous)
_______________________________________________________________________________________
9
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Typical Application Circuits (continued)
ON/OFF
R6
C3
V
IN
V
CC
7V TO 28V
LED
R3
R4
R5
13
R2
STRING
1 TO 6
LEDS
V
LED
C2
MAX618
L1
14
12
9
10
11
8
V
LED
LIM
V_IOUT RT/SYNC
EN
PGOOD CLKOUT SGND
N.C.
C10
D1
15
7
6
5
4
3
2
1
OVI
C9
C8
Q1
R12
DH
LX
16 CLP
V
V
CC
R11
EAOUT
17
18
CC
RS+
MAX4073T
R7
RS-
C7
OUT
EAN
BST
DL
MAX16818
C1
R10
19 DIFF
20 CSN
21 CSP
N.C.
R1
PGND
SGND SENSE- SENSE+ SGND
24 25
IN
V
V
DD
CC
26
22
23
27
28
V
CC
V
IN
R8
C5
C4
C6
Figure 2. Typical Application Circuit for an Input-Referred Buck-Boost LED Driver (Input: 7V to 28V, Output: 1 to 6 LEDs in Series)
10 ______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
Typical Application Circuits (continued)
ON/OFF
R6
C4
V
IN
V
CC
7V TO 28V
R3
R4
R5
13
V
LED
C3
L1
14
12
9
10
PGOOD CLKOUT SGND
N.C.
11
8
V
LED
LIM
V_IOUT RT/SYNC
EN
C1
C11
D1
15
7
6
5
4
3
2
1
OVI
C10
Q1
R12
DH
LX
16 CLP
R11
EAOUT
EAN
17
18
LED
STRING
C2
L2
C9
C8
BST
DL
MAX16818
R10
19 DIFF
20 CSN
21 CSP
R2
R7
N.C.
R1
PGND
SGND SENSE- SENSE+ SGND
24 25
IN
V
V
DD
CC
26
22
23
27
28
V
CC
V
IN
R8
C6
C5
C7
Figure 3. Typical Application Circuit for a SEPIC LED Driver
______________________________________________________________________________________ 11
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Typical Application Circuits (continued)
ON/OFF
R6
C3
V
CC
V
R3
R4
R5
13
IN
V
LED
7V TO 18V
MAX618
14
12
9
10
11
8
C2
LIM
V_IOUT RT/SYNC
EN
PGOOD CLKOUT SGND
N.C.
C11
15
7
6
5
4
3
2
1
OVI
Q1
C10
R12
DH
LX
16 CLP
V
LED
R11
L1
EAOUT
17
18
C9
C4
Q3
R7
C8
EAN
BST
DL
MAX16818
LED
STRING
Q2
C1
R10
19 DIFF
20 CSN
21 CSP
D2
N.C.
R2
R1
PGND
SGND SENSE- SENSE+ SGND
24 25
IN
V
V
DD
CC
26
22
23
27
28
V
CC
V
IN
R8
C6
C5
C7
Figure 4. Application Circuit for a Ground-Referred Buck-Boost LED Driver
12 ______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
Typical Application Circuits (continued)
V
CC
R4
V
IN
C3
R3
13
7V TO 28V
ON/OFF
10
14
12
9
11
8
C2
LIM
V_IOUT RT/SYNC
EN
PGOOD CLKOUT SGND
N.C.
C11
15
7
6
5
4
3
2
1
OVI
C10
C9
R10
R9
DH
LX
16 CLP
L1
EAOUT
EAN
17
18
C4
Q1
R5
C8
BST
DL
MAX16818
LED
STRING
D1
C1
R8
19 DIFF
20 CSN
21 CSP
N.C.
R2
R1
PGND
SGND SENSE- SENSE+ SGND
24 25
IN
V
V
DD
CC
26
22
23
27
28
V
CC
V
IN
R6
C6
C5
C7
Figure 5. Application Circuit for a Buck LED Driver
______________________________________________________________________________________ 13
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Functional Diagram
V
CC
I
S
EN
IN
0.5V x V
CC
5V
LDO
REGULATOR
UVLO
POR
TEMP SENSOR
MAX618
V
CC
TO INTERNAL
CIRCUITS
HICCUP MODE
CURRENT LIMIT
MAX16818
LIM
V
CM
126.7kΩ
100kΩ
S
R
Q
Q
R
T
0.5 x V
CLAMP
CLP
CSP
CSN
C
t
A
= 34.5
V
V
CM
CA
g
m
= 500μS
V
DD
PWM
COMPARATOR
A
V
= 4
CEA
BST
DH
LX
V_IOUT
SGND
V
HIGH
V
CLAMP
CLAMP
LOW
CPWM
Q
RAMP
S
R
2 x f (V/s)
S
CLK
RT/SYNC
OSCILLATOR
Q
DL
CLKOUT
DIFF
RAMP
GENERATOR
PGND
+0.6V
SENSE-
SENSE+
EAOUT
PGOOD
N
DIFF
AMP
0.1 x V
REF
ERROR AMP
VEA
EAN
0.12 x V
REF
OVP LATCH
LATCH
OVP COMP
SOFT-
START
V
V
= 0.6V
(0.6V)
REF
CLEAR ON UVLO RESET OR
ENABLE LOW
CM
OVI
Figure 6. MAX16818 Functional Diagram
14 ______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
P = V x I
D
IN CC
x (Q + Q )]
Detailed Description
I
= I + [f
CC
Q
SW G1 G2
The MAX16818 is a high-performance average-current-
mode PWM controller for high-power, high-brightness
LEDs (HBLEDs). Average current-mode control is the
ideal method for driving HBLEDs. This technique offers
inherently stable operation, reduces component derat-
ing and size by accurately controlling the inductor cur-
rent. The device achieves high efficiency at high
current (up to 30A) with a minimum number of external
components. The high- and low-side drivers source
and sink up to 4A for lower switching losses while dri-
ving high-gate-charge MOSFETs. The MAX16818’s
CLKOUT output is 180° out-of-phase with respect to the
high-side driver. CLKOUT drives a second MAX16818
LED driver out of phase, reducing the input-capacitor
ripple current.
where Q
and Q
are the total gate charge of the
G2
G1
low-side and high-side external MOSFETs at V
=
GATE
5V, I is 3.5mA (typ), and f
Q
cy of the converter.
is the switching frequen-
SW
Undervoltage Lockout (UVLO)
The MAX16818 includes an undervoltage lockout with
hysteresis and a power-on-reset circuit for converter
turn-on. The UVLO rising threshold is internally set at
4.35V with a 200mV hysteresis. Hysteresis at UVLO
eliminates chattering during startup.
Most of the internal circuitry, including the oscillator,
turns on when the input voltage reaches 4V. The
MAX16818 draws up to 3.5mA of current before the
input voltage reaches the UVLO threshold.
The MAX16818 consists of an inner average current loop
representing inductor current and an outer voltage loop
voltage-error amplifier (VEA) that directly controls LED
current. The combined action of the two loops results in
a tightly regulated LED current. The inductor current is
sensed across a current-sense resistor. The differential
amplifier senses LED current through a sense resistor in
series with the LEDs and the resulting sensed voltage is
compared against an internal 0.6V reference at the error-
amplifier input. The MAX16818 will adjust the LED cur-
rent to within 1% accuracy to maintain emitted spectrum
of the light in HBLEDs.
Soft-Start
The MAX16818 has an internal digital soft-start for a
monotonic, glitch-free rise of the output current. Soft-
start is achieved by the controlled rise of the error
amplifier dominant input in steps using a 5-bit counter
and a 5-bit DAC. The soft-start DAC generates a linear
ramp from 0 to 0.7V. This voltage is applied to the error
amplifier at a third (noninverting) input. As long as the
soft-start voltage is lower than the reference voltage,
the system converges to that lower reference value.
Once the soft-start DAC output reaches 0.6V, the refer-
ence takes over and the DAC output continues to climb
to 0.7V, assuring that it does not interfere with the refer-
ence voltage.
IN, V , and V
CC
DD
The MAX16818 accepts either a 4.75V to 5.5V or 7V to
28V input voltage range. All internal control circuitry
operates from an internally regulated nominal voltage of
Internal Oscillator
5V (V ). For input voltages of 7V or greater, the inter-
CC
CC
The internal oscillator generates a clock with the fre-
nal V
CC
regulator steps the voltage down to 5V. The
quency proportional to the inverse of R . The oscillator
T
V
output voltage is a regulated 5V output capable of
sourcing up to 60mA. Bypass the V
frequency is adjustable from 125kHz to 1.5MHz with
better than 8% accuracy using a single resistor con-
nected from RT/SYNC to SGND. The frequency accura-
cy avoids the over-design, size, and cost of passive
filter components like inductors and capacitors. Use
the following equation to calculate the oscillator fre-
quency:
to SGND with
CC
4.7µF and 0.1µF low-ESR ceramic capacitors for high-
frequency noise rejection and stable operation.
The MAX16818 uses V
to power the low-side and
DD
high-side drivers. Isolate V
from V
with a 1Ω resis-
DD
CC
tor and put a 1µF capacitor in parallel with a 0.1µF
capacitor to ground to prevent high-current noise spikes
created by the driver from disrupting internal circuitry.
For 120kΩ ≤ R ≤ 500kΩ:
T
10
6.25 x 10
The TQFN is a thermally enhanced package and can
dissipate up to 2.7W. The high-power packages allow
the high-frequency, high-current converter to operate
from a 12V or 24V bus. Calculate power dissipation in
the MAX16818 as a product of the input voltage and the
R
=
T
f
SW
For 40kΩ ≤ R ≤ 120kΩ:
T
total V
regulator output current (I ). I
includes qui-
10
CC
CC CC
6.40 x 10
R
=
T
escent current (I ) and gate-drive current (I ):
Q
DD
f
SW
______________________________________________________________________________________ 15
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
The oscillator also generates a 2V
voltage-ramp sig-
PWM comparator (CPWM) (Figure 7). The precision CA
P-P
nal for the PWM comparator and a 180° out-of-phase
clock signal for CLKOUT to drive a second LED regula-
tor out-of-phase.
amplifies the sense voltage across R by a factor of
S
34.5. The inverting input to the CEA senses the CA out-
put. The CEA output is the difference between the volt-
age-error amplifier output (EAOUT) and the amplified
voltage from the CA. The RC compensation network
connected to CLP provides external frequency compen-
sation for the CEA. The start of every clock cycle
enables the high-side drivers and initiates a PWM on-
cycle. Comparator CPWM compares the output voltage
from the CEA with a 0V to 2V ramp from the oscillator.
The PWM on-cycle terminates when the ramp voltage
exceeds the error voltage. Compensation for the outer
LED current loop varies based upon the topology.
Synchronization
The MAX16818 can be easily synchronized by con-
necting an external clock to RT/SYNC. If an external
clock is present, then the internal oscillator is disabled
and the external clock is used to run the device. If the
external clock is removed, the absence of clock for
32µs is detected and the circuit starts switching from
the internal oscillator. Pulling RT/SYNC to ground for at
least 50µs disables the converter. Use an open-collec-
tor transistor to synchronize the MAX16818 with the
external system clock.
MAX618
The MAX16818 outer LED current control loop consists
of the differential amplifier (DIFF AMP), reference volt-
age, and VEA. The unity-gain differential amplifier pro-
vides true differential remote sensing of the voltage
Control Loop
The MAX16818 uses an average-current-mode control
scheme to regulate the output current (Figure 7). The
main control loop consists of an inner current loop for
controlling the inductor current and an outer current
loop for regulating the LED current. The inner current
loop absorbs the inductor pole reducing the order of the
outer current loop to that of a single-pole system. The
across the LED current set resistor, R . The differential
LS
amplifier output connects to the inverting input (EAN) of
the VEA. The DIFF AMP is bypassed and the inverting
input is available to the pin for direct feedback. The
noninverting input of the VEA is internally connected to
an internal precision reference voltage, set to 0.6V. The
VEA controls the inner current loop (Figure 6). A feed-
back network compensates the outer loop using the
EAOUT and EAIN pins.
current loop consists of a current-sense resistor (R ), a
S
current-sense amplifier (CA), a current-error amplifier
(CEA), an oscillator providing the carrier ramp, and a
C
CF
R
CF
C
CFF
CSN
CSP
CLP
V
IN
CA
EAOUT
MAX16818
SENSE+
I
L
600mV
DIFF
AMP
CEA
LED
STRING
VEA
DRIVE
CPWM
SENSE-
EAN
C
OUT
V
REF
+ V = 1.2V
CM
R
LS
DIFF
R
S
Figure 7. MAX16818 Control Loop
16 ______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
Inductor Current-Sense Amplifier
The differential current-sense amplifier (CA) provides a
DC gain of 34.5. The maximum input offset voltage of
the current-sense amplifier is 1mV and the common-
mode voltage range is 0 to 5.5V (IN = 7V to 28V). The
current-sense amplifier senses the voltage across a
current-sense resistor. The maximum common-mode
Current-Error Amplifier
(For Inductor Currents)
The MAX16818 has a transconductance current-error
amplifier (CEA) with a typical g of 550µS and 320µA
m
output sink- and source-current capability. The current-
error amplifier output CLP serves as the inverting input
to the PWM comparator. CLP is externally accessible to
provide frequency compensation for the inner current
loops (Figure 7). Compensate (CEA) so the inductor
current negative slope, which becomes the positive
slope to the inverting input of the PWM comparator, is
less than the slope of the internally generated voltage
ramp (see the Compensation section).
voltage is 3.6V when V = 5V.
IN
Inductor Peak-Current Comparator
The peak-current comparator provides a path for fast
cycle-by-cycle current limit during extreme fault condi-
tions, such as an inductor malfunction (Figure 8). Note
the average current-limit threshold of 26.9mV still limits
the output current during short-circuit conditions. To
prevent inductor saturation, select an inductor with a
saturation current specification greater than the average
current limit. Proper inductor selection ensures that only
the extreme conditions trip the peak-current compara-
tor, such as an inductor with a shorted turn. The 60mV
threshold for triggering the peak-current limit is twice the
full-scale average current-limit voltage threshold. The
peak-current comparator has only a 260ns delay.
PWM Comparator and R-S Flip-Flop
The PWM comparator (CPWM) sets the duty cycle for
each cycle by comparing the output of the current-error
amplifier to a 2V
ramp. At the start of each clock
P-P
cycle, an R-S flip-flop resets and the high-side driver
(DH) goes high. The comparator sets the flip-flop as
soon as the ramp voltage exceeds the CLP voltage,
thus terminating the on-cycle (Figure 8).
V
DD
PEAK-CURRENT
COMPARATOR
60mV
CLP
A
= 34.5
V
CSP
CSN
MAX16818
g
= 550μS
m
CA
BST
DH
LX
CEA
SET
Q
S
R
VEA
EAN
CPWM
RAMP
2 x f (V/s)
S
EAOUT
CLK
Q
DL
CLR
PGND
SHDN
Figure 8. MAX16818 Phase Circuit
______________________________________________________________________________________ 17
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Differential Amplifier
BST
to power the low- and high-
The DIFF AMP facilitates remote sensing at the load
(Figure 7). It provides true differential LED current
The MAX16818 uses V
DD
side MOSFET drivers. The high-side driver derives its
power through a bootstrap capacitor and V supplies
(through the R sense resistor) sensing while rejecting
LS
DD
the common-mode voltage errors due to high-current
ground paths. The VEA provides the difference
between the differential amplifier output (DIFF) and the
desired LED current-sense voltage. The differential
amplifier has a bandwidth of 3MHz. The difference
between SENSE+ and SENSE- is regulated to 0.6V.
Connect SENSE+ to the positive side of the LED current-
sense resistor and SENSE- to the negative side of the
LED current-sense resistor (which is often PGND).
power internally to the low-side driver. Connect a
0.47µF low-ESR ceramic capacitor between BST and
LX. Connect a Schottky rectifier from BST to V . Keep
DD
the loop formed by the boost capacitor, rectifier, and IC
small on the PCB.
Protection
The MAX16818 includes output overvoltage protection
(OVP). During fault conditions when the load goes to
high impedance (opens), the controller attempts to
maintain LED current. The OVP protection disables the
MAX16818 whenever the voltage exceeds the thresh-
old, protecting the external circuits from undesirable
voltages.
MAX618
MOSFET Gate Drivers (DH, DL)
The high-side (DH) and low-side (DL) drivers drive the
gates of external n-channel MOSFETs (Figures 1–5).
The drivers’ 4A peak sink- and source-current capabili-
ty provides ample drive for the fast rise and fall times of
the switching MOSFETs. Faster rise and fall times result
in reduced cross-conduction losses. Due to physical
Current Limit
The VEA output is clamped to 930mV with respect to
the common-mode voltage (V ). Average-current-
CM
realities, extremely low gate charges and R
DS(ON)
mode control has the ability to limit the average current
sourced by the converter during a fault condition. When
a fault condition occurs, the VEA output clamps to
930mV with respect to the common-mode voltage
(0.6V) to limit the maximum current sourced by the con-
resistance of MOSFETs are typically exclusive of each
other. MOSFETs with very low R will have a high-
DS(ON)
er gate charge and vice versa. Choosing the high-side
MOSFET (Q1) becomes a trade-off between these two
attributes. Applications where the input voltage is much
higher than the output voltage result in a low duty cycle
where conduction losses are less important than
switching losses. In this case, choose a MOSFET with
verter to I
= 26.9mV / R . The hiccup current limit
S
LIMIT
overrides the average current limit. The MAX16818
includes hiccup current-limit protection to reduce the
power dissipation during a fault condition. The hiccup
current-limit circuit derives inductor current information
from the output of the current amplifier. This signal is
very low gate charge and a moderate R
DS(ON).
Conversely, for applications where the output voltage is
near the input voltage resulting in duty cycles much
compared against one half of V
. With no
CLAMP(EA)
greater than 50%, the R
losses become at least
DS(ON)
resistor connected from the LIM pin to ground, the hic-
cup current limit is set at 90% of the full-load average
equal, or even more important than the switching losses.
In this case, choose a MOSFET with very low R
DS(ON)
current limit. Use R
to increase the hiccup current
EXT
and moderate gate charge. Finally, for the applications
where the duty cycle is near 50%, the two loss compo-
nents are nearly equal, and a balanced MOSFET with
limit from 90% to 100% of the full load average limit.
The hiccup current limit can be disabled by connecting
LIM to SGND. In this case, the circuit follows the aver-
age current-limit action during overload conditions.
moderate gate charge and R
work best.
DS(ON)
In a buck topology, the low-side MOSFET (Q2) typically
operates in a zero voltage switching mode, thus it does
not have switching losses. Choose a MOSFET with very
Overvoltage Protection
The OVP comparator compares the OVI input to the
overvoltage threshold. A detected overvoltage event
latches the comparator output forcing the power stage
into the OVP state. In the OVP state, the high-side
MOSFET turns off and the low-side MOSFET latches on.
Connect OVI to the center tap of a resistor-divider from
low R
and moderate gate charge.
DS(ON)
Size both the high-side and low-side MOSFETs to han-
dle the peak and RMS currents during overload condi-
tions. The driver block also includes a logic circuit that
provides an adaptive nonoverlap time to prevent shoot-
through currents during transition. The typical nonover-
lap time between the high-side and low-side MOSFETs
is 35ns.
V
to SGND. In this case, the center tap is compared
LED
against 1.276V. Add an RC delay to reduce the sensitivity
of the overvoltage circuit and avoid nuisance tripping of
the converter. Disable the overvoltage function by con-
necting OVI to SGND.
18 ______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
span from the output to the input. This effectively
removes the boost-only restriction of the regulator in
Applications Information
Application Circuit Descriptions
This section provides some detail regarding the appli-
cation circuits in the Simplified Diagram and Figures
1–5. The discussion includes some description of the
topology as well as basic attributes.
Figure 1, allowing the voltage across the LEDs to be
greater than or less than the input voltage. LED current
sensing is not ground-referenced, so a high-side cur-
rent-sense amplifier is used to measure current.
SEPIC LED Driver
Figure 3 shows the MAX16818 configured as a SEPIC
LED driver. While buck topologies require the output to
be lesser than the input, and boost topologies require
the output to be greater than the input, a SEPIC topolo-
gy allows the output voltage to be greater than, equal
to, or less than the input. In a SEPIC topology, the volt-
age across C1 is the same as the input voltage, and L1
and L2 are the same inductance. Therefore, when Q1
conducts (on-time), both inductors ramp up current at
the same rate. The output capacitor supports the out-
put voltage during this time. During the off-time, L1 cur-
rent recharges C1 and combines with L2 to provide
current to recharge C2 and supply the load current.
Since the voltage waveform across L1 and L2 are
exactly the same, it is possible to wind both inductors
on the same core (a coupled inductor). Although volt-
ages on L1 and L2 are the same, RMS currents can be
quite different so the windings may have a different
gauge wire. Because of the dual inductors and seg-
mented energy transfer, the efficiency of a SEPIC con-
verter is somewhat lower than standard bucks or
boosts. As in the boost driver, the current-sense resis-
tor connects to ground, allowing the output voltage of
the LED driver to exceed the rated maximum voltage of
the MAX16818.
High-Frequency LED Current Pulser
The Simplified Diagram shows the MAX16818 providing
high-frequency, high-current pulses to the LEDs. The
basic topology must be a buck, since the inductor
always connects to the load in that configuration (in all
other topologies, the inductor disconnects from the
load at one time or another). The design minimizes the
current ripple by oversizing the inductor, which allows
for a very small (0.01µF) output capacitor. When MOS-
FET Q3 turns on, it diverts the current around the LEDs
at a very fast rate. Q3 also discharges the output
capacitor, but since the capacitor is so small, it does
not stress the MOSFET. Resistor R1 senses the LED/Q3
current and there is no reaction to the short that Q3
places across the LEDs. This design is superior in that
it does not attempt to actually change the inductor cur-
rent at high frequencies and yet the current in the LEDs
varies from zero to full in very small periods of time. The
efficiency of this technique is very high. Q3 must be
able to dissipate the LED current applied to its R
DS(ON)
at some maximum duty cycle. If the circuit needs to
control extremely high currents, use paralleled
MOSFETs. PGOOD is low during LED pulsed-current
operation.
Boost LED Driver
In Figure 1, the external components configure the
MAX16818 as a boost converter. The circuit applies the
input voltage to the inductor during the on-time, and
then during the off-time the inductor, which is in series
with the input capacitor, charges the output capacitor.
Because of the series connection between the input
voltage and the inductor, the output voltage can never
go lower than the input voltage. The design is nonsyn-
chronous, and since the current-sense resistor con-
nects to ground, the power supply can go to any output
voltage (above the input) as long as the components are
rated appropriately. R2 again provides the sense voltage
the MAX16818 uses to regulate the LED current.
Ground-Referenced Buck/Boost LED Driver
Figure 4 depicts a buck/boost topology. During the on-
time with this circuit, the current flows from the input
capacitor, through Q1, L1, and Q3 and back to the
input capacitor. During the off-time, current flows up
through Q2, L1, D1, and to the output capacitor C1.
This topology resembles a boost in that the inductor
sits between the input and ground during the on-time.
However, during the off-time the inductor resides
between ground and the output capacitor (instead of
between the input and output capacitors in boost
topologies), so the output voltage can be any voltage
less than, equal to, or greater than the input voltage. As
compared to the SEPIC topology, the buck/boost does
not require two inductors or a series capacitor, but it
does require two additional MOSFETs.
Input-Referenced LED Driver
The circuit in Figure 2 shows a step-up/step-down reg-
ulator. It is similar to the boost converter in Figure 1 in
that the inductor is connected to the input and the
MOSFET is essentially connected to ground. However,
rather than going from the output to ground, the LEDs
Buck Driver with Synchronous Rectification
In Figure 5, the input voltage can go from 7V to 28V and,
because of the ground-based current-sense resistor, the
______________________________________________________________________________________ 19
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
output voltage can be as high as the input. The synchro-
nous MOSFET keeps the power dissipation to a minimum,
especially when the input voltage is large when com-
pared to the voltage on the LED string. It is important to
keep the current-sense resistor, R1, inside the LC loop,
so that ripple current is available. To regulate the LED
current, R2 creates a voltage that the differential amplifier
compares to 0.6V. If power dissipation is a problem in R2,
add a noninverting amplifier and reduce the value of the
sense resistor accordingly.
surface-mount inductor series available from various
manufacturers.
For example, for a buck regulator and 2 LEDs in series,
calculate the minimum inductance at V
= 13.2V,
IN(MAX)
= 330kHz:
V
= 7.8V, ΔI = 400mA, and f
LED
L
SW
Buck regulators:
(13.2 − 7.8) x 7.8
13.2 x 330k x 0.4
L
MIN
=
= 24.2μH
MAX618
Inductor Selection
The switching frequencies, peak inductor current, and
allowable ripple at the output determine the value and
size of the inductor. Selecting higher switching frequen-
cies reduces the inductance requirement, but at the
cost of lower efficiency. The charge/discharge cycle of
the gate and drain capacitances in the switching
MOSFETs create switching losses. The situation wors-
ens at higher input voltages, since switching losses are
proportional to the square of the input voltage. The
MAX16818 can operate up to 1.5MHz, however for
For a boost regulator with four LEDs in series, calculate
the minimum inductance at V
= 13.2V, V
=
IN(MAX)
= 330kHz:
LED
15.6V, ΔI =400mA, and f
L
SW
Boost regulators:
(15.6 − 13.2) x 13.2
15.6 x 330k x 0.4
L
MIN
=
= 15.3μH
The average-current-mode control feature of the
MAX16818 limits the maximum peak inductor current
and prevents the inductor from saturating. Choose an
inductor with a saturating current greater than the
worst-case peak inductor current. Use the following
equation to determine the worst-case inductor current:
V
> +12V, use lower switching frequencies to limit the
IN
switching losses.
The following discussion is for buck or continuous
boost-mode topologies. Discontinuous boost, buck-
boost, and SEPIC topologies are quite different in
regards to component selection.
V
R
ΔI
CL
2
CL
L
=
+
LPEAK
S
Use the following equations to determine the minimum
inductance value:
where R is the inductor sense resistor and V
S
0.0282V.
=
CL
Buck regulators:
(V
− V
) x V
Switching MOSFETs
INMAX
V
LED
LED
L
MIN
=
When choosing a MOSFET for voltage regulators, con-
x f
x ΔI
INMAX
SW
L
sider the total gate charge, R
, power dissipation,
DS(ON)
and package thermal impedance. The product of the
MOSFET gate charge and on-resistance is a figure of
merit, with a lower number signifying better perfor-
mance. Choose MOSFETs optimized for high-frequen-
cy switching applications.
Boost regulators:
(V
− V
) x V
x ΔI
SW L
LED
INMAX
x f
INMAX
L
MIN
=
V
LED
The average current from the MAX16818 gate-drive
output is proportional to the total capacitance it drives
at DH and DL. The power dissipated in the MAX16818
is proportional to the input voltage and the average
where V
is the total voltage across the LED string.
LED
As a first approximation choose the ripple current, ΔI ,
L
equal to approximately 40% of the output current.
Higher ripple current allows for smaller inductors, but it
also increases the output capacitance for a given volt-
age ripple requirement. Conversely, lower ripple cur-
rent increases the inductance value, but allows the
output capacitor to reduce in size. This trade-off can be
altered once standard inductance and capacitance val-
ues are chosen. Choose inductors from the standard
drive current. See the IN, V , and V
section to
DD
CC
determine the maximum total gate charge allowed from
the combined driver outputs. The gate-charge and
drain-capacitance (CV2) loss, the cross-conduction
loss in the upper MOSFET due to finite rise/fall times,
2
and the I R loss due to RMS current in the MOSFET
R
account for the total losses in the MOSFET.
DS(ON)
20 ______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
Buck Regulator
Boost Regulator
_) caused by the MOS-
Estimate the power loss (PD
_) caused by the high-side
Estimate the power loss (PD
MOS
MOS
and low-side MOSFETs using the following equations:
FET using the following equations:
PD
= (Q x V
x f ) +
PD = (Q x V x f ) +
FET
G
DD
SW
MOS−HI
G
DD
SW
V
x I
x (t + t ) x f
2
⎛
⎞
IN
OUT R F SW
2
+ (R
x I
)
⎜
⎝
⎟
⎠
DS(ON)
RMS−HI
V
x I
x (t + t ) x f
2
⎛
⎞
IN
OUT
R
F
SW
⎜
⎝
⎟
⎠
D
3
2
2
I
=
(I
+ I
+ I
x I ) x
PK
RMS−HI
VALLEY
PK
VALLEY
2
+ (R
x I
)
DS(ON)
RMS−HI
where Q , R
, t , and t are the upper-switching
R F
G
DS(ON)
For a boost regulator in continuous mode, D = V
/
LEDs
MOSFET’s total gate charge, on-resistance at maximum
operating temperature, rise time, and fall time, respectively.
(V + V
), I
= (I
- Δ / 2) and I = (I
IN
LEDs VALLEY
OUT
L
PK
OUT
+ ΔI / 2).
L
The voltage across the MOSFET:
= V
D
3
2
2
I
=
(I
+ I
+ I x I ) x
VALLEY PK
RMS−HI
VALLEY
PK
V
+ V
F
MOSFET
LED
where V is the maximum forward voltage of the diode.
F
For the buck regulator, D = V
/ V , I
=
LEDs
+ ΔI / 2).
IN VALLEY
The output diode on a boost regulator must be rated to
(I
- ΔI / 2) and I = (I
OUT
L
PK
OUT L
handle the LED series voltage, V
. It should also
LED
have fast reverse-recovery characteristics and should
handle the average forward current that is equal to the
LED current.
PD
= (Q x V
x f ) +
DD SW
MOS−LO
G
2
(R
x I
)
DS(ON)
RMS − LO
Input Capacitors
For buck regulator designs, the discontinuous input
current waveform of the buck converter causes large
ripple currents in the input capacitor. The switching fre-
quency, peak inductor current, and the allowable peak-
to-peak voltage ripple reflected back to the source
dictate the capacitance requirement. Increasing
switching frequency or paralleling out-of-phase con-
verters lowers the peak-to-average current ratio, yield-
ing a lower input capacitance requirement for the same
(1−D)
3
2
2
+ I
I
=
(I
+ I
x I ) x
VALLEY PK
RMS − LO
VALLEY
PK
For example, from the typical specifications in the
Applications Information section with V = 7.8V, the
OUT
high-side and low-side MOSFET RMS currents are
0.77A and 0.63A, respectively, for a 1A buck regulator.
Ensure that the thermal impedance of the MOSFET
package keeps the junction temperature at least +25°C
below the absolute maximum rating. Use the following
equation to calculate the maximum junction tempera-
x θ ) + T , where θ and T are
JA A JA A
the junction-to-ambient thermal impedance and ambi-
ent temperature, respectively.
LED current. The input ripple is comprised of ΔV
(caused by the capacitor discharge) and ΔV
Q
ture: T = (PD
J
MOS
ESR
(caused by the ESR of the capacitor). Use low-ESR
ceramic capacitors with high-ripple-current capability at
the input. Assume the contributions from the ESR and
capacitor discharge are equal to 30% and 70%, respec-
tively. Calculate the input capacitance and ESR required
for a specified ripple using the following equation:
To guarantee that there is no shoot-through from V to
IN
PGND, the MAX16818 produces a nonoverlap time of
35ns. During this time, neither high- nor low-side MOS-
FET is conducting, and since the output inductor must
maintain current flow, the intrinsic body diode of the
low-side MOSFET becomes the conduction path. Since
this diode has a fairly large forward voltage, a Schottky
diode (in parallel to the low-side MOSFET) diverts current
flow from the MOSFET body diode because of its lower
forward voltage, which, in turn, increases efficiency.
ΔV
ESR
ESR
=
IN
ΔI
⎛
⎞
L
2
I
+
⎜
⎝
⎟
⎠
OUT
______________________________________________________________________________________ 21
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Buck:
Current Limit
In addition to the average current limit, the MAX16818
also has hiccup current limit. The hiccup current limit is
set to 10% below the average current limit to ensure that
the circuit goes in hiccup mode during continuous out-
put short circuit. Connecting a resistor from LIM to
ground increases the hiccup current limit, while shorting
LIM to ground disables the hiccup current-limit circuit.
I
x D(1−D)
OUT
C
=
IN
ΔV x f
Q
SW
where I
is the output current of the converter. For
OUT
example, at V = 13.2V, V
= 7.8V, I
= 1A, ΔI =
OUT L
IN
SW
LED
0.4A, and f
= 330kHz, the ESR and input capaci-
tance are calculated for the input peak-to-peak ripple
of 100mV or less yielding an ESR and capacitance
value of 25mΩ and 10µF.
Average Current Limit
The average-current-mode control technique of the
MAX16818 accurately limits the maximum output current.
The MAX16818 senses the voltage across the sense
MAX618
For boost regulator designs, the input-capacitor current
waveform is dominated by the inductor, a triangle wave
a magnitude of ΔI For simplicity’s sake, the current
waveform can be approximated by a square wave with
a magnitude that is half that of the triangle wave.
Calculate the input capacitance and ESR required for a
specified ripple using the following equation:
L.
resistor and limit the peak inductor current (I
)
L-PK
accordingly. The on-cycle terminates when the current-
sense voltage reaches 25.5mV (min). Use the following
equation to calculate the maximum current-sense resis-
tor value:
ΔV
ΔI
ESR
ESR
=
0.0255
IN
R
=
S
L
I
OUT
0.75 x 10−
3
Boost:
PD
=
R
R
S
ΔI
2
L
x D
C
=
where PD is the dissipation in the series resistors.
R
Select a 5% lower value of R to compensate for any
S
IN
ΔV x f
Q
SW
parasitics associated with the PCB. Also, select a non-
inductive resistor with the appropriate power rating.
Duty cycle, D, for a boost regulator is equal to (V
-
OUT
LED
V ) / V
IN
As an example, at V = 13.2V, V
=
OUT.
IN
15.6V, I
= 1A, ΔI = 0.4A, and f
= 330kHz, the
OUT
L
SW
Hiccup Current Limit
The hiccup current-limit value is always 10% lower than
the average current-limit threshold, when LIM is left
unconnected. Connect a resistor from LIM to SGND to
increase the hiccup current-limit value from 90% to
100% of the average current-limit value. The average
current-limit architecture accurately limits the average
output current to its current-limit threshold. If the hiccup
current limit is programmed to be equal or above the
average current-limit value, the output current does not
reach the point where the hiccup current limit can trig-
ger. Program the hiccup current limit at least 5% below
the average current limit to ensure that the hiccup cur-
rent-limit circuit triggers during overload. See the
ESR and input capacitance are calculated for the input
peak-to-peak ripple of 100mV or less yielding an ESR
and capacitance value of 250mΩ and 1µF, respectively.
Output Capacitor
For buck converters, the inductor always connects to
the load, so the inductance controls the ripple current.
The output capacitance shunts a fraction of this ripple
current and the LED string absorbs the rest. The
capacitor reactance (which includes the capacitance
and ESR) and the dynamic impedance of the LED
diode string form a conductance divider that splits the
ripple current between the LEDs and the capacitor. In
many cases, the capacitor is very large as compared to
the ESR, and this divider reduces to the ESR and the
LED resistance.
Hiccup Current Limit vs. R
Operating Characteristics.
graph in the Typical
EXT
Boost converters place a harsher requirement on the
output capacitors as they must sustain the full load dur-
ing the on-time of the MOSFET and are replenished
during the off-time. The ripple current in this case is the
full load current, and the holdup time is equal to the
duty cycle times the switching period.
22 ______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
Compensation
The main control loop consists of an inner current loop
(inductor current) and an outer LED current loop. The
MAX16818 uses an average current-mode control
scheme to regulate the LED current (Figure 7). The VEA
output provides the controlling voltage for the current
source. The inner current loop absorbs the inductor
pole reducing the order of the LED current loop to that
of a single-pole system.
In order to choose C , the external loop gain must be
CF
considered. The following equation describes the over-
all loop gain for a buck regulator, which is the ratio of a
small-signal change in the output of amplifier CA to the
output of amplifier CEA:
External Loop Buck:
ΔV
R
x V x A
IN V
CA
S
V
=
ΔV
x sL
CEA
RAMP
The major consideration when designing the current
control loop is making certain that the inductor down-
slope (which becomes an upslope at the output of the
CEA) does not exceed the internal ramp slope. This is a
necessary condition to avoid subharmonic oscillations
similar to those in peak current mode with insufficient
slope compensation. This requires that the gain at the
output of the CEA be limited based on the following
equation (Figure 6):
where A is the gain of the current amplifier (34.5) and
RAMP
Multiplying the external loop gain with the CEA amplifier
gain gives the total loop equation and solves for the fre-
quency that yields a gain of 1 results in:
V
V
is voltage peak (2V) of the internal ramp.
Total Loop Buck:
V
2πV
x f
SW
IN
Buck:
f
=
CMAX
OUT
V
× f
× L
× g
RAMP
SW
To be stable, the gain of the CEA amplifier must have a
zero placed before f . C creates a pole at the
R
R
≤
CF
A
× R × V
V
S
OUT m
x L
x V
OUT
CMAX
CF
origin and the combination of R and C creates the
f
CF
CF
SW
≤ 105
CF
zero. Lower frequency zeros result in less bandwidth,
R
S
but greater phase margin. The pole created by C
CFF
(in conjunction with R ) is for noise reduction and can
CF
where V
Boost:
= 2V, g = 550µs, A = 34.5.
m V
RAMP
be placed well past the crossover frequency.
The following equation describes the external loop gain
for a boost regulator:
V
× f
× L
− V ) × g
IN m
RAMP
SW
R
R
≤
CF
External Loop Boost:
A
× R × (V
V
S
OUT
x L
− V )
OUT IN
f
SW
ΔV
R
x V
x A
x sL
CA
S
OUT V
≤ 105
CF
=
R
x (V
S
ΔV
V
RAMP
CEA
Solving for the gain of the CEA amplifier,
Buck:
To get the total loop gain for a boost regulator, multiply
the external loop gain with the gain of the CEA amplifier
to arrive at the following:
ΔV
ΔV
V
V
x f
x L
CEA
RAMP
SW
Total Loop Boost:
g
× R
=
=
m
CF
x R x A
OUT S V
CA
f
x V
OUT
SW
f
=
CMAX
Boost:
× R
2π (V
− V )
IN
OUT
ΔV
ΔV
V
x f
x L
CEA
RAMP
SW
As in the buck regulator, the zero created by R and
g
=
CF
=
CF
m
(V
− V ) x R x A
CA
OUT IN S V
C
sits at a frequency lower than f
to maintain
CF
CMAX
stable operation
.
______________________________________________________________________________________ 23
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Power Dissipation
The TQFN is a thermally enhanced package and can dis-
sipate about 2.7W. The high-power package makes the
high-frequency, high-current LED driver possible to oper-
ate from a 12V or 24V bus. Calculate power dissipation in
the MAX16818 as a product of the input voltage and the
7) Avoid long traces between the V
bypass capaci-
DD
tors, the driver output of the MAX16818, the MOS-
FET gates, and PGND. Minimize the loop formed by
the V
bypass capacitors, bootstrap diode, boot-
CC
strap capacitor, the MAX16818, and the upper
MOSFET gate.
total V
regulator output current (I ). I
includes qui-
CC
CC CC
8) Distribute the power components evenly across the
board for proper heat dissipation.
escent current (I ) and gate drive current (I ):
Q
DD
9) Provide enough copper area at and around the
switching MOSFETs, inductor, and sense resistors
to aid in thermal dissipation.
P
= V x I
IN CC
D
MAX618
I
= I + f
x (Q + Q
G2
)
]
[
CC
Q
SW
G1
10) Use wide copper traces (2oz) to keep trace induc-
tance and resistance low to maximize efficiency.
Wide traces also cool heat-generating components.
where Q and Q are the total gate charge of the low-
G1
G2
side and high-side external MOSFETs at V
= 5V, I
Q
GATE
is estimated from the Supply Current (I ) vs. Frequency
Q
graph in the Typical Operating Characteristics, and f
SW
is the switching frequency of the LED driver. For boost
drivers, only consider one gate charge, Q
.
G1
Use the following equation to calculate the maximum
power dissipation (P ) in the chip at a given ambi-
Pin Configuration
DMAX
ent temperature (T ):
A
TOP VIEW
P
= 34.5 x (150 - T ) mW.
DMAX
A
17 16
21
19 18
15
20
PCB Layout Guidelines
Use the following guidelines to layout the switching
voltage regulator:
22
23
LIM
SGND
SENSE-
SENSE+
14
13
V_IOUT
RT/SYNC
EN
1) Place the IN, V , and V
CC
bypass capacitors
DD
24
12
11
close to the MAX16818.
25
SGND
IN
2) Minimize the area and length of the high current
loops from the input capacitor, upper switching
MOSFET, inductor, and output capacitor back to
the input capacitor negative terminal.
MAX16818
10 PGOOD
26
CLKOUT
SGND
V
9
8
CC 27
* EXPOSED PAD
V
DD
28
3) Keep short the current loop formed by the lower
switching MOSFET, inductor, and output capacitor.
+
6
2
3
4
5
7
1
4) Place the Schottky diodes close to the lower
MOSFETs and on the same side of the PCB.
TQFN
5) Keep the SGND and PGND isolated and connect
them at one single point.
6) Run the current-sense lines CSP and CSN very
close to each other to minimize the loop area.
Similarly, run the remote voltage-sense lines
SENSE+ and SENSE- close to each other. Do not
cross these critical signal lines through power cir-
cuitry. Sense the current right at the pads of the
current-sense resistors.
Chip Information
TRANSISTOR COUNT: 5654
PROCESS: BiCMOS
24 ______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX618
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
______________________________________________________________________________________ 25
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Package Information (continued)
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
MAX618
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
26 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2006 Maxim Integrated Products
is a registered trademark of Maxim Integrated Products, Inc.
Heaney
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