LTC1872ES6#TRPBF [Linear]
LTC1872 - Constant Frequency Current Mode Step-Up DC/DC Controller in SOT-23; Package: SOT; Pins: 6; Temperature Range: -40°C to 85°C;
| 型号： | LTC1872ES6#TRPBF |
| 厂家： | 
Linear |
| 描述： | LTC1872 - Constant Frequency Current Mode Step-Up DC/DC Controller in SOT-23; Package: SOT; Pins: 6; Temperature Range: -40°C to 85°C 开关 光电二极管 |
| 文件： | 总14页 (文件大小：255K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC1872
Constant Frequency
Current Mode Step-Up
DC/DC Controller in SOT-23
DescripTion
The LTC®1872 is a constant frequency current mode step-
up DC/DC controller providing excellent AC and DC load
and line regulation. The device incorporates an accurate
undervoltagelockoutfeaturethatshutsdowntheLTC1872
when the input voltage falls below 2.0V.
FeaTures
n
High Efficiency: Over 90%
n
High Output Currents Easily Achieved
n
Wide V Range: 2.5V to 9.8V
IN
n
n
n
n
V
Limited Only by External Components
OUT
Constant Frequency 550kHz Operation
Burst Mode™ Operation at Light Load
Current Mode Operation for Excellent Line and Load
Transient Response
The LTC1872 boasts a 2.ꢀ5 output voltage accuracy
and consumes only 270µA of quiescent current. For ap-
plications where efficiency is a prime consideration, the
LTC1872 is configured for Burst Mode operation, which
enhances efficiency at low output current.
n
n
n
n
Low Quiescent Current: 270µA
Shutdown Mode Draws Only 8µA Supply Current
2.ꢀ5 Reference Accuracy
In shutdown, the device draws a mere 8µA. The high
ꢀꢀ0kHz constant operating frequency allows the use of a
small external inductor.
Tiny 6-Lead SOT-23 Package
applicaTions
The LTC1872 is available in a small footprint 6-lead
SOT-23.
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners.
n
Lithium-Ion-Powered Applications
n
Cellular Telephones
n
Wireless Modems
n
Portable Computers
n
Scanners
Typical applicaTion
Efficiency vs Load Current
V
IN
3.3V
100
95
90
85
80
75
70
65
C1
10µF
10V
R1
V
V
= 3.3V
OUT
IN
0.03Ω
= 5V
147k
1
5
L1
4.7µH
I
/RUN
V
IN
TH
LTC1872
220pF
80.6k
V
5V
1A
OUT
2
3
4
6
–
GND
SENSE
NGATE
+
C2
2× 22F
6.3V
D1
M1
V
FB
422k
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: MURATA GRM42-2X5R226K6.3
D1: IR10BQ015
L1: MURATA LQN6C4R7M04
M1: IRLMS2002
R1: DALE 0.25W
1
10
100
1000
LOAD CURRENT (mA)
1872 TA01
1872 TA01b
Figure 1. LTC1872 High Output Current 3.3V to 5V Boost Converter
1872fa
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For more information www.linear.com/LTC1872
LTC1872
absoluTe MaxiMuM raTings
pin conFiguraTion
(Note 1)
Input Supply Voltage (V )......................... –0.3V to 10V
IN
TOP VIEW
–
SENSE , NGATE Voltages ............. –0.3V to (V + 0.3V)
IN
I
TH
/RUN 1
GND 2
6 NGATE
5 V
V , I /RUN Voltages .............................. –0.3V to 2.4V
FB TH
IN
–
NGATE Peak Output Current (<10µs) ........................ 1A
Storage Ambient Temperature Range ....–6ꢀ°C to 1ꢀ0°C
Operating Temperature Range (Note 2)....–40°C to 8ꢀ°C
Junction Temperature (Note 3) ............................ 1ꢀ0°C
Lead Temperature (Soldering, 10 sec)...................300°C
V
3
4 SENSE
FB
S6 PACKAGE
6-LEAD PLASTIC SOT-23
= 1ꢀ0°C, θ = 230°C/W
T
JMAX
JA
orDer inForMaTion
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC1872ES6#PBF
LTC1872ES6#TRPBF
LTMK
6-Lead Plastic SOT-23
–40°C to 8ꢀ°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on nonstandard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
elecTrical characTerisTics The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER
CONDITIONS
Typicals at V = 4.2V (Note 4)
MIN
TYP
MAX
UNITS
Input DC Supply Current
Normal Operation
Sleep Mode
IN
2.4V ≤ V ≤ 9.8V
270
230
8
420
370
22
µA
µA
µA
µA
IN
2.4V ≤ V ≤ 9.8V
IN
Shutdown
UVLO
2.4V ≤ V ≤ 9.8V, V /RUN = 0V
IN ITH
V
IN
< UVLO Threshold
6
10
l
l
Undervoltage Lockout Threshold
V
V
Falling
Rising
1.ꢀꢀ
1.8ꢀ
2.00
2.10
2.3ꢀ
2.40
V
V
IN
IN
Shutdown Threshold (at I /RUN)
0.1ꢀ
0.2ꢀ
0.3ꢀ
0.ꢀ
0.ꢀꢀ
0.8ꢀ
V
TH
Start-Up Current Source
V
ITH
/RUN = 0V
µA
l
l
Regulated Feedback Voltage
0°C to 70°C(Note ꢀ)
–40°C to 8ꢀ°C(Note ꢀ)
0.780
0.770
0.800
0.800
0.820
0.830
V
V
V
Input Current
(Note ꢀ)
10
ꢀꢀ0
40
ꢀ0
nA
kHz
ns
FB
Oscillator Frequency
Gate Drive Rise Time
Gate Drive Fall Time
V
C
C
= 0.8V
ꢀ00
114
6ꢀ0
FB
= 3000pF
= 3000pF
LOAD
LOAD
40
ns
Peak Current Sense Voltage
(Note 6)
120
mV
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 2: The LTC1872E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 8ꢀ°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 5: The LTC1872 is tested in a feedback loop that servos V to the
output of the error amplifier.
Note 6: Guaranteed by design at duty cycle = 305. Peak current sense
FB
voltage is V /6.67 at duty cycle <405, and decreases as duty cycle
REF
Note 3: T is calculated from the ambient temperature T and power
increases due to slope compensation as shown in Figure 2.
J
A
dissipation P according to the following formula:
D
T = T + (P • θ °C/W)
J
A
D
JA
1872fa
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For more information www.linear.com/LTC1872
LTC1872
Typical perForMance characTerisTics
Undervoltage Lockout Trip
Voltage vs Temperature
Reference Voltage
vs Temperature
Normalized Oscillator Frequency
vs Temperature
825
820
815
810
805
800
795
790
785
780
775
10
8
2.24
2.20
2.16
2.12
2.08
2.04
2.00
1.96
1.92
1.88
1.84
V
= 4.2V
V
= 4.2V
IN
V
IN
FALLING
IN
6
4
2
0
–2
–4
–6
–8
–10
–55 –35 –15
5
25 45 65 85 105 125
–55 –35 –15
5
25 45 65 85 105 125
85
105 125
–55 –35 –15
5
25 45 65
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
1872 G01
1872 G02
1872 G03
Maximum Current Sense Trip
Voltage vs Duty Cycle
Shutdown Threshold
vs Temperature
600
560
520
480
440
400
360
320
280
240
200
130
120
110
100
90
V
= 4.2V
V
A
= 4.2V
= 25°C
IN
IN
T
80
70
60
50
60 70
DUTY CYCLE (%)
–55 –35 –15
5
45
85 105 125
20 30 40 50
80 90 100
25
65
TEMPERATURE (°C)
1872 G04
1872 G05
pin FuncTions
–
I /RUN (Pin 1): This pin performs two functions. It
SENSE (Pin 4): The Negative Input to the Current Com-
TH
serves as the error amplifier compensation point as well
as the run control input. Nominal voltage range for this
pin is 0.7V to 1.9V. Forcing this pin below 0.3ꢀV causes
the device to be shut down. In shutdown all functions are
disabled and the NGATE pin is held low.
parator.
V
(Pin 5): Supply Pin. Must be closely decoupled to
GND Pin 2.
IN
NGATE (Pin 6): Gate Drive for the External N-Channel
MOSFET. This pin swings from 0V to V .
IN
GND (Pin 2): Ground Pin.
V (Pin3):Receivesthefeedbackvoltagefromanexternal
FB
resistive divider across the output.
1872fa
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LTC1872
FuncTional DiagraM
–
V
SENSE
IN
5
4
+
–
ICMP
V
IN
RS
NGATE
6
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
SLOPE
COMP
R
OSC
Q
S
–
+
FREQ
FOLDBACK
BURST
CMP
OVP
+
–
+
–
0.3V
V
+
SLEEP
REF
0.15V
60mV
V
IN
EAMP
V
REF
+
–
0.8V
0.5µA
V
FB
I
/RUN
TH
1
3
+
V
IN
V
IN
0.3V
–
0.35V
+
SHDN
UV
SHDN
CMP
VOLTAGE
REFERENCE
V
REF
0.8V
–
GND
2
UNDERVOLTAGE
LOCKOUT
1.2V
1872FD
(Refer to Functional Diagram)
operaTion
Main Control Loop
relative to the 0.8V reference, which in turn causes the
TH
current matches the new load current.
I /RUN voltage to increase until the average inductor
The LTC1872 is a constant frequency current mode
switchingregulator.Duringnormaloperation,theexternal
N-channel power MOSFET is turned on each cycle by the
oscillator and turned off when the current comparator
(ICMP) resets the RS latch. The peak inductor current
at which ICMP resets the RS latch is controlled by the
The main control loop is shut down by pulling the ITH/
RUN pin low. Releasing ITH/RUN allows an internal 0.ꢀµA
current source to charge up the external compensation
network. When the ITH/RUN pin reaches 0.3ꢀV, the main
control loop is enabled with the ITH/RUN voltage then
pulled up to its zero current level of approximately 0.7V.
Astheexternalcompensationnetworkcontinuestocharge
up, the corresponding output current trip level follows,
allowing normal operation.
voltage on the I /RUN pin, which is the output of the
TH
error amplifier EAMP. An external resistive divider con-
nected between V
and ground allows the EAMP to
OUT
receive an output feedback voltage V . When the load
FB
current increases, it causes a slight decrease in V
FB
1872fa
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LTC1872
operaTion
Comparator OVP guards against transient overshoots
>7.ꢀ5 by turning off the external N-channel power
MOSFET and keeping it off until the fault is removed.
Overvoltage Protection
The overvoltage comparator in the LTC1872 will turn the
external MOSFET off when the feedback voltage has risen
7.ꢀ5abovethereferencevoltageof0.8V.Thiscomparator
has a typical hysteresis of 20mV.
Burst Mode Operation
The LTC1872 enters Burst Mode operation at low load
currents. In this mode, the peak current of the inductor is
Slope Compensation and Inductor’s Peak Current
set as if V /RUN = 1V (at low duty cycles) even though
ITH
The inductor’s peak current is determined by:
the voltage at the I /RUN pin is at a lower value. If the
TH
V
10 R
ITH−0.7
inductor’saveragecurrentisgreaterthantheloadrequire-
IPK
=
ment, the voltage at the I /RUN pin will drop. When the
(
)
TH
SENSE
I /RUN voltage goes below 0.8ꢀV, the sleep signal goes
TH
when the LTC1872 is operating below 405 duty cycle.
However, once the duty cycle exceeds 405, slope com-
pensationbeginsandeffectivelyreducesthepeakinductor
current. The amount of reduction is given by the curves
in Figure 2.
high, turning off the external MOSFET. The sleep signal
goes low when the I /RUN voltage goes above 0.92ꢀV
TH
and the LTC1872 resumes normal operation. The next
oscillator cycle will turn the external MOSFET on and the
switching cycle repeats.
Short-Circuit Protection
Undervoltage Lockout
Since the power switch in a boost converter is not in
series with the power path from input to load, turning off
the switch provides no protection from a short-circuit at
the output. External means such as a fuse in series with
the boost inductor must be employed to handle this fault
condition.
TopreventoperationoftheN-channelMOSFETbelowsafe
input voltage levels, an undervoltage lockout is incorpo-
rated into the LTC1872. When the input supply voltage
drops below approximately 2.0V, the N-channel MOSFET
andallcircuitryisturnedoffexcepttheundervoltageblock,
which draws only several microamperes.
110
100
90
80
70
60
50
I
= 0.4I
PK
RIPPLE
AT 5% DUTY CYCLE
= 0.2I
40
30
20
10
I
RIPPLE
PK
AT 5% DUTY CYCLE
V
= 4.2V
IN
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
1872 F02
Figure 2. Maximum Output Current vs Duty Cycle
1872fa
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LTC1872
applicaTions inForMaTion
The basic LTC1872 application circuit is shown in
Figure 1. External component selection is driven by the
load requirement and begins with the selection of L1 and
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
ofasmallerinductorforthesameamountofinductorripple
current. However, this is at the expense of efficiency due
to an increase in MOSFET gate charge losses.
R
(= R1). Next, the power MOSFET and the output
SENSE
diodeD1isselectedfollowedbyC (=C1)andC (=C2).
IN
OUT
R
Selection for Output Current
SENSE
The inductance value also has a direct effect on ripple
R
is chosen based on the required output current.
SENSE
current. The ripple current, I
, decreases with higher
RIPPLE
With the current comparator monitoring the voltage de-
inductance or frequency and increases with higher V
.
OUT
veloped across R
, the threshold of the comparator
SENSE
The inductor’s peak-to-peak ripple current is given by:
determinestheinductor’speakcurrent.Theoutputcurrent
ꢀ
ꢂ
ꢁ
ꢃ
VIN
f L
( )
VOUT+VD−V
IN ꢅ
ꢄ
the LTC1872 can provide is given by:
IRIPPLE
=
VOUT +VD
ꢀ
ꢂ
ꢁ
ꢃ
ꢅ
ꢄ
0.12 IRIPPLE
V
IN
VOUT +VD
IOUT
=
−
R
2
wherefistheoperatingfrequency.Acceptinglargervalues
of I allows the use of low inductances, but results
SENSE
RIPPLE
where I
is the inductor peak-to-peak ripple current
in higher output voltage ripple and greater core losses.
A reasonable starting point for setting ripple current is:
RIPPLE
(see Inductor Value Calculation section) and V is the
D
forward drop of the output diode at the full rated output
current.
ꢀ
ꢂ
ꢁ
ꢃ
ꢅ
ꢄ
V
OUT +VD
I
RIPPLE =0.4 I
(
)
OUT MAX
(
)
V
IN
A reasonable starting point for setting ripple current is:
VOUT +VD
In Burst Mode operation, the ripple current is normally set
such that the inductor current is continuous during the
burst periods. Therefore, the peak-to-peak ripple current
must not exceed:
IRIPPLE = O.4 I
(
)
(
)
OUT
V
IN
Rearranging the above equation, it becomes:
ꢀ
ꢂ
ꢁ
ꢃ
ꢅ
ꢄ
0.03
RSENSE
1
I
V
IN
IRIPPLE
≤
RSENSE
=
10
( )
V
+VD
(
)
OUT
OUT
for Duty Cycle <40%
This implies a minimum inductance of:
ꢀ
ꢂ
ꢁ
ꢃ
However,foroperationthatisabove405dutycycle,slope
compensation’s effect has to be taken into consideration
to select the appropriate value to provide the required
amount of current. Using the scaling factor (SF, in 5) in
VIN
0.03
V
OUT+VD−V
VOUT +VD
IN ꢅ
LMIN
=
ꢀ
ꢂ
ꢁ
ꢃ
ꢅ
ꢄ
ꢄ
f
R
SENSE
Figure 2, the value of R
is:
SENSE
A smaller value than L
could be used in the circuit;
MIN
however, the inductor current will not be continuous
during burst periods.
ꢀ
ꢂ
ꢃ
ꢅ
ꢄ
V
SF
IN
RSENSE
=
10 I
( )
100 V +VD
(
)
(
)
ꢁ
OUT
OUT
1872fa
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LTC1872
applicaTions inForMaTion
Inductor Selection
It is important to adequately specify the diode peak cur-
rent and average power dissipation so as not to exceed
the diode ratings.
When selecting the inductor, keep in mind that inductor
saturation current has to be greater than the current limit
set by the current sense resistor. Also, keep in mind that
the DC resistance of the inductor will affect the efficiency.
OfftheshelfinductorsareavailablefromMurata,Coilcraft,
Toko, Panasonic, Coiltronics and many other suppliers.
Schottky diodes are recommended for low forward drop
and fast switching times. Remember to keep lead length
short and observe proper grounding (see Board Layout
Checklist) to avoid ringing and increased dissipation.
Power MOSFET Selection
C and C
Selection
IN
OUT
The main selection criteria for the power MOSFET are the
To prevent large input voltage ripple, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current for a boost
converter is approximately equal to:
threshold voltage V
, the “on” resistance R
,
GS(TH)
DS(ON)
and total gate charge.
reverse transfer capacitance C
RSS
Since the LTC1872 is designed for operation down to low
input voltages, a logic level threshold MOSFET (R
DS(ON)
CIN Required IRMS ≈ 0.3 I
guaranteed at V = 2.ꢀV) is required for applications
(
)
RIPPLE
GS
that work close to this voltage. When these MOSFETs are
where I
is as defined in the Inductor Value Calcula-
used, make sure that the input supply to the LTC1872 is
RIPPLE
tion section.
less than the absolute maximum V rating, typically 8V.
GS
Notethatcapacitormanufacturer’sripplecurrentratingsare
oftenbasedon2000hoursoflife.Thismakesitadvisableto
further derate the capacitor, or to choose a capacitor rated
at a higher temperature than required. Several capacitors
may be paralleled to meet the size or height requirements
in the design. Due to the high operating frequency of the
TherequiredminimumR
oftheMOSFETisgoverned
DS(ON)
by its allowable power dissipation given by:
PP
RDS(ON)
≅
2
1+δp
DC I
(
)
(
)
IN
where P is the allowable power dissipation and δp is the
P
temperature dependency of R
. (1 + δp) is generally
DS(ON)
LTC1872, ceramic capacitors can also be used for C .
IN
given for a MOSFET in the form of a normalized R
DS(ON)
Always consult the manufacturer if there is any question.
vs temperature curve, but δp = 0.00ꢀ/°C can be used as
an approximation for low voltage MOSFETs. DC is the
maximum operating duty cycle of the LTC1872.
The selection of C
is driven by the required effective
OUT
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
The output ripple (∆V ) is approximated by:
Output Diode Selection
OUT
ꢀ
ꢃ
ꢅ
ꢄ
VOUT +VD IRIPPLE
Under normal load conditions, the average current con-
ducted by the diode in a boost converter is equal to the
output load current:
ΔVOUT ≈ I •
+
•
ꢂ O
V
2
ꢁ
IN
1
ꢉ2
ꢋ
ꢆ
ꢈ
ꢈ
ꢇ
I
D(avg) = IOUT
ꢀ
ꢃ2
1
2
ꢂ
ꢅ
ESR +
ꢂ
ꢅ
ꢋ
2πfCOUTꢄ
ꢁ
ꢊ
1872fa
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LTC1872
applicaTions inForMaTion
where f is the operating frequency, C
is the output
Setting Output Voltage
OUT
capacitanceandI
istheripplecurrentintheinductor.
RIPPLE
The LTC1872 develops a 0.8V reference voltage between
the feedback (Pin 3) terminal and ground (see Figure 4).
By selecting resistor R1, a constant current is caused to
flow through R1 and R2 to set the overall output voltage.
The regulated output voltage is determined by:
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
productofanyaluminumelectrolyticatasomewhathigher
price. The output capacitor RMS current is approximately
equal to:
ꢀ
ꢃ
ꢅ
ꢄ
R2
R1
VOUT =0.8V 1+
ꢂ
ꢁ
105
100
95
IPK • DC−DC2
V
REF
where I is the peak inductor current and DC is the switch
PK
V
ITH
duty cycle.
90
Whenusingelectrolyticoutputcapacitors, iftherippleand
ESR requirements are met, there is likely to be far more
capacitance than required.
85
80
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum
electrolytic and dry tantalum capacitors are both available
in surface mount configurations. An excellent choice of
tantalum capacitors is the AVX TPS and KEMET Tꢀ10
series of surface mount tantalum capacitors. Also,
ceramic capacitors in XꢀR pr X7R dielectrics offer excel-
lent performance.
75
2.0
2.2
2.4
2.6
2.8
3.0
INPUT VOLTAGE (V)
1872 F03
Figure 3. Line Regulation of VREF and VITH
V
OUT
R2
LTC1872
3
V
FB
R1
Low Supply Operation
Although the LTC1872 can function down to approxi-
mately 2.0V, the maximum allowable output current is
1872 F04
Figure 4. Setting Output Voltage
reduced when V decreases below 3V. Figure 3 shows
IN
the amount of change as the supply is reduced down to
2V. Also shown in Figure 3 is the effect of V on V as
IN
REF
V goes below 2.3V.
IN
1872fa
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LTC1872
applicaTions inForMaTion
For most applications, an 80k resistor is suggested for
R1. To prevent stray pickup, locate resistors R1 and R2
close to LTC1872.
2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each
time a MOSFET gate is switched from low to high to
low again, a packet of charge, dQ, moves from V
to ground. The resulting dQ/dt is a current out of V
IN
Efficiency Considerations
IN
which is typically much larger than the contoller’s DC
Theefficiencyofaswitchingregulatorisequaltotheoutput
power divided by the input power times 1005. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
supply current. In continuous mode, I
= f(Qp).
GATECHG
2
3. I R losses are predicted from the DC resistances of
the MOSFET, inductor and current sense resistor.
The MOSFET R
multiplied by duty cycle times
DS(ON)
the average output current squared can be summed
with I R losses in the inductor ESR in series with the
current sense resistor.
Efficiency = 1005 – (η1 + η2 + η3 + ...)
2
where η1, η2, etc. are the individual losses as a percent-
age of input power.
4. Theoutputdiodeisamajorsourceofpowerlossathigh
currents. The diode loss is calculated by multiplying
the forward voltage by the load current.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
lossesinLTC1872circuits:1)LTC1872DCbiascurrent,2)
2
ꢀ. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
MOSFET gate charge current, 3) I R losses and 4) voltage
drop of the output diode.
1. The V current is the DC supply current, given in the
IN
2
Transition Loss = 2(V ) I
C
(f)
IN IN(MAX) RSS
electricalcharacteristics, thatexcludesMOSFETdriver
and control currents. V current results in a small loss
IN
Other losses, including C and C
losses, and inductor core losses, generally account for
less than 25 total additional loss.
ESR dissipative
IN
OUT
which increases with V .
IN
1872fa
9
For more information www.linear.com/LTC1872
LTC1872
applicaTions inForMaTion
PC Board Layout Checklist
4. Connect the end of R
as close to V (Pin ꢀ) as
SENSE
IN
+
possible. The V pin is the SENSE of the current
IN
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC1872. These items are illustrated graphically in
the layout diagram in Figure ꢀ. Check the following in
your layout:
comparator.
–
ꢀ. The trace from SENSE (Pin 4) to the Sense resistor
should be kept short. The trace should connect close
to R
.
SENSE
1. The Schottky diode should be closely connected
between the output capacitor and the drain of the
external MOSFET.
6. Keep the switching node NGATE away from sensitive
small signal nodes.
7. The V pin should connect directly to the feedback
FB
2. The (+) plate of C should connect to the sense resis-
resistors. The resistive divider R1 and R2 must be
IN
tor as closely as possible. This capacitor provides AC
current to the inductor.
connected between the (+) plate of C
ground.
and signal
OUT
3. The input decoupling capacitor (0.1µF) should be
connected closely between V (Pin ꢀ) and ground
IN
(Pin 2).
V
V
IN
1
2
3
6
5
4
I
/RUN NGATE
LTC1872
TH
M1
L1
R
S
R
ITH
GND
V
IN
+
0.1µF
C
IN
D1
–
V
SENSE
C
FB
ITH
OUT
+
R2
C
OUT
R1
BOLD LINES INDICATE HIGH CURRENT PATHS
1872 F05
Figure 5. LTC1872 Layout Diagram (See PC Board Layout Checklist)
1872fa
10
For more information www.linear.com/LTC1872
LTC1872
Typical applicaTion
LTC1872 12V/500mA Boost Converter
V
IN
3V TO 9.8V
C1
10µF
10V
R1
0.033Ω
1
5
L1
I
/RUN
V
TH
IN
10µH
LTC1872
10k
V
2
3
4
6
OUT
–
GND
SENSE
NGATE
12V
+
C2
47µF
16V
220pF
D1
M1
V
FB
1.1M
1872 TA02
78.7k
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: AVX TPSE476M016R0150
D1: IR10BQ015
L1: COILTRONICS UP2B-100
M1: Si9804DV
R1: DALE 0.25W
LTC1872 Three-Cell White LED Driver
V
= 3 AA CELLS ≈ 2.7V TO 4.8V
IN
C1
10µF
10V
R1
0.27Ω
AA
AA
AA
1
5
L1
I
/RUN
V
TH
IN
150µH
V
≈ 28.8V
OUT
LTC1872
10k
220pF
(WITH 8 LEDs)
2
3
4
6
–
GND
SENSE
NGATE
+
C2
15µF
35V
C3
15mA
D0
M1
V
0.1µF
CERAMIC
FB
D1
D2
1 TO 8
WHITE
LEDs
•
•
•
D8
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: AVX TPSD156M035R0300
D0: MOTOROLA MBR0540
L1: COILCRAFT DO1608C-154
M1: Si9804
R1: DALE 0.25W
53.6Ω
1872 TA04
D1-D7: CMD333UWC
1872fa
11
For more information www.linear.com/LTC1872
LTC1872
package DescripTion Dimensions in inches (millimeters) unless otherwise noted.
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
2.90 BSC
(NOTE 4)
0.62
MAX
0.95
REF
1.22 REF
1.4 MIN
1.50 – 1.75
(NOTE 4)
2.80 BSC
3.85 MAX 2.62 REF
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
DATUM ‘A’
0.01 – 0.10
1.00 MAX
0.30 – 0.50 REF
1.90 BSC
0.09 – 0.20
(NOTE 3)
S6 TSOT-23 0302
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
1872fa
12
For more information www.linear.com/LTC1872
LTC1872
revision hisTory
REV
DATE
DESCRIPTION
PAGE NUMBER
A
09/1ꢀ Revised package drawing
12
1872fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
13
LTC1872
Typical applicaTion
LTC1872 –2.5V to 3.3V/0.5A Boost Converter
LTC1872 2.7V to 9.8V Input
to 3.3V/1.2A Output SEPIC Converter
R1
0.034Ω
C2
V
IN
2¥ 100F
2.7V TO 9.8V
+
C
IN
R
CS
0.03Ω
1
5
10V
L1
4.7µH
10µF
I
/RUN
V
IN
TH
C
C1
220pF
10V, X5R
R
C1
10k
LTC1872
10k
V
3.3V
0.5A
OUT
V
1
5
OUT
2
3
4
6
–
I
/RUN
V
IN
TH
3.3V/1.2A
GND
SENSE
NGATE
+
C01
D1
220pF
C1
L1A
L1B
LTC1872
180µF
4V, SP
D1
332k
V
M1
FB
MBRM120
2
3
4
6
–
GND
SENSE
NGATE
CS
4.7µF
10V
V
FB
+
R
U1
f2
80.6k
0.1µF
CERAMIC
100µF
10V
R
f1
252k
M1
1872 TA05
80.6k
180k
C
, CS; TOKO, MURATA OR TAIYO YUDEN
: PANASONIC EEFUE0G181R
FOR V
= 5V CHANGE
V
IN
01
OUT
IN
–2.5V
C
R
C
TO 427kΩ AND
f1
1872 TA03
L1: BH ELECTRONICS 511-1012
M1: IRLMS2002
R
TO 150µF, 6V PANASONIC
C1, C2: AVX TPSE107M010R0100
D1: MOTOROLA MBR2045CT
L1: COILTRONICS UP2B-4R7
M1: Si9804DV
01
SP TYPE CAPACITOR
R1: DALE 0.25W
: DALE OR IRC
U1: PANASONIC 2SB709A
CS
relaTeD parTs
PART NUMBER
LT1304
DESCRIPTION
COMMENTS
Micropower DC/DC Converter with Low-Battery Detector
1.7MHz, Single Cell Micropower DC/DC Converter
1.4MHz, Single Cell DC/DC Converter in ꢀ-Lead SOT-23
Low Voltage Current Mode PWM Controller
120µA Quiescent Current, 1.ꢀV ≤ V ≤ 8V
IN
LT1610
30µA Quiescent Current, V Down to 1V
IN
LT1613
Internally Compensated, V Down to 1V
IN
LT1619
8-Lead MSOP Package, 1.9V ≤ V ≤ 18V
IN
LT1680
High Power DC/DC Step-Up Controller
Operation Up to 60V, Fixed Frequency Current Mode
8-Pin N-Channel Drive, 3.ꢀV ≤ V ≤ 36V
LTC1624
LT161ꢀ
High Efficiency SO-8 N-Channel Switching Regulator Controller
Micropower Step-Up DC/DC Converter in SOT-23
IN
20µA Quiescent Current, V Down to 1V
IN
LTC1700
LTC1772
No RSENSE Synchronous Current Mode DC/DC Step-Up Controller
Constant Frequency Current Mode Step-Down DC/DC Controller
9ꢀ5 Efficient, 0.9V ≤ V ≤ ꢀV, ꢀꢀ0kHz Operation
IN
V
IN
2.ꢀV to 9.8V, I
up to 4A, SOT-23 Package
OUT
LTC3401/LTC3402 1A/2A, 3MHz Micropower Synchronous Boost Converter
10-Lead MSOP Package, 0.ꢀV ≤ V ≤ ꢀV
IN
1872fa
1872f LT/TP 0915 4K REV A • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 9ꢀ03ꢀ-7417
14
●
●
LINEAR TECHNOLOGY CORPORATION 2000
(408)432-1900 FAX: (408) 434-0ꢀ07 www.linear.com/LTC1872
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