LTC1628-SYNC [Linear]
High Efficiency, 2-Phase Synchronous Step-Down Switching Regulator; 高效率,两相同步降压型开关稳压器型号: | LTC1628-SYNC |
厂家: | Linear |
描述: | High Efficiency, 2-Phase Synchronous Step-Down Switching Regulator |
文件: | 总32页 (文件大小:375K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LTC1628-SYNC
High Efficiency, 2-Phase
Synchronous Step-Down Switching Regulator
U
FEATURES
DESCRIPTIO
The LTC®1628-SYNC is a high performance dual step-
down switching regulator controller that drives all
N-channel synchronous power MOSFET stages. A con-
stant frequency current mode architecture allows phase-
lockable frequency of up to 300kHz. Power loss and noise
due to the ESR of the input capacitors are minimized by
operating the two controller output stages out of phase.
■
Out-of-Phase Controllers Reduce Required Input
Capacitance and Power Supply Induced Noise
OPTI-LOOPTM Compensation Minimizes COUT
Dual N-Channel MOSFET Synchronous Drive
±1% Output Voltage Accuracy
■
■
■
■
■
■
■
■
■
■
■
■
■
■
■
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Power Good Output Voltage Monitor
Phase-Lockable Fixed Frequency 150kHz to 300kHz
Wide VIN Range: 3.5V to 36V Operation
Very Low Dropout Operation: 99% Duty Cycle
Adjustable Soft-Start Current Ramping
Foldback Output Current Limiting
Latched Short-Circuit Shutdown with Defeat Option
Output Overvoltage Protection
OPTI-LOOP compensation allows the transient response
tobeoptimizedoverawiderangeofoutputcapacitanceand
ESRvalues. Theprecision0.8Vreferenceandpowergood
output indicator are compatible with future microproces-
sor generations, and a wide 3.5V to 30V (36V maximum)
input supply range encompasses all battery chemistries.
Remote Output Voltage Sense
A RUN/SS pin for each controller provides both soft-start
and optional timed, short-circuit shutdown. Current
foldback limits MOSFET dissipation during short-circuit
conditions when overcurrent latchoff is disabled. Output
overvoltage protection circuitry latches on the bottom
MOSFET until VOUT returns to normal. The FCB mode pin
can select among Burst Mode, constant frequency mode
and continuous inductor current mode or regulate a
secondarywinding. TheLTC1628-SYNCincludesapower
good output pin that indicates when both outputs are
within 7.5% of their designed set point.
Low Shutdown IQ: 20µA
5V and 3.3V Standby Regulators
Small 28-Lead SSOP Package
Selectable Constant Frequency or Burst ModeTM
Operation
U
APPLICATIO S
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Notebook and Palmtop Computers, PDAs
■
Telecom Systems
■
Portable Instruments
■
Battery-Operated Digital Devices
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode and OPTI-LOOP are trademarks of Linear Technology Corporation.
■
DC Power Distribution Systems
U
V
IN
5.2V TO 28V
TYPICAL APPLICATIO
C
IN
+
1µF
22µF
4.7µF
D3
CERAMIC
50V
D4
V
PGOOD INTV
IN
CC
CERAMIC
M1
M3
M4
TG1
TG2
L1
6.3µH
L2
6.3µH
C
, 0.1µF
B2
BOOST1
SW1
BOOST2
SW2
LTC1628-SYNC
C
, 0.1µF
B1
M2
BG1
BG2
D1
D2
f
PLLIN
SENSE1
PGND
IN
+
+
SENSE2
R
R
SENSE2
0.01Ω
SENSE1
1000pF
1000pF
0.01Ω
–
–
SENSE1
SENSE2
V
V
3.3V
5A
OUT1
5V
5A
OUT2
V
V
OSENSE2
OSENSE1
R2
105k
1%
R4
63.4k
1%
I
I
TH1
TH2
C
C
C2
C
C1
220pF
C
56µF
6V
OUT1
OUT
RUN/SS1
SGND
RUN/SS2
+
+
R1
20k
1%
R3
20k
1%
220pF
R
C2
47µF
6V
R
C1
C
C
SS1
0.1µF
SS2
0.1µF
15k
15k
SP
SP
M1, M2, M3, M4: FDS6680A
1628 F01
Figure 1. High Efficiency Dual 5V/3.3V Step-Down Converter
1
LTC1628-SYNC
W W
U W
U W
U
ABSOLUTE AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Supply Voltage (VIN).........................36V to –0.3V
ORDER PART
NUMBER
TOP VIEW
Top Side Driver Voltages
1
2
PGOOD
TG1
28
27
26
25
24
23
22
21
20
19
18
17
16
15
RUN/SS1
(BOOST1, BOOST2) ...................................42V to –0.3V
Switch Voltage (SW1, SW2) .........................36V to –5V
INTVCC, EXTVCC, RUN/SS1, RUN/SS2, (BOOST1-SW1),
(BOOST2-SW2), PGOOD .............................7V to –0.3V
SENSE1+, SENSE2+, SENSE1–,
+
SENSE1
LTC1628CG-SYNC
LTC1628IG-SYNC
–
3
SW1
SENSE1
4
BOOST1
V
OSENSE1
5
V
IN
PLLFLTR
PLLIN
FCB
6
BG1
SENSE2– Voltages........................ (1.1)INTVCC to –0.3V
PLLIN, PLLFLTR, FCB, Voltage ............ INTVCC to –0.3V
7
EXTV
CC
8
INTV
CC
I
TH1
9
PGND
BG2
SGND
ITH1, TH2, VOSENSE1, VOSENSE2 Voltages ...2.7V to –0.3V
I
10
11
12
13
14
3.3V
OUT
Peak Output Current <10µs (TG1, TG2, BG1, BG2) ... 3A
INTVCC Peak Output Current ................................ 50mA
Operating Temperature Range
BOOST2
SW2
I
TH2
V
OSENSE2
–
TG2
SENSE2
SENSE2
LTC1628CG-SYNC................................... 0°C to 85°C
LTC1628IG-SYNC............................... –40°C to 85°C
Junction Temperature (Note 2)............................. 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
+
RUN/SS2
G PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 95°C/W
*PGOOD ON THE LTC1628-SYNC
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loops
V
Regulated Feedback Voltage
Feedback Current
(Note 3); I
(Note 3)
Voltage = 1.2V
TH1, 2
●
0.792
0.800
–5
0.808
–50
V
nA
OSENSE1, 2
I
VOSENSE1, 2
V
V
Reference Voltage Line Regulation
Output Voltage Load Regulation
V
= 3.6V to 30V (Note 3)
IN
0.002
0.02
%/V
REFLNREG
LOADREG
(Note 3)
Measured in Servo Loop; ∆I Voltage = 1.2V to 0.7V
Measured in Servo Loop; ∆I Voltage = 1.2V to 2.0V
●
●
0.1
–0.1
0.5
–0.5
%
%
TH
TH
g
g
Transconductance Amplifier g
I
I
= 1.2V; Sink/Source 5uA; (Note 3)
= 1.2V; (Note 3)
1.3
3
mmho
MHz
m1, 2
m
TH1, 2
TH1, 2
Transconductance Amplifier GBW
mGBW1, 2
I
Input DC Supply Current
Normal Mode
(Note 4)
IN
RUN/SS1, 2
Q
V
V
= 15V; EXTV Tied to V ; V = 5V
OUT1 OUT1
350
20
µA
µA
CC
Shutdown
= 0V, V
= Open;
35
0.84
–0.1
4.8
STBYMD
V
Forced Continuous Threshold
Forced Continuous Pin Current
●
0.76
0.800
–0.18
4.3
V
µA
V
FCB
I
V
= 0.85V
–0.30
FCB
FCB
V
Burst Inhibit (Constant Frequency)
Threshold
Measured at FCB pin
BINHIBIT
UVLO
Undervoltage Lockout
V
Ramping Down
●
●
3.5
0.86
–60
4
V
V
IN
V
Feedback Overvoltage Lockout
Sense Pins Total Source Current
Measured at V
0.84
–85
0.88
OVL
OSENSE1, 2
I
(Each Channel); V
–
– = V + + = 0V
SENSE1 , 2
µA
SENSE
SENSE1 , 2
2
LTC1628-SYNC
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
98
TYP
99.4
1.2
1.5
4.1
2
MAX
UNITS
%
DF
MAX
Maximum Duty Factor
In Dropout
I
Soft-Start Charge Current
V
V
V
= 1.9V
RUN/SS1, 2
0.5
1.0
µA
V
RUN/SS1, 2
V
V
ON RUN/SS Pin ON Threshold
LT RUN/SS Pin Latchoff Arming Threshold
RUN/SS Discharge Current
V Rising
RUN/SS1, RUN/SS2
1.9
4.5
4
RUN/SS1, 2
RUN/SS1, 2
SCL1, 2
V
Rising from 3V
V
RUN/SS1, RUN/SS2
I
Soft Short Condition V
= 0.5V;
0.5
µA
OSENSE1, 2
V
= 4.5V
RUN/SS1, 2
I
Shutdown Latch Disable Current
Maximum Current Sense Threshold
V
= 0.5V
1.6
5
µA
SDLHO
OSENSE1, 2
V
V
V
= 0.7V,V
= 0.7V,V
–
SENSE1 , 2
SENSE1 , 2
–
–
= 5V
= 5V
65
62
75
75
85
88
mV
mV
SENSE(MAX)
OSENSE1, 2
OSENSE1, 2
–
●
TG Transition Time:
Rise Time
Fall Time
(Note 5)
TG1, 2 t
TG1, 2 t
C
C
= 3300pF
50
50
90
90
ns
ns
r
f
LOAD
LOAD
= 3300pF
BG Transition Time:
Rise Time
Fall Time
(Note 5)
LOAD
LOAD
BG1, 2 t
BG1, 2 t
C
C
= 3300pF
= 3300pF
40
40
90
80
ns
ns
r
f
TG/BG t
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
1D
C
C
= 3300pF Each Driver
= 3300pF Each Driver
90
ns
LOAD
LOAD
BG/TG t
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
2D
90
ns
ns
t
Minimum On-Time
Tested with a Square Wave (Note 6)
180
ON(MIN)
INTV Linear Regulator
CC
INTVCC
V
V
V
V
V
Internal V Voltage
6V < V < 30V, V = 4V
4.8
4.5
5.0
0.2
80
5.2
1.0
160
V
%
CC
IN
EXTVCC
INT
INTV Load Regulation
I
I
I
= 0 to 20mA, V
= 4V
LDO
LDO
CC
CC
CC
CC
EXTVCC
EXT
EXTV Voltage Drop
= 20mA, V
= 5V
mV
V
CC
EXTVCC
EXTV Switchover Voltage
= 20mA, EXTV Ramping Positive
●
4.7
0.2
EXTVCC
LDOHYS
CC
CC
EXTV Hysteresis
V
CC
Oscillator and Phase-Locked Loop
f
f
f
Nominal Frequency
Lowest Frequency
Highest Frequency
PLLIN Input Resistance
V
V
V
= 1.2V
= 0V
190
120
280
220
140
310
50
250
160
360
kHz
kHz
kHz
kΩ
NOM
LOW
HIGH
PLLFLTR
PLLFLTR
PLLFLTR
≥ 2.4V
R
PLLIN
I
Phase Detector Output Current
Sinking Capability
Sourcing Capability
PLLFLTR
f
f
< f
OSC
> f
OSC
–15
15
µA
µA
PLLIN
PLLIN
3.3V Linear Regulator
V
V
V
3.3V Regulator Output Voltage
3.3V Regulator Load Regulation
3.3V Regulator Line Regulation
No Load
= 0 to 10mA
●
3.25
3.35
0.5
3.45
2
V
%
%
3.3OUT
3.3IL
I
3.3
6V < V < 30V
0.05
0.2
3.3VL
IN
PGOOD Output
V
PGOOD Voltage Low
I
= 2mA
= 5V
0.1
0.3
V
PGL
PGOOD
I
PGOOD Leakage Current
PGOOD Trip Level, Either Controller
V
V
±1
µA
PGOOD
PGOOD
V
with Respect to Set Output Voltage
Ramping Negative
Ramping Positive
PG
OSENSE
V
V
–6
6
–7.5
7.5
–9.5
9.5
%
%
OSENSE
OSENSE
3
LTC1628-SYNC
ELECTRICAL CHARACTERISTICS
Note 1: Absolute Maximum Ratings are those values beyond which the life
Note 4: Dynamic supply current is higher due to the gate charge being
of a device may be impaired.
delivered at the switching frequency. See Applications Information.
Note 2: T is calculated from the ambient temperature T and power
Note 5: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
J
A
dissipation P according to the following formulas:
D
LTC1628-SYNC: T = T + (P • 95 °C/W)
Note 3: The LTC1628-SYNC is tested in a feedback loop that servos V
to a specified voltage and measures the resultant
Note 6: The minimum on-time condition is specified for an inductor
J
A
D
peak-to-peak ripple current ≥40% of I
(see minimum on-time
MAX
ITH1,
considerations in the Applications Information section).
2
V
OSENSE1, 2.
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Output Current
and Mode (Figure 14)
Efficiency vs Output Current
(Figure 14)
Efficiency vs Input Voltage
(Figure 14)
100
90
80
70
60
50
40
30
20
10
0
100
90
80
70
60
50
100
90
80
70
60
50
V
= 5V
= 3A
Burst Mode
OPERATION
OUT
OUT
V
= 7V
IN
I
V
IN
= 10V
FORCED
CONTINUOUS
MODE
V
= 15V
IN
V
= 20V
IN
CONSTANT
FREQUENCY
(BURST DISABLE)
V
OUT
= 15V
= 5V
IN
V
= 5V
V
OUT
0.001
0.01
0.1
1
10
0.001
0.01
0.1
1
5
15
25
35
10
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
INPUT VOLTAGE (V)
1628 G01
1628 G02
1628 G03
Supply Current vs Input Voltage
and Mode (Figure 14)
INTVCC and EXTVCC Switch
Voltage vs Temperature
EXTVCC Voltage Drop
1000
800
600
400
200
0
250
200
150
100
50
5.05
5.00
4.95
4.90
4.85
4.80
4.75
4.70
INTV VOLTAGE
CC
BOTH
CONTROLLERS ON
EXTV SWITCHOVER THRESHOLD
CC
STANDBY
SHUTDOWN
0
0
5
10
15
20
25
30
35
0
10
20
30
40
50
–50 –25
0
25
50
TEMPERATURE (°C)
75
100 125
INPUT VOLTAGE (V)
CURRENT (mA)
1628 G05
1628 G04
1628 G06
4
LTC1628-SYNC
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Maximum Current Sense Threshold
vs Percent of Nominal Output
Voltage (Foldback)
Maximum Current Sense Threshold
Internal 5V LDO Line Regulation
vs Duty Factor
75
50
25
0
5.1
5.0
80
70
60
50
40
30
20
10
0
I
= 1mA
LOAD
4.9
4.8
4.7
4.6
4.5
4.4
0
20
40
60
80
100
50
20
INPUT VOLTAGE (V)
30
35
0
25
75
100
0
5
10
15
25
DUTY FACTOR (%)
PERCENT ON NOMINAL OUTPUT VOLTAGE (%)
1628 G08
1628 G09
1628 G07
Maximum Current Sense Threshold
vs VRUN/SS (Soft-Start)
Maximum Current Sense Threshold
vs Sense Common Mode Voltage
Current Sense Threshold
vs ITH Voltage
90
80
80
76
72
68
64
60
80
60
40
20
V
= 1.6V
SENSE(CM)
70
60
50
40
30
20
10
0
–10
–20
–30
0
0
1
2
3
4
5
6
0
1
2
3
4
5
0
0.5
1
1.5
(V)
2
2.5
V
(V)
COMMON MODE VOLTAGE (V)
V
ITH
RUN/SS
1628 G10
1628 G11
1628 G12
Load Regulation
VITH vs VRUN/SS
SENSE Pins Total Source Current
0.0
–0.1
–0.2
–0.3
–0.4
2.5
2.0
1.5
1.0
100
50
V
= 0.7V
FCB = 0V
= 15V
OSENSE
V
IN
FIGURE 1
0
–50
–100
0.5
0
0
1
2
3
LOAD CURRENT (A)
4
5
0
1
2
3
4
5
6
0
2
4
6
V
(V)
V
COMMON MODE VOLTAGE (V)
RUN/SS
SENSE
1628 G14
1628 G13
1628 G15
5
LTC1628-SYNC
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Maximum Current Sense
Threshold vs Temperature
Dropout Voltage vs Output Current
(Figure 14)
RUN/SS Current vs Temperature
80
78
76
74
72
70
4
3
2
1
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
V
= 5V
OUT
R
= 0.015Ω
SENSE
R
= 0.010Ω
SENSE
0
0
50
75 100 125
0
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0
OUTPUT CURRENT (A)
–50 –25
0
25
–50 –25
0
25
125
50
75 100
TEMPERATURE (°C)
TEMPERATURE (°C)
1628 G18
1628 G17
1628 G25
Soft-Start Up (Figure 14)
Load Step (Figure 14)
Load Step (Figure 14)
VOUT
5V/DIV
VOUT
200mV/DIV
VOUT
200mV/DIV
VRUN/SS
5V/DIV
IOUT
2A/DIV
IOUT
2A/DIV
IOUT
2A/DIV
V
IN = 15V
5ms/DIV
1628 G19
V
IN = 15V
20µs/DIV
1628 G20
VIN = 15V
VOUT = 5V
20µs/DIV
1628 G21
VOUT = 5V
VOUT = 5V
LOAD STEP = 0A TO 3A
Burst Mode OPERATION
LOAD STEP = 0A TO 3A
CONTINUOUS MODE
Input Source/Capacitor
Instantaneous Current (Figure 14)
Constant Frequency (Burst Inhibit)
Operation (Figure 14)
Burst Mode Operation (Figure 14)
IIN
VOUT
20mV/DIV
2A/DIV
VOUT
20mV/DIV
VIN
200mV/DIV
VSW1
10V/DIV
VSW2
10V/DIV
IOUT
0.5A/DIV
IOUT
0.5A/DIV
VIN = 15V
1µs/DIV
1628 G22
VIN = 15V
10µs/DIV
1628 G23
VIN = 15V
VOUT = 5V
VFCB = 5V
IOUT = 20mA
2µs/DIV
1628 G24
VOUT = 5V
VOUT = 5V
IOUT5 = IOUT3.3 = 2A
VFCB = OPEN
IOUT = 20mA
6
LTC1628-SYNC
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Current Sense Pin Input Current
vs Temperature
EXTVCC Switch Resistance
vs Temperature
Oscillator Frequency
vs Temperature
35
33
31
29
27
25
10
8
350
300
V
OUT
= 5V
V
= 5V
PLLFLTR
250
200
150
100
50
V
= OPEN
= 0V
PLLFLTR
6
V
PLLFLTR
4
2
0
0
–50 –25
0
25
50
75 100 125
–50 –25
0
25
50
75 100 125
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
75
TEMPERATURE (°C)
TEMPERATURE (°C)
1628 G26
1628 G27
1628 G28
Undervoltage Lockout
vs Temperature
Shutdown Latch Thresholds
vs Temperature
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
3.50
3.45
3.40
3.35
LATCH ARMING
LATCHOFF
THRESHOLD
3.30
3.25
3.20
0
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
125
–50 –25
0
25
75
50
75 100
TEMPERATURE (°C)
1628 G29
1628 G30
7
LTC1628-SYNC
U
U
U
PI FU CTIO S
RUN/SS1, RUN/SS2 (Pins 1, 15): Combination of soft-
start,runcontrolinputsandshort-circuitdetectiontimers.
A capacitor to ground at each of these pins sets the ramp
time to full output current. Forcing either of these pins
back below 1.0V causes the IC to shut down the circuitry
requiredforthatparticularcontroller.Latchoffovercurrent
protection is also invoked via this pin as described in the
Applications Information section.
SENSE1+, SENSE2+ (Pins 2, 14): The (+) Input to the
Differential Current Comparators. The Ith pin voltage and
controlled offsets between the SENSE– and SENSE+ pins
in conjunction with RSENSE set the current trip threshold.
3.3VOUT (Pin 10): Output of a linear regulator capable of
supplying 10mA DC with peak currents as high as 50mA.
PGND (Pin 20): Driver Power Ground. Connects to the
sources of bottom (synchronous) N-channel MOSFETs, an-
odes of the Schottky rectifiers and the (–) terminal(s) of CIN.
INTVCC (Pin 21): Output of the Internal 5V Linear Low
Dropout Regulator and the EXTVCC Switch. The driver and
control circuits are powered from this voltage source. Must
be decoupled to power ground with a minimum of 4.7µF
tantalum or other low ESR capacitor.
EXTVCC (Pin 22): External Power Input to an Internal
Switch Connected to INTVCC. This switch closes and
supplies VCC power, bypassing the internal low dropout
regulator, whenever EXTVCC is higher than 4.7V. See
EXTVCC connectioninApplicationssection. Donotexceed
7V on this pin.
SENSE1–, SENSE2– (Pins 3, 13): The (–) Input to the
Differential Current Comparators.
VOSENSE1, VOSENSE2 (Pins 4, 12): Receives the remotely-
sensed feedback voltage for each controller from an
external resistive divider across the output.
BG1, BG2 (Pins 23, 19): High Current Gate Drives for
Bottom (Synchronous) N-Channel MOSFETs. Voltage
swing at these pins is from ground to INTVCC.
PLLFLTR (Pin 5): The Phase-Locked Loop’s Lowpass
Filter is Tied to This Pin. Alternatively, this pin can be
driven with an AC or DC voltage source to vary the
frequency of the internal oscillator.
VIN (Pin 24): Main Supply Pin. A bypass capacitor should
be tied between this pin and the signal ground pin.
PLLIN (Pin 6): External Synchronization Input to Phase
Detector. This pin is internally terminated to SGND with
50kΩ. The phase-locked loop will force the rising top gate
signal of controller 1 to be synchronized with the rising
edge of the PLLIN signal.
BOOST1, BOOST2 (Pins 25, 18): Bootstrapped Supplies
to the Top Side Floating Drivers. Capacitors are connected
between the boost and switch pins and Schottky diodes
aretied betweentheboostandINTVCCpins.Voltageswing
at the boost pins is from INTVCC to (VIN + INTVCC).
FCB(Pin7):ForcedContinuousControl Input. Thisinput
acts on both controllers and is normally used to regulate
a secondary winding. Pulling this pin below 0.8V will
force continuous synchronous operation. Do not leave
this pin floating.
SW1, SW2 (Pins 26, 17): Switch Node Connections to
Inductors. Voltage swing at these pins is from a Schottky
diode (external) voltage drop below ground to VIN.
TG1, TG2 (Pins 27, 16): High Current Gate Drives for Top
N-Channel MOSFETs. These are the outputs of floating
drivers with a voltage swing equal to INTVCC – 0.5V
superimposed on the switch node voltage SW.
ITH1, TH2 (Pins 8, 11): Error Amplifier Output and Switch-
I
ingRegulatorCompensationPoint.Eachassociatedchan-
nels’ current comparator trip point increases with this
control voltage.
PGOOD (Pin 28): Open-Drain Logic Output. PGOOD is
pulled to ground when the voltage on either VOSENSE pin is
not within ±7.5% of its set point.
SGND (Pin 9): Small Signal Ground common to both
controllers, must be routed separately from high current
grounds to the common (–) terminals of the COUT
capacitors.
8
LTC1628-SYNC
U
U
W
FU CTIO AL DIAGRA
PLLIN
INTV
V
IN
CC
F
PHASE DET
IN
D
C
B
DUPLICATE FOR SECOND
CONTROLLER CHANNEL
50k
BOOST
TG
PLLFLTR
B
DROP
OUT
+
CLK1
CLK2
–
TOP
BOT
R
LP
C
IN
D
OSCILLATOR
1
DET
BOT
FCB
C
LP
SW
TOP ON
0.86V
S
Q
Q
+
SWITCH
LOGIC
INTV
CC
R
V
OSENSE1
PGOOD
BG
–
+
0.74V
0.86V
C
OUT
PGND
B
+
–
0.55V
–
+
V
OUT
SHDN
R
SENSE
V
OSENSE2
–
+
INTV
CC
0.74V
BINH
I1
I2
V
SEC
3V
+
–
+
–
+
4.5V
0.8V
–
+ +
–
+
0.18µA
FCB
–
SENSE
SENSE
D
SEC
C
SEC
30k
30k
R6
3mV
0.86V
4(V
–
)
FB
+
–
FCB
R5
SLOPE
COMP
45k
45k
2.4V
3.3V
OUT
V
OSENSE
R2
V
FB
+
–
V
REF
–
EA
+
0.80V
0.86V
R1
OV
V
IN
+
–
V
IN
C
C
+
–
4.8V
I
TH
5V
1.2µA
EXTV
INTV
LDO
REG
CC
SHDN
RST
RUN
SOFT
START
R
C
C
C2
6V
4(V
)
FB
CC
5V
+
RUN/SS
INTERNAL
SUPPLY
SGND
C
SS
1628 FD/F02
Figure 2
U
OPERATIO
(Refer to Functional Diagram)
Main Control Loop
The peak inductor current at which I1 resets the RS latch
is controlled by the voltage on the ITH pin, which is the
outputofeacherroramplifierEA.TheVOSENSE pinreceives
the voltage feedback signal, which is compared to the
internalreferencevoltagebytheEA. Whentheloadcurrent
increases, itcausesaslightdecreaseinVOSENSErelativeto
the 0.8V reference, which in turn causes the ITH voltage to
The LTC1628-SYNC uses a constant frequency, current
mode step-down architecture with the two controller
channels operating 180 degrees out of phase. During
normaloperation, eachtopMOSFETisturnedonwhenthe
clock for that channel sets the RS latch, and turned off
when the main current comparator, I1, resets the RS latch.
9
LTC1628-SYNC
U
OPERATIO
(Refer to Functional Diagram)
increase until the average inductor current matches the
new load current. After the top MOSFET has turned off, the
bottom MOSFET is turned on until either the inductor
currentstartstoreverse, asindicatedbycurrentcompara-
tor I2, or the beginning of the next cycle.
the output voltage drops. There is 60mV of hysteresis in
the burst comparator B tied to the ITH pin. This hysteresis
produces output signals to the MOSFETs that turn them
on for several cycles, followed by a variable “sleep”
interval depending upon the load current. The resultant
output voltage ripple is held to a very small value by
having the hysteretic comparator after the error amplifier
gain block.
The top MOSFET drivers are biased from floating boot-
strap capacitor CB, which normally is recharged during
each off cycle through an external diode when the top
MOSFET turns off. As VIN decreases to a voltage close to
VOUT, the loop may enter dropout and attempt to turn on
the top MOSFET continuously. The dropout detector de-
tects this and forces the top MOSFET off for about 400ns
every tenth cycle to allow CB to recharge.
Frequency Synchronization
The phase-locked loop allows the internal oscillator to be
synchronized to an external source via the PLLIN pin. The
output of the phase detector at the PLLFLTR pin is also the
DC frequency control input of the oscillator that operates
over a 140kHz to 310kHz range corresponding to a DC
voltageinputfrom0Vto2.4V.Whenlocked,thePLLaligns
the turn on of the top MOSFET to the rising edge of the
synchronizingsignal.WhenPLLINisleftopen,thePLLFLTR
pingoeslow,forcingtheoscillatortominimumfrequency.
The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 1.2µA
current source to charge soft-start capacitor CSS. When
CSS reaches1.5V,themaincontrolloopisenabledwiththe
ITH voltageclampedatapproximately30%ofitsmaximum
value. As CSS continues to charge, the ITH pin voltage is
gradually released allowing normal, full-current opera-
tion. When both RUN/SS1 and RUN/SS2 are low, all
LTC1628-SYNC controller functions are shut down,
including the 5V and 3.3V regulators.
Continuous Current (PWM) Operation
Tying the FCB pin to ground will force continuous current
operation. This is the least efficient operating mode, but
may be desirable in certain applications. The output can
source or sink current in this mode. When sinking current
while in forced continuous operation, current will be
forced back into the main power supply potentially boost-
ing the input supply to dangerous voltage levels—
BEWARE!
Low Current Operation
The FCB pin is a multifunction pin providing two func-
tions:1)toprovideregulationforasecondarywindingby
temporarily forcing continuous PWM operation on
bothcontrollers;and2)selectbetweentwomodesoflow
current operation. When the FCB pin voltage is below
0.8V, the controller forces continuous PWM current
mode operation. In this mode, the top and bottom
MOSFETsarealternatelyturnedontomaintaintheoutput
voltage independent of direction of inductor current.
When the FCB pin is below VINTVCC – 2V but greater than
0.8V, the controller enters Burst Mode operation. Burst
Mode operation sets a minimum output current level
beforeinhibitingthetopswitchandturnsoffthesynchro-
nous MOSFET(s) when the inductor current goes nega-
tive. This combination of requirements will, at low cur-
rents, force the ITH pin below a voltage threshold that will
temporarily inhibit turn-on of both output MOSFETs until
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin.
When the EXTVCC pin is left open, an internal 5V low
dropoutlinearregulatorsuppliesINTVCC power.IfEXTVCC
is taken above 4.7V, the 5V regulator is turned off and an
internalswitchisturnedonconnectingEXTVCC toINTVCC.
This allows the INTVCC power to be derived from a high
efficiency external source such as the output of the regu-
lator itself or a secondary winding, as described in the
Applications Information section.
10
LTC1628-SYNC
U
OPERATIO
(Refer to Functional Diagram)
Output Overvoltage Protection
THEORY AND BENEFITS OF 2-PHASE OPERATION
An overvoltage comparator, OV, guards against transient
overshoots (>7.5%) as well as other more serious condi-
tions that may overvoltage the output. In this case, the top
MOSFETisturnedoffandthebottomMOSFETisturnedon
until the overvoltage condition is cleared.
The LTC1628-SYNC dual high efficiency DC/DC controller
brings the considerable benefits of 2-phase operation to
portable applications for the first time. Notebook comput-
ers, PDAs, handheldterminalsandautomotiveelectronics
will all benefit from the lower input filtering requirement,
reducedelectromagneticinterference(EMI)andincreased
efficiency associated with 2-phase operation.
Power Good (PGOOD) Pin
ThePGOODpinisconnectedtoanopendrainofaninternal
MOSFET.TheMOSFETturnsonandpullsthepinlowwhen
either output is not within ±7.5% of the nominal output
level as determined by the resistive feedback divider.
When both outputs meet the ±7.5% requirement, the
MOSFET is turned off within 10µs and the pin is allowed to
be pulled up by an external resistor to a source of up to 7V.
Whytheneedfor2-phaseoperation?UpuntiltheLTC1628
family, constant-frequency dual switching regulators op-
erated both channels in phase (i.e., single-phase opera-
tion). Thismeansthatbothswitchesturnedonatthesame
time, causing current pulses of up to twice the amplitude
of those for one regulator to be drawn from the input
capacitorandbattery.Theselargeamplitudecurrentpulses
increased the total RMS current flowing from the input
capacitor, requiring the use of more expensive input
capacitorsandincreasingbothEMIandlossesintheinput
capacitor and battery.
Foldback Current, Short-Circuit Detection
and Short-Circuit Latchoff
TheRUN/SScapacitorsareusedinitiallytolimittheinrush
current of each switching regulator. After the controller
has been started and been given adequate time to charge
up the output capacitors and provide full load current, the
RUN/SS capacitor is used in a short-circuit time-out
circuit. If the output voltage falls to less than 70% of its
nominal output voltage, the RUN/SS capacitor begins
discharging on the assumption that the output is in an
overcurrent and/or short-circuit condition. If the condi-
tion lasts for a long enough period as determined by the
size of the RUN/SS capacitor, the controller will be shut
down until the RUN/SS pin(s) voltage(s) are recycled.
This built-in latchoff can be overridden by providing a
>5µA pull-up at a compliance of 5V to the RUN/SS pin(s).
This current shortens the soft start period but also pre-
vents net discharge of the RUN/SS capacitor(s) during an
overcurrent and/or short-circuit condition. Foldback cur-
rent limiting is also activated when the output voltage falls
below 70% of its nominal level whether or not the short-
circuit latchoff circuit is enabled. Even if a short is present
and the short-circuit latchoff is not enabled, a safe, low
outputcurrentisprovidedduetointernalcurrentfoldback
and actual power wasted is low due to the efficient nature
of the current mode switching regulator.
With 2-phase operation, the two channels of the dual-
switching regulator are operated 180 degrees out of
phase. This effectively interleaves the current pulses
drawn by the switches, greatly reducing the overlap time
where they add together. The result is a significant reduc-
tion in total RMS input current, which in turn allows less
expensive input capacitors to be used, reduces shielding
requirements for EMI and improves real world operating
efficiency.
Figure 3 compares the input waveforms for a representa-
tive single-phase dual switching regulator to the new
LTC1628-SYNC 2-phase dual switching regulator. An
actual measurement of the RMS input current under these
conditions shows that 2-phase operation dropped the
input current from 2.53ARMS to 1.55ARMS. While this is an
impressive reduction in itself, remember that the power
losses are proportional to IRMS2, meaning that the actual
power wasted is reduced by a factor of 2.66. The reduced
input ripple voltage also means less power is lost in the
inputpowerpath, whichcouldincludebatteries, switches,
trace/connector resistances and protection circuitry. Im-
provements in both conducted and radiated EMI also
directly accrue as a result of the reduced RMS input
current and voltage.
11
LTC1628-SYNC
U
OPERATIO
(Refer to Functional Diagram)
5V SWITCH
20V/DIV
3.3V SWITCH
20V/DIV
INPUT CURRENT
5A/DIV
INPUT VOLTAGE
500mV/DIV
IIN(MEAS) = 2.53ARMS
IIN(MEAS) = 1.55ARMS
DC236 F03a
DC236 F03b
(b)
(a)
Figure 3. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for
Dual Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input
Ripple with the LTC1628-SYNC 2-Phase Regulator Allows Less Expensive Input Capacitors,
Reduces Shielding Requirements for EMI and Improves Efficiency
Of course, the improvement afforded by 2-phase opera-
operation. In addition, isolation between the two channels
becomes more critical with 2-phase operation because
switch transitions in one channel could potentially disrupt
the operation of the other channel.
tion is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
voltage VIN (Duty Cycle = VOUT/VIN). Figure 4 shows how
theRMSinputcurrentvariesforsingle-phaseand2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.
The LTC1628-SYNC is proof that these hurdles have been
surmounted. Thenewdeviceoffersuniqueadvantagesfor
the ever-expanding number of high efficiency power sup-
plies required in portable electronics.
It can readily be seen that the advantages of 2-phase
operation are not just limited to a narrow operating range,
but in fact extend over a wide region. A good rule of thumb
for most applications is that 2-phase operation will reduce
theinputcapacitorrequirementtothatforjustonechannel
operating at maximum current and 50% duty cycle.
3.0
SINGLE PHASE
DUAL CONTROLLER
2.5
2.0
A final question: If 2-phase operation offers such an
advantage over single-phase operation for dual switching
regulators, why hasn’t it been done before? The answer is
that, while simple in concept, it is hard to implement.
Constant-frequency current mode switching regulators
require an oscillator derived “slope compensation” signal
to allow stable operation of each regulator at over 50%
duty cycle. This signal is relatively easy to derive in single-
phasedualswitchingregulators,butrequiredthedevelop-
ment of a new and proprietary technique to allow 2-phase
1.5
2-PHASE
DUAL CONTROLLER
1.0
0.5
V
V
= 5V/3A
O1
O2
= 3.3V/3A
0
0
10
20
INPUT VOLTAGE (V)
30
40
1628 F04
Figure 4. RMS Input Current Comparison
12
LTC1628-SYNC
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APPLICATIO S I FOR ATIO
U
2.5
Figure 1 on the first page is a basic LTC1628-SYNC
applicationcircuit.Externalcomponentselectionisdriven
by the load requirement, and begins with the selection of
2.0
1.5
1.0
0.5
0
RSENSE andtheinductorvalue.Next,thepowerMOSFETs
and D1 are selected. Finally, CIN and COUT are selected.
The circuit shown in Figure 1 can be configured for
operation up to an input voltage of 28V (limited by the
external MOSFETs).
RSENSE Selection For Output Current
120
170
220
270
320
RSENSE is chosen based on the required output current.
OPERATING FREQUENCY (kHz)
1628 F05
The LTC1628-SYNC current comparator has a maximum
threshold of 75mV/RSENSE and an input common mode
range of SGND to 1.1(INTVCC). The current comparator
threshold sets the peak of the inductor current, yielding a
maximum average output current IMAX equal to the peak
value less half the peak-to-peak ripple current, ∆IL.
Figure 5. PLLFLTR Pin Voltage vs Frequency
isincreasedthegatechargelosseswillbehigher,reducing
efficiency (see Efficiency Considerations). The maximum
switching frequency is approximately 310kHz.
AllowingamarginforvariationsintheLTC1628-SYNCand
external component values yields:
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFET gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
50mV
R
=
SENSE
I
MAX
When using the controller in very low dropout conditions,
the maximum output current level will be reduced due to
the internal compensation required to meet stability crite-
rion for buck regulators operating at greater than 50%
duty factor. A curve is provided to estimate this reducton
in peak output current level depending upon the operating
duty factor.
Theinductorvaluehasadirecteffectonripplecurrent.The
inductor ripple current ∆IL decreases with higher induc-
tance or frequency and increases with higher VIN:
Operating Frequency
The LTC1628-SYNC uses a constant frequency phase-
lockable architecture with the frequency determined by an
internal capacitor. This capacitor is charged by a fixed
current plus an additional current which is proportional to
the voltage applied to the PLLFLTR pin. Refer to Phase-
Locked Loop and Frequency Synchronization in the Appli-
cations Information section for additional information.
1
V
OUT
V
IN
∆I =
V
1–
L
OUT
(f)(L)
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL=0.3(IMAX). The maximum ∆IL
occurs at the maximum input voltage.
A graph for the voltage applied to the PLLFLTR pin vs
frequency is given in Figure 5. As the operating frequency
13
LTC1628-SYNC
W U U
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APPLICATIO S I FOR ATIO
The inductor value also has secondary effects. The transi-
tion to Burst Mode operation begins when the average
inductor current required results in a peak current below
25% of the current limit determined by RSENSE. Lower
inductor values (higher ∆IL) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
The peak-to-peak drive levels are set by the INTVCC
voltage. This voltage is typically 5V during start-up (see
EXTVCC Pin Connection). Consequently, logic-level
threshold MOSFETs must be used in most applications.
The only exception is if low input voltage is expected
(VIN < 5V); then, sub-logic level threshold MOSFETs
(VGS(TH) < 3V) should be used. Pay close attention to the
BVDSS specification for the MOSFETs as well; most of the
logic level MOSFETs are limited to 30V or less.
SelectioncriteriaforthepowerMOSFETsincludethe“ON”
Inductor Core Selection
resistance RDS(ON), reverse transfer capacitance CRSS
,
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron
cores, forcing the use of more expensive ferrite,
molypermalloy, or Kool Mµ® cores. Actual core loss is
independent of core size for a fixed inductor value, but it
is very dependent on inductance selected. As inductance
increases, core losses go down. Unfortunately, increased
inductance requires more turns of wire and therefore
copper losses will increase.
input voltage and maximum output current. When the
LTC1628-SYNC is operating in continuous mode the duty
cycles for the top and bottom MOSFETs are given by:
V
V
OUT
Main SwitchDuty Cycle =
IN
V – V
IN
OUT
Synchronous SwitchDuty Cycle =
V
IN
The MOSFET power dissipations at maximum output
current are given by:
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that induc-
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
2
) (
V
V
OUT
P
=
I
1+δ R
+
(
)
MAIN
MAX
DS(ON)
IN
2
k V
I
C
f
(
) (
)(
)( )
IN
MAX RSS
Molypermalloy (from Magnetics, Inc.) is a very good, low
losscorematerialfortoroids,butitismoreexpensivethan
ferrite. A reasonable compromise from the same manu-
facturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Be-
cause they generally lack a bobbin, mounting is more
difficult. However, designsforsurfacemountareavailable
that do not increase the height significantly.
2
) (
V – V
IN
OUT
P
=
I
(
1+δ R
)
SYNC
MAX
DS(ON)
V
IN
where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
BothMOSFETshaveI2RlosseswhilethetopsideN-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V the
high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increasetothepointthattheuseofahigherRDS(ON)device
with lower CRSS actually provides higher efficiency. The
Power MOSFET and D1 Selection
Two external power MOSFETs must be selected for each
controller in the LTC1628-SYNC: One N-channel MOSFET
for the top (main) switch, and one N-channel MOSFET for
the bottom (synchronous) switch.
Kool Mµ is a registered trademark of Magnetics, Inc.
14
LTC1628-SYNC
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APPLICATIO S I FOR ATIO
U
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during a
short-circuit when the synchronous switch is on close to
100% of the period.
the capacitor is important for capacitor power dissipation
as well as overall battery efficiency. All of the power (RMS
ripple current • ESR) not only heats up the capacitor but
wastes power from the battery.
Theterm(1+δ)isgenerallygivenforaMOSFETintheform
of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the MOS-
FET characteristics. The constant k = 1.7 can be used to
estimate the contributions of the two terms in the main
switch dissipation equation.
Medium voltage (20V to 35V) ceramic, tantalum, OS-CON
and switcher-rated electrolytic capacitors can be used as
inputcapacitors,buteachhasdrawbacks:ceramicvoltage
coefficients are very high and may have audible piezoelec-
tric effects; tantalums need to be surge-rated; OS-CONs
suffer from higher inductance, larger case size and limited
surface-mount applicability; electrolytics’ higher ESR and
dryout possibility require several to be used. Multiphase
systems allow the lowest amount of capacitance overall.
As little as one 22µF or two to three 10µF ceramic capaci-
tors are an ideal choice in a 20W to 35W power supply due
to their extremely low ESR. Even though the capacitance
at 20V is substantially below their rating at zero-bias, very
low ESR loss makes ceramics an ideal candidate for
highest efficiency battery operated systems. Also con-
sider parallel ceramic and high quality electrolytic capaci-
tors as an effective means of achieving ESR and bulk
capacitance goals.
The Schottky diode D1 shown in Figure 1 conducts during
the dead-time between the conduction of the two power
MOSFETs. This prevents the body diode of the bottom
MOSFET from turning on, storing charge during the dead-
time and requiring a reverse recovery period that could
cost as much as 3% in efficiency at high VIN. A 1A to 3A
Schottky is generally a good compromise for both regions
of operation due to the relatively small average current.
Larger diodes result in additional transition losses due to
their larger junction capacitance.
CIN and COUT Selection
Incontinuousmode, thesourcecurrentofthetopN-chan-
nel MOSFET is a square wave of duty cycle VOUT/VIN. To
preventlargevoltagetransients, alowESRinputcapacitor
sized for the maximum RMS current of one channel must
beused. ThemaximumRMScapacitorcurrentisgivenby:
The selection of CIN is simplified by the multiphase archi-
tecture and its impact on the worst-case RMS current
drawnthroughtheinputnetwork(battery/fuse/capacitor).
It can be shown that the worst case RMS current occurs
when only one controller is operating. The controller with
the highest (VOUT)(IOUT) product needs to be used in the
formula below to determine the maximum RMS current
requirement. Increasing the output current, drawn from
the other out-of-phase controller, will actually decrease
the input RMS ripple current from this maximum value
(see Figure 4). The out-of-phase technique typically re-
duces the input capacitor’s RMS ripple current by a factor
of 30% to 70% when compared to a single phase power
supply solution.
1/2
]
V
V − V
OUT
(
)
OUT IN
[
C RequiredI
≈I
IN
RMS MAX
V
IN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst case condition is com-
monlyusedfordesignbecauseevensignificantdeviations
donotoffermuchrelief.Notethatcapacitormanufacturer’s
ripple current ratings are often based on only 2000 hours
of life. This makes it advisable to further derate the
capacitor, or to choose a capacitor rated at a higher
temperaturethanrequired.Severalcapacitorsmayalsobe
paralleled to meet size or height requirements in the
design. Always consult the manufacturer if there is any
question.
The type of input capacitor, value and ESR rating have
efficiency effects that need to be considered in the selec-
tion process. The capacitance value chosen should be
sufficient to store adequate charge to keep high peak
battery currents down. 20µF to 40µF is usually sufficient
for a 25W output supply operating at 200kHz. The ESR of
15
LTC1628-SYNC
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APPLICATIO S I FOR ATIO
The benefit of the LTC1628-SYNC multiphase can be
calculated by using the equation above for the higher
power controller and then calculating the loss that would
have resulted if both controller channels switch on at the
same time. The total RMS power lost is lower when both
controllers are operating due to the interleaving of current
pulses through the input capacitor’s ESR. This is why the
input capacitor’s requirement calculated above for the
worst-case controller is adequate for the dual controller
design. Remember that input protection fuse resistance,
battery resistance and PC board trace resistance losses
are also reduced due to the reduced peak currents in a
multiphase system. The overall benefit of a multiphase
design will only be fully realized when the source imped-
ance of the power supply/battery is included in the effi-
ciency testing. The drains of the two top MOSFETS should
be placed within 1cm of each other and share a common
CIN(s). Separating the drains and CIN may produce unde-
sirable voltage and current resonances at VIN.
capacitance increases the ripple voltage due to the dis-
charging term but can be compensated for by using
capacitors of very low ESR to maintain the ripple voltage
at or below 50mV. The ITH pin OPTI-LOOP compensation
components can be optimized to provide stable, high
performance transient response regardless of the output
capacitors selected.
Manufacturers such as Nichicon, United Chemicon and
Sanyo can be considered for high performance through-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest (ESR)(size)
product of any aluminum electrolytic at a somewhat
higher price. An additional ceramic capacitor in parallel
with OS-CON capacitors is recommended to reduce the
inductance effects.
In surface mount applications multiple capacitors may
need to be used in parallel to meet the ESR, RMS current
handling and load step requirements of the application.
Aluminum electrolytic, dry tantalum and special polymer
capacitors are available in surface mount packages. Spe-
cial polymer surface mount capacitors offer very low ESR
buthavelowerstoragecapacityperunitvolumethanother
capacitor types. These capacitors offer a very cost-effec-
tiveoutputcapacitorsolutionandareanidealchoicewhen
combined with a controller having high loop bandwidth.
Tantalum capacitors offer the highest capacitance density
and are often used as output capacitors for switching
regulators having controlled soft-start. Several excellent
surge-tested choices are the AVX TPS, AVX TPSV or the
KEMET T510 series of surface mount tantalums, available
in case heights ranging from 2mm to 4mm. Aluminum
electrolytic capacitors can be used in cost-driven applica-
tionsprovidingthatconsiderationisgiventoripplecurrent
ratings, temperature and long term reliability. A typical
application will require several to many aluminum electro-
lytic capacitors in parallel. A combination of the above
mentioned capacitors will often result in maximizing per-
formance and minimizing overall cost. Other capacitor
types include Nichicon PL series, NEC Neocap, Cornell
Dubilier ESRE and Sprague 595D series. Consult manu-
facturers for other specific recommendations.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically once the ESR require-
ment is satisfied the capacitance is adequate for filtering.
The output ripple (∆VOUT) is determined by:
1
∆VOUT ≈ ∆IL ESR +
8fCOUT
Wheref=operatingfrequency,COUT =outputcapacitance,
and ∆IL= ripple current in the inductor. The output ripple
is highest at maximum input voltage since ∆IL increases
with input voltage. With ∆IL = 0.3IOUT(MAX) the output
ripple will typically be less than 50mV at max VIN assum-
ing:
COUT Recommended ESR < 2 RSENSE
and COUT > 1/(8fRSENSE
)
The first condition relates to the ripple current into the
ESR of the output capacitance while the second term
guarantees that the output capacitance does not signifi-
cantly discharge during the operating frequency period
due to ripple current. The choice of using smaller output
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LTC1628-SYNC
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U
INTVCC Regulator
EXTVCC Connection
An internal P-channel low dropout regulator produces 5V
at the INTVCC pin from the VIN supply pin. INTVCC powers
the drivers and internal circuitry within the LTC1628-
SYNC.TheINTVCC pinregulatorcansupplyapeakcurrent
of 50mA and must be bypassed to ground with a mini-
mum of 4.7µF tantalum, 10µF special polymer, or low
ESR type electrolytic capacitor. A 1µF ceramic capacitor
placed directly adjacent to the INTVCC and PGND IC pins
is highly recommended. Good bypassing is necessary to
supplythehightransientcurrentsrequiredbythe MOSFET
gate drivers and to prevent interaction between channels.
The LTC1628-SYNC contains an internal P-channel MOS-
FET switch connected between the EXTVCC and INTVCC
pins. When the voltage applied to EXTVCC rises above
4.7V, the internal regulator is turned off and the switch
closes, connecting the EXTVCC pin to the INTVCC pin
thereby supplying internal power. The switch remains
closed as long as the voltage applied to EXTVCC remains
above 4.5V. This allows the MOSFET driver and control
power to be derived from the output during normal opera-
tion (4.7V < VOUT < 7V) and from the internal regulator
when the output is out of regulation (start-up, short-
circuit). If more current is required through the EXTVCC
switch than is specified, an external Schottky diode can be
added between the EXTVCC and INTVCC pins. Do not apply
greater than 7V to the EXTVCC pin and ensure that
EXTVCC < VIN.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maxi-
mum junction temperature rating for the LTC1628-SYNC
to be exceeded. The system supply current is normally
dominated by the gate charge current. Additional external
loading of the INTVCC and 3.3V linear regulators also
needs to be taken into account for the power dissipation
calculations. The total INTVCC current can be supplied by
either the 5V internal linear regulator or by the EXTVCC
input pin. When the voltage applied to the EXTVCC pin is
less than 4.7V, all of the INTVCC current is supplied by the
internal 5V linear regulator. Power dissipation for the IC in
this case is highest: (VIN)(IINTVCC), and overall efficiency
is lowered. The gate charge current is dependent on
operatingfrequencyasdiscussedintheEfficiencyConsid-
erations section. The junction temperature can be esti-
mated by using the equations given in Note 2 of the
ElectricalCharacteristics.Forexample,theLTC1628-SYNC
VIN current is limited to less than 24mA from a 24V supply
when not using the EXTVCC pin as follows:
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by a
factor of (Duty Cycle)/(Efficiency). For 5V regulators this
supply means connecting the EXTVCC pin directly to VOUT
.
However, for 3.3V and other lower voltage regulators,
additional circuitry is required to derive INTVCC power
from the output.
The following list summarizes the four possible connec-
tions for EXTVCC:
1. EXTVCCLeftOpen(orGrounded).ThiswillcauseINTVCC
to be powered from the internal 5V regulator resulting in
an efficiency penalty of up to 10% at high input voltages.
2. EXTVCC Connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
TJ = 70°C + (24mA)(24V)(95°C/W) = 125°C
UseoftheEXTVCC inputpinreducesthejunctiontempera-
ture to:
3. EXTVCC Connected to an External supply. If an external
supply is available in the 5V to 7V range, it may be used to
powerEXTVCC providingitiscompatiblewiththeMOSFET
gate drive requirements.
TJ = 70°C + (24mA)(5V)(95°C/W) = 81°C
Dissipationshouldbecalculatedtoalsoincludeanyadded
current drawn from the internal 3.3V linear regulator. To
prevent maximum junction temperature from being ex-
ceeded, the input supply current must be checked operat-
ing in continuous mode at maximum VIN.
4. EXTVCC Connected to an Output-Derived Boost Net-
work. For3.3Vandotherlowvoltageregulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage that has been boosted to greater
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LTC1628-SYNC
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than 4.7V. This can be done with either the inductive boost
winding as shown in Figure 6a or the capacitive charge
pump shown in Figure 6b. The charge pump has the
advantage of simple magnetics.
compared with the internal precision 0.800V voltage ref-
erence by the error amplifier. The output voltage is given
by the equation:
R2
R1
V
= 0.8V 1+
OUT
Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors CB connected to the BOOST
pinssupplythegatedrivevoltagesforthetopsideMOSFETs.
Capacitor CB in the functional diagram is charged though
external diode DB from INTVCC when the SW pin is low.
When one of the topside MOSFETs is to be turned on, the
driver places the CB voltage across the gate-source of the
desiredMOSFET.ThisenhancestheMOSFETandturnson
the topside switch. The switch node voltage, SW, rises to
VIN and the BOOST pin follows. With the topside MOSFET
where R1 and R2 are defined in Figure 2.
SENSE+/SENSE– Pins
The common mode input range of the current comparator
sense pins is from 0V to (1.1)INTVCC. Continuous linear
operation is guaranteed throughout this range allowing
output voltage setting from 0.8V to 7.7V, depending upon
the voltage applied to EXTVCC. A differential NPN input
stage is biased with internal resistors from an internal
2.4V source as shown in the Functional Diagram. This
requires that current either be sourced or sunk from the
SENSE pins depending on the output voltage. If the output
voltage is below 2.4V current will flow out of both SENSE
pinstothemainoutput.Theoutputcanbeeasilypreloaded
by the VOUT resistive divider to compensate for the current
comparator’s negative input bias current. The maximum
current flowing out of each pair of SENSE pins is:
on, the boost voltage is above the input supply: VBOOST
=
VIN + VINTVCC. The value of the boost capacitor CB needs
to be 100 times that of the total input capacitance of the
topside MOSFET(s). The reverse breakdown of the exter-
nal Schottky diode must be greater than VIN(MAX). When
adjusting the gate drive level, the final arbiter is the total
input current for the regulator. If a change is made and the
input current decreases, then the efficiency has improved.
If there is no change in input current, then there is no
change in efficiency.
ISENSE+ + ISENSE– = (2.4V – VOUT)/24k
Output Voltage
SinceVOSENSE isservoedtothe0.8Vreferencevoltage, we
can choose R1 in Figure 2 to have a maximum value to
absorb this current.
The LTC1628-SYNC output voltages are each set by an
external feedback resistive divider carefully placed across
the output capacitor. The resultant feedback signal is
+
V
IN
V
IN
1µF
OPTIONAL EXTV
CONNECTION
CC
+
+
5V < V
< 7V
SEC
C
IN
C
IN
0.22µF
BAT85
BAT85
BAT85
V
V
V
IN
IN
SEC
+
N-CH
LTC1628-SYNC
N-CH
LTC1628-SYNC
1µF
VN2222LL
TG1
TG1
R
SENSE
R
SENSE
V
V
OUT
OUT
L1
T1
1:N
EXTV
SW
EXTV
FCB
CC
SW
CC
R6
R5
+
+
C
OUT
C
OUT
BG1
BG1
N-CH
N-CH
PGND
SGND
PGND
1628 F06b
1628 F06a
Figure 6b. Capacitive Charge Pump for EXTVCC
Figure 6a. Secondary Output Loop & EXTVCC Connection
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U
V
INTV
IN
CC
R
0.8V
2.4V – V
3.3V OR 5V
RUN/SS
*
R1
= 24k
R
*
SS
(MAX)
SS
D1
OUT
RUN/SS
C
SS
for VOUT < 2.4V
C
SS
Regulating an output voltage of 1.8V, the maximum value
of R1 should be 32K. Note that for an output voltage above
2.4V, R1 has no maximum value necessary to absorb the
sense currents; however, R1 is still bounded by the
*OPTIONAL TO DEFEAT OVERCURRENT LATCHOFF
(a)
1628 F07
(b)
Figure 7. RUN/SS Pin Interfacing
V
OSENSE feedback current.
Soft-Start/Run Function
Fault Conditions: Overcurrent Latchoff
The RUN/SS1 and RUN/SS2 pins are multipurpose pins
that provide a soft-start function and a means to shut
down the LTC1628-SYNC. Soft-start reduces the input
powersource’ssurgecurrentsbygraduallyincreasingthe
controller’s current limit (proportional to VITH). This pin
can also be used for power supply sequencing.
The RUN/SS pins also provide the ability to latch off the
controller(s) when an overcurrent condition is detected.
The RUN/SS capacitor, CSS, is used initially to turn on and
limit the inrush current. After the controller has been
started and been given adequate time to charge up the
outputcapacitorandprovidefullloadcurrent, theRUN/SS
capacitorisusedforashort-circuittimer. Iftheregulator’s
output voltage falls to less than 70% of its nominal value
after CSS reaches 4.1V, CSS begins discharging on the
assumption that the output is in an overcurrent condition.
If the condition lasts for a long enough period as deter-
mined by the size of the CSS and the specified discharge
current, the controller will be shut down until the RUN/SS
pin voltage is recycled. If the overload occurs during start-
up, the time can be approximated by:
An internal 1.2µA current source charges up the CSS
capacitor. When the voltage on RUN/SS1 (RUN/SS2)
reaches 1.5V, the particular controller is permitted to start
operating. As the voltage on RUN/SS increases from 1.5V
to 3.0V, the internal current limit is increased from 25mV/
RSENSE to 75mV/RSENSE. The output current limit ramps
up slowly, taking an additional 1.25s/µF to reach full
current. The output current thus ramps up slowly, reduc-
ing the starting surge current required from the input
power supply. If RUN/SS has been pulled all the way to
ground there is a delay before starting of approximately:
tLO1 ≈ [CSS(4.1 – 1.5 + 4.1 – 3.5)]/(1.2µA)
= 2.7 • 106 (CSS)
If the overload occurs after start-up the voltage on CSS will
begin discharging from the zener clamp voltage:
1.5V
1.2µA
tDELAY
=
=
C
SS = 1.25s / µF CSS
(
)
t
LO2 ≈ [CSS (6 – 3.5)]/(1.2µA) = 2.1 • 106 (CSS)
3V − 1.5V
1.2µA
This built-in overcurrent latchoff can be overridden by
providing a pull-up resistor to the RUN/SS pin as shown
in Figure 7. This resistance shortens the soft-start period
and prevents the discharge of the RUN/SS capacitor
during an over current condition. Tying this pull-up resis-
tor to VIN as in Figure 7a, defeats overcurrent latchoff.
Diode-connecting this pull-up resistor to INTVCC, as in
Figure 7b, eliminates any extra supply current during
controller shutdown while eliminating the INTVCC loading
from preventing controller start-up.
tIRAMP
C
SS = 1.25s / µF CSS
(
)
By pulling both RUN/SS pins below 1V, the LTC1628-
SYNC is put into low current shutdown (IQ = 20µA). The
RUN/SS pins can be driven directly from logic as shown
in Figure 7. Diode D1 in Figure 7 reduces the start delay
but allows CSS to ramp up slowly providing the soft-start
function. EachRUN/SSpinhasaninternal6Vzenerclamp
(See Functional Diagram).
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Why should you defeat overcurrent latchoff? During the
prototyping stage of a design, there may be a problem
with noise pickup or poor layout causing the protection
circuit to latch off. Defeating this feature will easily allow
troubleshooting of the circuit and PC layout. The internal
short-circuit and foldback current limiting still remains
active, thereby protecting the power supply system from
failure. After the design is complete, a decision can be
made whether to enable the latchoff feature.
The resulting short-circuit current is:
25mV
RSENSE
1
2
ISC
=
+ ∆IL(SC)
Fault Conditions: Overvoltage Protection (Crowbar)
The overvoltage crowbar is designed to blow a system
input fuse when the output voltage of the regulator rises
much higher than nominal levels. The crowbar causes
huge currents to flow, that blow the fuse to protect against
a shorted top MOSFET if the short occurs while the
controller is operating.
The value of the soft-start capacitor CSS may need to be
scaled with output voltage, output capacitance and load
current characteristics. The minimum soft-start capaci-
tance is given by:
A comparator monitors the output for overvoltage condi-
tions. The comparator (OV) detects overvoltage faults
greater than 7.5% above the nominal output voltage.
When this condition is sensed, the top MOSFET is turned
off and the bottom MOSFET is turned on until the overvolt-
age condition is cleared. The output of this comparator is
only latched by the overvoltage condition itself and will
thereforeallowaswitchingregulatorsystemhavingapoor
PC layout to function while the design is being debugged.
The bottom MOSFET remains on continuously for as long
as the OV condition persists; if VOUT returns to a safe level,
normal operation automatically resumes. A shorted top
MOSFET will result in a high current condition which will
open the system fuse. The switching regulator will regu-
late properly with a leaky top MOSFET by altering the duty
cycle to accommodate the leakage.
C
SS > (COUT )(VOUT) (10–4) (RSENSE
)
The minimum recommended soft-start capacitor of
CSS = 0.1µF will be sufficient for most applications.
Fault Conditions: Current Limit and Current Foldback
The LTC1628-SYNC current comparator has a maximum
sense voltage of 75mV resulting in a maximum MOSFET
current of 75mV/RSENSE. The maximum value of current
limit generally occurs with the largest VIN at the highest
ambient temperature, conditions that cause the highest
power dissipation in the top MOSFET.
The LTC1628-SYNC includes current foldback to help
further limit load current when the output is shorted to
ground. The foldback circuit is active even when the
overload shutdown latch described above is overridden. If
theoutputfallsbelow70%ofitsnominaloutputlevel,then
themaximumsensevoltageisprogressivelyloweredfrom
75mV to 25mV. Under short-circuit conditions with very
low duty cycles, the LTC1628-SYNC will begin cycle
skipping in order to limit the short-circuit current. In this
situation the bottom MOSFET will be dissipating most of
the power but less than in normal operation. The short-
circuit ripple current is determined by the minimum on-
time tON(MIN) of the LTC1628-SYNC (less than 200ns), the
input voltage and inductor value:
Phase-Locked Loop and Frequency Synchronization
The LTC1628-SYNC has a phase-locked loop comprised
of an internal voltage controlled oscillator and phase
detector. This allows the top MOSFET turn-on to be locked
to the rising edge of an external source. The frequency
range of the voltage controlled oscillator is ±50% around
the center frequency fO. A voltage applied to the PLLFLTR
pin of 1.2V corresponds to a frequency of approximately
220kHz. The nominal operating frequency range of the
LTC1628-SYNC is 140kHz to 310kHz.
∆IL(SC) = tON(MIN) (VIN/L)
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The phase detector used is an edge sensitive digital type
which provides zero degrees phase shift between the
external and internal oscillators. This type of phase detec-
tor will not lock up on input frequencies close to the
harmonics of the VCO center frequency. The PLL hold-in
range, ∆fH, is equal to the capture range, ∆fC:
U
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest time duration
that the LTC1628-SYNC is capable of turning on the top
MOSFET. Itisdeterminedbyinternaltimingdelaysandthe
gate charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that
∆fH = ∆fC = ±0.5 fO (150kHz-300kHz)
The output of the phase detector is a complementary pair
of current sources charging or discharging the external
filter network on the PLLFLTR pin. A simplified block
diagram is shown in Figure 7.
VOUT
V (f)
IN
tON(MIN)
<
Ifthedutycyclefallsbelowwhatcanbeaccommodatedby
the minimum on-time, the LTC1628-SYNC will begin to
skip cycles. The output voltage will continue to be regu-
lated, but the ripple voltage and current will increase.
If the external frequency (fPLLIN) is greater than the oscil-
lator frequency f0SC, current is sourced continuously,
pulling up the PLLFLTR pin. When the external frequency
is less than f0SC, current is sunk continuously, pulling
down the PLLFLTR pin. If the external and internal fre-
quencies are the same but exhibit a phase difference, the
currentsourcesturnonforanamountoftimecorrespond-
ing to the phase difference. Thus the voltage on the
PLLFLTR pin is adjusted until the phase and frequency of
the external and internal oscillators are identical. At this
stable operating point the phase comparator output is
open and the filter capacitor CLP holds the voltage. The
LTC1628-SYNC PLLIN pin must be driven from a low
impedance source such as a logic gate located close to the
pin.WhenusingmultipleLTC1628-SYNC’s(orLTC1629’s,
as shown in Figure 14) for a phase-locked system, the
PLLFLTR pin of the master oscillator should be biased at
a voltage that will guarantee the slave oscillator(s) ability
to lock onto the master’s frequency. A DC voltage of 0.7V
to 1.7V applied to the master oscillator’s PLLFLTR pin is
recommended in order to meet this requirement. The
resultant operating frequency can range from 170kHz to
270kHz.
The minimum on-time for the LTC1628-SYNC is generally
less than 200ns. However, as the peak sense voltage
decreasestheminimumon-timegraduallyincreasesupto
about 300ns. This is of particular concern in forced
continuous applications with low ripple current at light
loads. If the duty cycle drops below the minimum on-time
limit in this situation, a significant amount of cycle skip-
ping can occur with correspondingly larger current and
voltage ripple.
FCB Pin Operation
The FCB pin can be used to regulate a secondary winding
or as a logic level input. Continuous operation is forced on
both controllers when the FCB pin drops below 0.8V.
During continuous mode, current flows continuously in
the transformer primary. The secondary winding(s) draw
current only when the bottom, synchronous switch is on.
When primary load currents are low and/or the VIN/VOUT
ratio is low, the synchronous switch may not be on for a
sufficient amount of time to transfer power from the
outputcapacitortothesecondaryload.Forcedcontinuous
operationwillsupportsecondarywindingsprovidingthere
is sufficient synchronous switch duty factor. Thus, the
FCB input pin removes the requirement that power must
be drawn from the inductor primary in order to extract
The loop filter components (CLP, RLP) smooth out the
current pulses from the phase detector and provide a
stable input to the voltage controlled oscillator. The filter
components CLP and RLP determine how fast the loop
acquires lock. Typically RLP =10kΩ and CLP is 0.01µF to
0.1µF.
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power from the auxiliary windings. With the loop in
continuous mode, the auxiliary outputs may nominally be
loaded without regard to the primary output load.
loop is reduced depending upon the maximum load step
specifications. Voltage positioning can easily be added to
the LTC1628-SYNC by loading the ITH pin with a resistive
divider having a Thevenin equivalent voltage source equal
to the midpoint operating voltage range of the error
amplifier, or 1.2V (see Figure 8).
The secondary output voltage VSEC is normally set as
shown in Figure 6a by the turns ratio N of the transformer:
VSEC (N + 1) VOUT
The resistive load reduces the DC loop gain while main-
taining the linear control range of the error amplifier. The
maximum output voltage deviation can theoretically be
reduced to half or alternatively the amount of output
capacitance can be reduced for a particular application. A
complete explanation is included in Design Solutions 10.
(See www.linear-tech.com)
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current,
then VSEC will droop. An external resistive divider from
VSEC to the FCB pin sets a minimum voltage VSEC(MIN)
:
R6
R5
V
≈ 0.8V 1+
SEC(MIN)
INTV
CC
where R5 and R6 are shown in Figure 2.
R
T2
T1
I
TH
If VSEC drops below this level, the FCB voltage forces
temporary continuous switching operation until VSEC is
again above its minimum.
LTC1628-SYNC
R
R
C
C
C
1628 F08
In order to prevent erratic operation if no external connec-
tions are made to the FCB pin, the FCB pin has a 0.18µA
internal current source pulling the pin high. Include this
current when choosing resistor values R5 and R6.
Figure 8. Active Voltage Positioning Applied
to the LTC1628-SYNC
Efficiency Considerations
The following table summarizes the possible states avail-
able on the FCB pin:
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
Table 1
FCB Pin
Condition
0V to 0.75V
Forced Continuous Both Controllers
(Current Reversal Allowed—
Burst Inhibited)
0.85V < V < 4.3V
Minimum Peak Current Induces
Burst Mode Operation
FCB
%Efficiency = 100% – (L1 + L2 + L3 + ...)
No Current Reversal Allowed
whereL1, L2, etc. aretheindividuallossesasapercentage
of input power.
Feedback Resistors
>4.8V
Regulating a Secondary Winding
Burst Mode Operation Disabled
Constant Frequency Mode Enabled
No Current Reversal Allowed
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1628-SYNC circuits: 1) LTC1628-SYNC VIN
current (including loading on the 3.3V internal regulator),
2) INTVCC regulator current, 3) I2R losses, 4) Topside
MOSFET transition losses.
No Minimum Peak Current
Voltage Positioning
Voltage positioning can be used to minimize peak-to-peak
output voltage excursions under worst-case transient
loading conditions. The open-loop DC gain of the control
1. The VIN current has two components: the first is the DC
supply current given in the Electrical Characteristics table,
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U
which excludes MOSFET driver and control currents; the
second is the current drawn from the 3.3V linear regulator
output.VINcurrenttypicallyresultsinasmall(<0.1%)loss.
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high input
voltages (typically 15V or greater). Transition losses can
be estimated from:
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high to
low again, a packet of charge dQ moves from INTVCC to
ground. The resulting dQ/dt is a current out of INTVCC that
is typically much larger than the control circuit current. In
continuous mode, IGATECHG =f(QT+QB), where QT and QB
are the gate charges of the topside and bottom side
MOSFETs.
2
Transition Loss = (1.7) VIN IO(MAX) CRSS
f
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is very
important to include these “system” level losses during
the design phase. The internal battery and fuse resistance
losses can be minimized by making sure that CIN has
adequate charge storage and very low ESR at the switch-
ing frequency. A 25W supply will typically require a mini-
mum of 20µF to 40µF of capacitance having a maximum
of 20mΩ to 50mΩ of ESR. The LTC1628-SYNC 2-phase
architecture typically halves this input capacitance re-
quirement over competing solutions. Other losses includ-
ing Schottky conduction losses during dead-time and
inductor core losses generally account for less than 2%
total additional loss.
SupplyingINTVCC powerthroughtheEXTVCC switchinput
from an output-derived source will scale the VIN current
required for the driver and control circuits by a factor of
(Duty Cycle)/(Efficiency). For example, in a 20V to 5V
application, 10mA of INTVCC current results in approxi-
mately2.5mAofVIN current. Thisreducesthemid-current
loss from 10% or more (if the driver was powered directly
from VIN) to only a few percent.
Checking Transient Response
3. I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor,
and input and output capacitor ESR. In continuous mode
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to ∆ILOAD (ESR), where ESR is the effective
series resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. OPTI-
LOOP compensation allows the transient response to be
optimized over a wide range of output capacitance and
ESR values. The availability of the ITH pin not only allows
optimization of control loop behavior but also provides a
DC coupled and AC filtered closed loop response test
point. The DC step, rise time and settling at this test point
truly reflects the closed loop response. Assuming a pre-
dominantly second order system, phase margin and/or
damping factor can be estimated using the percentage of
the average output current flows through L and RSENSE
,
but is “chopped” between the topside MOSFET and the
synchronous MOSFET. If the two MOSFETs have approxi-
mately the same RDS(ON), then the resistance of one
MOSFET can simply be summed with the resistances of L,
RSENSE and ESR to obtain I2R losses. For example, if each
RDS(ON) = 30mΩ, RL = 50mΩ, RSENSE = 10mΩ and RESR
= 40mΩ (sum of both input and output capacitance
losses), then the total resistance is 130mΩ. This results in
losses ranging from 3% to 13% as the output current
increases from 1A to 5A for a 5V output, or a 4% to 20%
loss for a 3.3V output. Efficiency varies as the inverse
square of VOUT for the same external components and
output power level. The combined effects of increasingly
lower output voltages and higher currents required by
high performance digital systems is not doubling but
quadrupling the importance of loss terms in the switching
regulator system!
23
LTC1628-SYNC
W U U
U
APPLICATIO S I FOR ATIO
overshoot seen at this pin. The bandwidth can also be
estimated by examining the rise time at the pin. The ITH
external components shown in the Figure 1 circuit will
provide an adequate starting point for most applications.
approximately 25 • CLOAD. Thus a 10µF capacitor would
require a 250µs rise time, limiting the charging current to
about 200mA.
Automotive Considerations: Plugging into the
Cigarette Lighter
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80% of
full-load current having a rise time of 1µs to 10µs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop. Placing a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This is
why it is better to look at the ITH pin signal which is in the
feedback loop and is the filtered and compensated control
loop response. The gain of the loop will be increased by
increasing RC and the bandwidth of the loop will be
increased by decreasing CC. If RC is increased by the same
factor that CC is decreased, the zero frequency will be kept
the same, thereby keeping the phase shift the same in the
most critical frequency range of the feedback loop. The
outputvoltagesettlingbehaviorisrelatedtothestabilityof
the closed-loop system and will demonstrate the actual
overall supply performance.
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserveorevenrechargebatterypacksduringoperation.
But before you connect, be advised: you are plugging into
thesupplyfromhell. Themainpowerlineinanautomobile
is the source of a number of nasty potential transients,
including load-dump, reverse-battery, and double-bat-
tery.
Load-dump is the result of a loose battery cable. When the
cablebreaksconnection,thefieldcollapseinthealternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
just what it says, while double-battery is a consequence of
tow-truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
ThenetworkshowninFigure9isthemoststraightforward
approach to protect a DC/DC converter from the ravages
of an automotive power line. The series diode prevents
current from flowing during reverse-battery, while the
transient suppressor clamps the input voltage during
load-dump. Note that the transient suppressor should not
conduct during double-battery operation, but must still
clamptheinputvoltagebelowbreakdownoftheconverter.
Although the LTC1628-SYNC has a maximum input volt-
age of 36V, most applications will be limited to 30V by the
MOSFET BVDSS.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
dischargedbypasscapacitorsareeffectivelyputinparallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than1:50, the switch rise time
should be controlled so that the load rise time is limited to
50A I RATING
PK
V
IN
12V
LTC1628-SYNC
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
1628 F09
Figure 9. Automotive Application Protection
24
LTC1628-SYNC
W U U
APPLICATIO S I FOR ATIO
U
Design Example
Choosing 1% resistors; R1 = 25.5k and R2 = 32.4k yields
an output voltage of 1.816V.
As a design example for one channel, assume VIN
=
12V(nominal), VIN = 22V(max), VOUT = 1.8V, IMAX = 5A,
and f = 300kHz.
The power dissipation on the top side MOSFET can be
easily estimated. Choosing a Siliconix Si4412DY results
in; RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input
voltage with T(estimated) = 50°C:
Theinductancevalueischosenfirstbasedona30%ripple
current assumption. The highest value of ripple current
occursatthemaximuminputvoltage. TiethePLLFLTRpin
to the INTVCC pin for 300kHz operation. The minimum
inductance for 30% ripple current is:
2
( )
1.8V
22V
P
=
(
5 1+(0.005)(50°C – 25°C)
MAIN
[
]
2
0.042Ω + 1.7 22V 5A 100pF 300kHz
)
(
) ( )(
)(
)
V
(f)(L)
V
OUT
V
IN
OUT
∆I =
1–
L
= 220mW
Ashort-circuittogroundwillresultinafoldedbackcurrent
of:
A 4.7µH inductor will produce 23% ripple current and a
3.3µH will result in 33%. The peak inductor current will be
the maximum DC value plus one half the ripple current, or
5.84A, for the 3.3µH value. Increasing the ripple current
will also help ensure that the minimum on-time of 200ns
is not violated. The minimum on-time occurs at maximum
VIN:
25mV 1 200ns(22V)
ISC
=
+
= 3.2A
0.01Ω
2
3.3µH
with a typical value of RDS(ON) and δ = (0.005/°C)(20) =
0.1. TheresultingpowerdissipatedinthebottomMOSFET
is:
VOUT
1.8V
tON(MIN)
=
=
= 273ns
2
22V – 1.8V
22V
= 434mW
V
IN(MAX)f 22V(300kHz)
PSYNC
=
3.2A 1.1 0.042Ω
(
) ( )(
)
The RSENSE resistor value can be calculated by using the
maximum current sense voltage specification with some
accommodation for tolerances:
which is less than under full-load conditions.
CIN is chosen for an RMS current rating of at least 3A at
temperature assuming only this channel is on. COUT is
chosen with an ESR of 0.02Ω for low output ripple. The
output ripple in continuous mode will be highest at the
maximum input voltage. The output voltage ripple due to
ESR is approximately:
60mV
5.84A
RSENSE
≤
≈ 0.01Ω
Since the output voltage is below 2.4V the output resistive
divider will need to be sized to not only set the output
voltage but also to absorb the SENSE pins specified input
current.
VORIPPLE = RESR (∆IL) = 0.02Ω(1.67A) = 33mVP–P
0.8V
R1
= 24k
= 24K
(MAX)
2.4V – V
OUT
0.8V
2.4V – 1.8V
= 32k
25
LTC1628-SYNC
W U U
U
APPLICATIO S I FOR ATIO
PC Board Layout Checklist
2. Are the signal and power grounds kept separate? The
combined LTC1628-SYNC signal ground pin and the
ground return of CINTVCC must return to the combined
COUT (–) terminals. The path formed by the top N-channel
MOSFET, Schottky diode and the CIN capacitor should
have short leads and PC trace lengths. The output capaci-
tor (–) terminals should be connected as close as possible
to the (–) terminals of the input capacitor by placing the
capacitors next to each other and away from the Schottky
loop described above.
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1628-SYNC. These items are also illustrated graphi-
cally in the layout diagram of Figure 10. The Figure 11
illustrates the current waveforms present in the various
branches of the 2-phase synchronous regulators operat-
ing in the continuous mode. Check the following in your
layout:
1. Are the top N-channel MOSFETs M1 and M3 located
within 1cm of each other with a common drain connection
at CIN? Do not attempt to split the input decoupling for the
two channels as it can cause a large resonant loop.
3. Do the LTC1628-SYNC VOSENSE pins resistive dividers
connect to the (+) terminals of COUT? The resistive divider
must be connected between the (+) terminal of COUT and
R
PU
V
PULL-UP
(<7V)
1
2
28
27
26
25
24
23
22
21
20
19
18
17
16
15
PGOOD
RUN/SS1
PGOOD
TG1
L1
R
SENSE
+
V
SENSE1
OUT1
3
–
SENSE1
SW1
R2
M1
M2
C
B1
R1
4
D1
V
BOOST1
OSENSE1
5
PLLFLTR
PLLIN
FCB
V
IN
f
IN
6
C
C
OUT1
BG1
R
IN
7
C
IN
INTV
EXTV
CC
CC
CC
C
VIN
GND
LTC1628-SYNC
INTV
8
I
TH1
V
IN
C
INTVCC
9
SGND
PGND
BG2
OUT2
D2
10
11
12
13
14
3.3V
3.3V
OUT
I
BOOST2
SW2
TH2
C
B2
M3
M4
L2
V
OSENSE2
R
R3
R4
SENSE
–
V
SENSE2
SENSE2
TG2
OUT2
+
RUN/SS2
1628 F10
Figure 10. LTC1628-SYNC Recommended Printed Circuit Layout Diagram
26
LTC1628-SYNC
W U U
APPLICATIO S I FOR ATIO
U
SW1
L1
R
SENSE1
V
OUT1
+
D1
C
OUT1
R
L1
V
IN
R
IN
+
C
IN
SW2
L2
R
SENSE2
V
OUT2
+
D2
C
OUT2
R
L2
BOLD LINES INDICATE
HIGH, SWITCHING
CURRENT LINES.
KEEP LINES TO A
MINIMUM LENGTH.
1628 F11
Figure 11. Branch Current Waveforms
signal ground. The R2 and R4 connections should not be
along the high current input feeds from the input
capacitor(s).
An additional 1µF ceramic capacitor placed immediately
next to the INTVCC and PGND pins can help improve noise
performance substantially.
4. Are the SENSE – and SENSE+ leads routed together
with minimum PC trace spacing? The filter capacitor
between SENSE+ and SENSE– should be as close as
possible to the IC. Ensure accurate current sensing with
Kelvin connections at the SENSE resistor.
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), and boost nodes (BOOST1, BOOST2) away
from sensitive small-signal nodes, especially from the
oppositeschannel’svoltageandcurrentsensingfeedback
pins. All of these nodes have very large and fast moving
signals and therefore should be kept on the “output side”
of the LTC1628-SYNC and occupy minimum PC trace
area.
5. Is the INTVCC decoupling capacitor connected close to
the IC, between the INTVCC and the power ground pins?
This capacitor carries the MOSFET drivers current peaks.
27
LTC1628-SYNC
U
W
U U
APPLICATIO S I FOR ATIO
7. Use a modified “star ground” technique: a low imped-
ance, large copper area central grounding point on the
same side of the PC board as the input and output
capacitors with tie-ins for the bottom of the INTVCC
decoupling capacitor, the bottom of the voltage feedback
resistive divider and the SGND pin of the IC.
Short-circuit testing can be performed to verify proper
overcurrent latchoff, or 5µA can be provided to the RUN/
SS pin(s) by resistors from VIN to prevent the short-circuit
latchoff from occurring.
ReduceVIN fromitsnominalleveltoverifyoperationofthe
regulator in dropout. Check the operation of the
undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
PC Board Layout Debugging
Start with one controller on at a time. It is helpful to use a
DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output
switching node (SW pin) to synchronize the oscilloscope
to the internal oscillator and probe the actual output
voltage as well. Check for proper performance over the
operating voltage and current range expected in the appli-
cation. The frequency of operation should be maintained
over the input voltage range down to dropout and until the
output load drops below the low current operation thresh-
old—typically 10% to 20% of the maximum designed
current level in Burst Mode operation.
Investigate whether any problems exist only at higher
output currents or only at higher input voltages. If prob-
lems coincide with high input voltages and low output
currents,lookforcapacitivecouplingbetweentheBOOST,
SW, TG, and possibly BG connections and the sensitive
voltage and current pins. The capacitor placed across the
current sensing pins needs to be placed immediately
adjacent to the pins of the IC. This capacitor helps to
minimize the effects of differential noise injection due to
high frequency capacitive coupling. If problems are en-
countered with high current output loading at lower input
voltages,lookforinductivecouplingbetweenCIN,Schottky
and the top MOSFET components to the sensitive current
and voltage sensing traces. In addition, investigate com-
mon ground path voltage pickup between these compo-
nents and the SGND pin of the IC.
The duty cycle percentage should be maintained from
cycle to cycle in a well-designed, low noise PCB imple-
mentation. Variation in the duty cycle at a subharmonic
rate can suggest noise pickup at the current or voltage
sensing inputs or inadequate loop compensation. Over-
compensation of the loop can be used to tame a poor PC
layout if regulator bandwidth optimization is not required.
Only after each controller is checked for their individual
performance should both controllers be turned on at the
same time. A particularly difficult region of operation is
when one controller channel is nearing its current com-
parator trip point when the other channel is turning on its
topMOSFET. Thisoccursaround50%dutycycleoneither
channel due to the phasing of the internal clocks and may
cause minor duty cycle jitter.
An embarrassing problem, which can be missed in an
otherwise properly working switching regulator, results
when the current sensing leads are hooked up backwards.
Theoutputvoltageunderthisimproperhookupwillstillbe
maintained but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
28
LTC1628-SYNC
U
TYPICAL APPLICATIO S
59k
1M
100k
MBRS1100T3
V
+
PULL-UP
33µF
25V
(<7V)
T1, 1:1.8
10µH
1
2
28
27
26
25
24
23
22
21
20
19
18
17
16
15
PGOOD
TG1
PGOOD
RUN/SS1
0.015Ω
V
0.1µF
OUT1
5V
+
SENSE1
3A; 4A PEAK
180pF
1000pF
INTV
3
8
–
SW1
SENSE1
105k, 1%
5
M1
M2
0.1µF
D1
MBRM
140T3
LT1121
ON/OFF
4
BOOST1
V
OSENSE1
3
2
1
20k
1%
220k
V
5
OUT2
V
IN
PLLFLTR
PLLIN
FCB
CC
12V
120mA
6
150µF, 6.3V
PANASONIC SP
BG1
+
33pF
1µF
25V
10Ω
22µF
50V
100k
7
CMDSH-3TR
EXTV
CC
0.1µF
GND
LTC1628-SYNC
INTV
8
I
CC
TH1
1µF
10V
15k
4.7µF
1000pF
1000pF
9
180µF, 4V
PANASONIC SP
PGND
BG2
SGND
V
33pF
IN
7V TO
28V
CMDSH-3TR
10
11
12
13
14
3.3V
3.3V
OUT
D2
MBRM
140T3
BOOST2
SW2
I
TH2
15k
0.1µF
M3
M4
V
OSENSE2
20k
1%
V
OUT2
3.3V
5A; 6A PEAK
–
TG2
SENSE2
SENSE2
63.4k
1%
0.01Ω
1000pF
L1
6.3µH
+
RUN/SS2
180pF
0.1µF
1628 F12
V
V
: 7V TO 28V
IN
: 5V, 3A/3.3V, 6A, 12V, 150mA
OUT
SWITCHING FREQUENCY = 300kHz
MI, M2, M3, M4: NDS8410A
L1: SUMIDA CEP123-6R3MC
T1: 10µH 1:1.8 — DALE LPE6562-A262 GAPPED E-CORE OR BH ELECTRONICS #501-0657 GAPPED TOROID
Figure 12. LTC1628-SYNC High Efficiency Low Noise 5V/3A, 3.3V/5A, 12V/120mA Regulator
29
LTC1628-SYNC
U
TYPICAL APPLICATIO S
V
PULL-UP
(<7V)
1
28
27
26
25
24
23
22
21
20
19
18
17
16
15
L1
PGOOD
RUN/SS1
PGOOD
TG1
8µH
0.015Ω
V
0.1µF
OUT1
2
3
+
–
SENSE1
SENSE1
5V
3A; 4A PEAK
27pF
1000pF
105k
1%
SW1
20k
1%
M1A
M1B
0.1µF
D1
MBRM
140T3
4
V
BOOST1
OSENSE1
5
PLLFLTR
PLLIN
FCB
V
IN
0.01µF
10k
1000pF
47µF
6.3V
6
f
SYNC
BG1
33pF
10Ω
22µF
50V
7
CMDSH-3TR
EXTV
INTV
CC
CC
0.1µF
GND
LTC1628-SYNC
8
I
TH1
1µF
10V
15k
4.7µF
220pF
9
SGND
PGND
BG2
56µF, 4V
V
33pF
IN
CMDSH-3TR
5.2V TO
28V
10
11
12
13
14
3.3V
3.3V
OUT
D2
MBRM
140T3
I
BOOST2
SW2
TH2
15k
0.1µF
M2A
M2B
220pF
V
OSENSE2
20k
1%
V
OUT2
–
3.3V
SENSE2
SENSE2
TG2
0.015Ω
63.4k
1%
3A; 4A PEAK
1000pF
L2
8µH
+
27pF
RUN/SS2
0.1µF
: 5.2V TO 28V
1628 F13
V
IN
V
SWITCHING FREQUENCY = 300kHz
MI, M2: FDS8936A
L1, L2: 8µH SUMIDA CEP1238R0MC
OUTPUT CAPACITORS: PANASONIC SP SERIES
: 5V, 4A/3.3V, 4A
OUT
Figure 13. LTC1628-SYNC 5V/4A, 3.3V/4A Regulator with External Frequency Synchronization
30
LTC1628-SYNC
U
PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
G Package
28-Lead Plastic SSOP (0.209)
(LTC DWG # 05-08-1640)
10.07 – 10.33*
(0.397 – 0.407)
28 27 26 25 24 23 22 21 20 19 18
16 15
17
7.65 – 7.90
(0.301 – 0.311)
5
7
8
1
2
3
4
6
9 10 11 12 13 14
5.20 – 5.38**
(0.205 – 0.212)
1.73 – 1.99
(0.068 – 0.078)
0° – 8°
0.65
(0.0256)
BSC
0.13 – 0.22
0.55 – 0.95
(0.005 – 0.009)
(0.022 – 0.037)
0.05 – 0.21
(0.002 – 0.008)
0.25 – 0.38
(0.010 – 0.015)
NOTE: DIMENSIONS ARE IN MILLIMETERS
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.152mm (0.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.254mm (0.010") PER SIDE
G28 SSOP 1098
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
31
LTC1628-SYNC
U
TYPICAL APPLICATIO
I
IN
12V
IN
C
IN
I
I
*
IN
1
0°
BUCK: 2.5V/15A
BUCK: 2.5V/15A
OPEN
PHASMD TG1
180°
I
1
2
3
4
2.5V /30A
O
TG2
U1
LTC1629
I
I
90°
2
3
I
CLKOUT
I
I
1.5V /15A
O
90°
BUCK: 1.5V/15A
BUCK: 1.8V/15A
TG1
270°
1.8V /15A
O
TG2
LTC1628-SYNC
U2
*INPUT RIPPLE CURRENT CANCELLATION
INCREASES THE RIPPLE FREQUENCY AND
REDUCES THE RMS INPUT RIPPLE CURRENT
THUS, SAVING INPUT CAPACITORS
I
90°
4
PLLIN
1628 F14
Figure 14. Multioutput Multiphase Application
RELATED PARTS
PART NUMBER
LTC1438/LTC1439
LTC1438-ADJ
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Mobile Pentium® III Processors, 550kHz,
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≤ 3.5V, Current Mode Ensures
OUT
Accurate Current Sharing, 3.5V ≤ V ≤ 36V
IN
LTC1735
LTC1736
High Efficiency Synchronous Step-Down Switching Regulator
Output Fault Protection, 16-Pin SSOP
High Efficiency Synchronous Controller with 5-Bit Mobile VID Control
Output Fault Protection, 24-Pin SSOP,
3.5V ≤ V ≤ 36V
IN
LTC1929/
LTC1929-PG
2-Phase Synchronous Controllers
Up to 42A, Uses All Surface Mount Components,
No Heat Sinks, 3.5V ≤ V ≤ 36V
IN
Adaptive Power, No R
and PolyPhase are trademarks of Linear Technology Corporation. Pentium is a registered trademark of Intel Corporation.
SENSE
1628sf LT/TP 0700 4K • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
32
LINEAR TECHNOLOGY CORPORATION 1998
●
●
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
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