LT1676I [Linear]
Wide Input Range, High Efficiency, Step-Down Switching Regulator; 宽输入范围，高效率，降压型开关稳压器
| 型号： | LT1676I |
| 厂家： | 
Linear |
| 描述： | Wide Input Range, High Efficiency, Step-Down Switching Regulator |
| 文件： | 总16页 (文件大小：139K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT1676
Wid e Inp ut Ra ng e ,
Hig h Effic ie nc y, Ste p -Do wn
Switc hing Re g ula to r
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FEATURES
DESCRIPTIO
The LT®1676 is a wide input range, high efficiency Buck
(step-down) switching regulator. The monolithic die in-
cludes all oscillator, control and protection circuitry. The
part can accept input voltages as high as 60V and contains
an output switch rated at 700mA peak current. Current
mode control offers excellent dynamic input supply rejec-
tion and short-circuit protection.
■
Wide Input Range: 7.4V to 60V
700mA Peak Switch Current Rating
Adaptive Switch Drive Maintains Efficiency at High
Load Without Pulse Skipping at Light Load
True Current Mode Control
100kHz Fixed Operating Frequency
Synchronizable to 250kHz
Low Supply Current in Shutdown: 30µA
■
■
■
■
■
■
The LT1676 contains several features to enhance effi-
ciency. The internal control circuitry is normally powered
via the VCC pin, thereby minimizing power drawn directly
■
Available in 8-Pin SO and PDIP Packages
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from the V supply (see Applications Information). The
IN
APPLICATIO S
■
action of the LT1676 switch circuitry is also load depen-
dent. At medium to high loads, the output switch circuitry
maintains highrisetimeforgoodefficiency. Atlightloads,
rise time is deliberately reduced to avoid pulse skipping
behavior.
Automotive DC/DC Converters
Telecom 48V Step-Down Converters
Cellular Phone Battery Charger Accessories
IEEE 1394 Step-Down Converters
■
■
■
The available SO-8 package and 100kHz switching fre-
quency allow for minimal PC board area requirements.
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATIO
V
Efficiency vs V and ILOAD
IN
IN
8V TO 50V
5
90
80
V
IN
1
2
3
SHDN
V
CC
220µH*
5V
400mA
36.5k
V
SW
70
+
+
39µF
63V
100µF
10V
MBR160
LT1676
60
50
40
30
20
1%
7
8
FB
6
SYNC
V
C
2200pF
V
= 12V
= 24V
IN
12.1k
1%
V
IN
100pF
22k
GND
4
V
= 36V
= 48V
IN
V
IN
1676 F01
*65T #30 ON MAGNETICS
MPP #55030
1
10
100
1000
I
(mA)
LOAD
1676 TA01
Figure 1
1
LT1676
W W
U W
U
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ABSOLUTE MAXIMUM RATINGS
(Note 1)
PACKAGE/ORDER INFORMATION
Supply Voltage ........................................................ 60V
Switch Voltage......................................................... 60V
SHDN, SYNC Pin Voltage........................................... 7V
ORDER PART
TOP VIEW
NUMBER
SHDN
1
2
3
4
8
7
6
5
V
C
LT1676CN8
LT1676CS8
LT1676IN8
LT1676IS8
V
CC
FB
V Pin Voltage ....................................................... 30V
CC
V
SW
SYNC
FB Pin Voltage ........................................................... 3V
Operating Junction Temperature Range
LT1676C................................................ 0°C to 125°C
LT1676I ............................................ –40°C to 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
GND
V
IN
N8 PACKAGE
8-LEAD PDIP
S8 PACKAGE
8-LEAD PLASTIC SO
S8 PART MARKING
T
T
JMAX = 125°C, θJA = 130°C/ W (N8)
JMAX = 125°C, θJA = 110°C/ W (S8)
1676
1676I
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
V = 48V, VSW open, VCC = 5V, V = 1.4V unless otherwise noted.
IN
C
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Power Supplies
Minimum Input Voltage
V
IN(MIN)
6.7
620
3.2
7.0
7.4
V
V
●
●
I
V
Supply Current
Supply Current
Dropout Voltage
V = 0V
800
900
µA
µA
VIN
IN
C
I
V
CC
V = 0V
C
4.0
5.0
mA
mA
VCC
●
●
V
V
CC
(Note 2)
2.8
30
3.1
V
VCC
Shutdown Mode I
V
SHDN
= 0V
50
75
µA
µA
VIN
●
●
Feedback Amplifier
Reference Voltage
V
1.225
1.215
1.240
1.255
1.265
V
V
REF
I
FB Pin Input Bias Current
600
650
1500
nA
IN
g
Feedback Amplifier Transconductance
∆lc = ±10µA
400
200
1000
1500
µmho
µmho
m
●
●
I
, I
Feedback Amplifier Source or Sink Current
60
45
100
2.0
170
220
µA
µA
SRC SNK
V
Feedback Amplifier Clamp Voltage
Reference Voltage Line Regulation
Voltage Gain
V
%/V
V/V
CL
12V ≤ V ≤ 60V
●
0.01
IN
200
600
Output Switch
V
Output Switch On Voltage
Switch Current Limit
I
= 0.5A
1.0
1.5
1.0
V
A
ON
SW
I
(Note 3)
●
0.55
0.9
0.70
LIM
Current Amplifier
Control Pin Threshold
Control Voltage to Switch Transconductance
Duty Cycle = 0%
1.1
2
1.25
V
A/V
2
LT1676
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
V = 48V, VSW open, VCC = 5V, V = 1.4V unless otherwise noted.
IN
C
SYMBOL
Timing
f
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Switching Frequency
90
85
100
110
115
kHz
kHz
●
●
Maximum Switch Duty Cycle
Minimum Switch On Time
85
90
%
ns
t
High dV/dt Mode, R = 50Ω (Note 4)
300
ON(MIN)
L
Boost Operation
V Pin Boost Threshold
1.35
0.2
V
V/ns
V/ns
C
dV/dt Below Threshold
dV/dt Above Threshold
1.6
Sync Function
Minimum Sync Amplitude
Synchronization Range
SYNC Pin Input R
●
●
1.5
40
2.2
V
kHz
kΩ
130
0.2
250
SHDN Pin Function
V
SHDN
Shutdown Mode Threshold
0.5
V
V
●
0.8
Upper Lockout Threshold
Lower Lockout Threshold
Shutdown Pin Current
Switching Action On
Switching Action Off
1.260
1.245
V
V
I
V
SHDN
= 0V
12
2.5
20
10
µA
µA
SHDN
V
SHDN
= 1.25V
Note 3: Switch current limit is DC trimmed and tested in production.
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Inductor dl/dt rate will cause a somewhat higher current limit in actual
application.
Note 2: Control circuitry powered from V .
CC
Note 4: Minimum switch on time is production tested with a 50Ω resistive
load to ground.
U W
TYPICAL PERFORMANCE CHARACTERISTICS
Minimum Input Voltage vs
Temperature
Switch-On Voltage vs
Switch Current
Switch Current Limit vs
Duty Cycle
1.50
1.25
1.00
0.75
0.50
0.25
0
7.4
7.2
7.0
6.8
6.6
6.4
6.2
6.0
1000
800
600
400
200
0
T = 25°C
A
25°C
–55°C
125°C
400
600 700
0
100 200 300
500
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
75
10 20 30
50 60 70
0
40
80 100
90
SWITCH CURRENT (mA)
DUTY CYCLE (%)
1676 G02
LT1676 G01
1676 G03
3
LT1676
TYPICAL PERFORMANCE CHARACTERISTICS
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SHDN Pin Shutdown Threshold
vs Temperature
SHDN Pin Input Current
vs Voltage
SHDN Pin Lockout Thresholds
vs Temperature
1.30
1.28
1.26
1.24
1.22
1.20
900
800
700
600
500
400
300
200
5
0
UPPER THRESHOLD
LOWER THRESHOLD
–5
–10
–15
–20
25°C
–55°C
125°C
50
TEMPERATURE (°C)
100 125
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
75
–50 –25
0
25
75
0
1
2
3
4
5
SHDN PIN VOLTAGE (V)
LT1676 G06
LT1676 G04
1676 G05
Switching Frequency
vs Temperature
Minimum Synchronization Voltage
vs Temperature
Switch Minimum On-Time
vs Temperature
600
500
400
300
200
100
0
106
104
102
100
98
2.25
2.00
1.75
1.50
1.25
1.00
0.75
V
= 48V
= 50Ω
IN
R
L
FB =
96
94
–50 –25
50
TEMPERATURE (°C)
100 125
50
TEMPERATURE (°C)
100 125
–50 –25
25
50
TEMPERATURE (°C)
75
100 125
0
25
75
–50 –25
0
25
75
0
1676 G09
1676 G07
1676 G08
V Pin Switching Threshold,
C
Boost Threshold, Clamp Voltage
vs Temperature
Feedback Amplifier Output
Current vs FB Pin Voltage
Error Amplifier Transconductance
vs Temperature
100
50
750
700
650
600
550
500
450
400
2.2
2.0
1.8
1.6
1.4
1.2
1.0
0.8
25°C
–55°C
125°C
CLAMP
VOLTAGE
0
BOOST
THRESHOLD
–50
–100
–150
SWITCHING
THRESHOLD
50
TEMPERATURE (°C)
100 125
1.0
1.1
1.2
1.3
1.4
1.5
–50 –25
0
25
75
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
75
FB PIN VOLTAGE (V)
1676 G11
LT1676 G12
LT1676 G10
4
LT1676
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PIN FUNCTIONS
SHDN (Pin 1): When pulled below the shutdown mode
threshold, nominally 0.30V, this pin turns off the regula-
short as possible to minimize electromagnetic radiation
and voltage spikes.
tor and reduces V input current to a few tens of micro-
amperes (shutdown mode).
IN
GND (Pin 4): This is the device ground pin. The internal
reference and feedback amplifier are referred to it. Keep
When this pin is held above the shutdown mode thresh-
old, but below the lockout threshold, the part will be
operational with the exception that output switching
action will be inhibited (lockout mode). A user-adjustable
undervoltage lockout can be implemented by driving this
the ground path connection to the FB divider and the V
compensation capacitor free of large ground currents.
C
V (Pin 5): This is the high voltage supply pin for the
IN
outputswitch.Italsosupplies powertotheinternalcontrol
circuitry during start-up conditions or if the V pin is left
CC
pin from an external resistor divider to V . This action is
IN
open. A high quality bypass capacitor that meets the input
ripple current requirements is needed here. (See Applica-
tions Information.)
logically“ANDed”withtheinternalUVLO,setatnominally
6.7V, such that minimum V can be increased above
IN
6.7V, but not decreased (see Applications Information).
SYNC (Pin 6): Pin used to synchronize internal oscillator
to the external frequency reference. It is directly logic
compatible and can be driven with any signal between
10% and 90% duty cycle. The sync function is internally
disabled if the FB pin voltage is low enough to cause
oscillatorslowdown.Ifunused,this pinshouldbegrounded.
If unused, this pin should be left open. However, the high
impedance nature of this pin renders it susceptible to
coupling from the high speed V node, so a small
SW
capacitor to ground, typically 100pF or so is recom-
mended when the pin is left “open.”
V (Pin 2): This pin is used to power the internal control
CC
FB (Pin 7): This is the inverting input to the feedback
amplifier. The noninverting input of this amplifier is inter-
nally tied to the 1.24V reference. This pin also slows down
the frequency of the internal oscillator when its voltage is
abnormally low, e.g., 2/3 of normal or less. This feature
helps maintain proper short-circuit protection.
circuitry off of the switching supply output. Proper use of
this pin enhances overall power supply efficiency. During
start-up conditions, internal control circuitry is powered
directly from V . If the output capacitor is located more
IN
than an inch from the VCC pin, a separate 0.1µF bypass
capacitor to ground may be required right at the pin.
V (Pin 8): This is the control voltage pin which is the
C
VSW (Pin 3): This is the emitter node of the output switch
output of the feedback amplifier and the input of the
currentcomparator.Frequencycompensationoftheover-
allloopis effectedbyplacingacapacitor, (orinmostcases
a series RC combination) between this node and ground.
and has large currents flowing through it. This node
moves at a high dV/dt rate, especially when in “boost”
mode. Keep the traces to the switching components as
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TIMING DIAGRAMS
High dV/dt Mode
Low dV/dt Mode
V
IN
V
IN
V
SW
V
SW
0
0
SWDR
SWON
BOOST
SWOFF
SWDR
SWON
BOOST
SWOFF
1676 TD01
1676 TD02
5
LT1676
W
BLOCK DIAGRA
2
1
5
V
IN
V
CC
R1
R
SENSE
V
BG
SHDN
BIAS
V
B
Q3
I
SWDR
COMP
Q4
Q2
SWDR
SWON
BOOST
SWOFF
Q1
OSC
LOGIC
SYNC
GND
6
4
V
SW
3
D1
I
SWON
BOOST
COMP
I
I
V
8
7
C
V
TH
FB
AMP
BOOST
FB
SWOFF
Q5
gm
I
V
BG
1676 BD
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OPERATIO
The LT1676 is a current mode switching regulator IC that
has been optimized for high efficiency operation in high
input voltage, low output voltage Buck topologies. The
Block Diagram shows an overall view of the system.
Several of the blocks are straightforward and similar to
those found in traditional designs, including: Internal Bias
Regulator, Oscillator and Feedback Amplifier. The novel
portion includes an elaborate Output Switch section and
Logic Section to provide the control signals required by
the switch section.
Output Switch Theory
One of the classic problems in delivering low output
voltage from high input voltage at good efficiency is that
minimizing AC switching losses requires very fast volt-
age (dV/dt) and current (dI/dt) transition at the output
device. This is in spite of the fact that in a bipolar
implementation, slow lateral PNPs must be included in
the switching signal path.
Fast positive-going slew rate action is provided by lateral
PNP Q3 driving the Darlington arrangement of Q1 and Q2.
The extra β available from Q2 greatly reduces the drive
requirements of Q3.
The LT1676 operates much the same as traditional
current mode switchers, the major difference being its
specialized output switch section. Due to space con-
straints, this discussion will not reiterate the basics of
current mode switcher/controllers and the “Buck” topol-
ogy. A good source of information on these topics is
Application Note 19.
Although desirable for dynamic reasons, this topology
alone will yield a large DC forward voltage drop. A second
lateral PNP, Q4, acts directly on the base of Q1 to reduce
the voltage drop after the slewing phase has taken place.
To achieve the desired high slew rate, PNPs Q3 and Q4 are
“force-fed” packets of charge via the current sources
controlled by the boost signal.
6
LT1676
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OPERATIO
Please refer to the High dV/dt Mode Timing Diagram. A threshold reference, V . (Remember that in a current
TH
typical oscillator cycle is as follows: The logic section first
generates an SWDR signal that powers up the current
comparatorandallows ittimetosettle.About1µs later,the
SWON signal is asserted and the BOOST signal is pulsed
for a few hundred nanoseconds. After a short delay, the
mode switching topology, the V voltage determines the
C
peak switch current.) When the V signal is above V , the
C
TH
previously described “high dV/dt” action is performed.
When the V signal is below V , the boost pulses are
C
TH
absent, as can be seen in the Low dV/dt Mode Timing
Diagram. Now the DC current, activated by the SWON
signal alone, drives Q4 and this transistor drives Q1 by
itself. The absence of a boost pulse, plus the lack of a
second NPN driver, result in a much lower slew rate which
aids light load controllability.
V
SW pin slews rapidly to V . Later, after the peak switch
IN
current indicated by the control voltage V has been
C
reached (current mode control), the SWON and SWDR
signals are turned off, and SWOFF is pulsed for several
hundred nanoseconds. The use of an explicit turn-off
device, i.e., Q5, improves turn-off response time and thus
aids both controllability and efficiency.
A further aid to overall efficiency is provided by the
specialized bias regulator circuit, which has a pair of
The system as previously described handles heavy loads
(continuous mode) at good efficiency, but it is actually
counterproductive for light loads. The method of jam-
ming charge into the PNP bases makes it difficult to turn
them off rapidly and achieve the very short switch ON
times required by light loads in discontinuous mode.
Furthermore, the high leading edge dV/dt rate similarly
adversely affects light load controllability.
inputs, V and V . The V pin is normally connected to
the switching supply output. During start-up conditions,
IN CC CC
the LT1676 powers itself directly from V . However, after
IN
the switching supply output voltage reaches about 2.9V,
the bias regulator uses this supply as its input. Previous
generation Buck controller ICs without this provision
typically required hundreds of milliwatts of quiescent
power when operating at high input voltage. This both
degraded efficiency and limited available output current
due to internal heating.
The solution is to employ a “boost comparator” whose
inputs are the V control voltage and a fixed internal
C
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APPLICATIONS INFORMATION
Selecting a Power Inductor
V
V – V
IN OUT
OUT
L =
There are several parameters to consider when selecting
a power inductor. These include inductance value, peak
current rating (to avoid core saturation), DC resistance,
construction type, physical size, and of course, cost.
f•I
V
IN
PK
For example, substituting 48V, 5V, 200mA and 100kHz
respectively for V , VOUT, IPK and f yields a value of about
220µH. Notethatthelefthalfofthis expressionis indepen-
IN
Inatypicalapplication,properinductancevalueis dictated
bymatchingthediscontinuous/continuous crossoverpoint
withtheLT1676internallow-to-highdV/dtthreshold. This
is the best compromise between maintaining control with
light loads while maintaining good efficiency with heavy
loads. The fixed internal dV/dt threshold has a nominal
dent of input voltage while the right half is only a weak
function of V when V is much greater than VOUT. This
IN
IN
means that a single inductor value will work well over a
range of “high” input voltage. And although a progres-
sively smaller inductor is suggested as V begins to
IN
approach VOUT, note that the much higher ON duty cycles
under these conditions are much more forgiving with
respect to controllability and efficiency issues. Therefore
when a wide input voltage range must be accommodated,
say 10V to 50V for 5VOUT, the user should choose an
inductance value based on the maximum input voltage.
value of 1.4V, which referred to the V pin threshold and
C
control voltage to switch transconductance, corresponds
to a peak current of about 200mA. Standard Buck con-
verter theory yields the following expression for induc-
tance at the discontinuous/continuous crossover:
7
LT1676
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APPLICATIONS INFORMATION
Once the inductance value is decided, inductor peak
current rating and resistance need to be considered. Here,
the inductor peak current rating refers to the onset of
saturation in the core material, although manufacturers
sometimes specify a “peak current rating” which is
derived from a worst-case combination of core saturation
andself-heatingeffects.Inductorwindingresistancealone
limits the inductor’s current carrying capability as the I2R
power threatens to overheat the inductor. If applicable,
remember to include the condition of output short circuit.
Although the peak current rating of the inductor can be
exceeded in short-circuit operation, as core saturation per
se is not destructive to the core, excess resistive self-
heating is still a potential problem.
internalswitchwillrampupV currentintothediodeinan
attempt to get it to recover. Then, when the diode has
IN
finallyturnedoff,sometens ofnanoseconds later,theV
SW
node voltage ramps up at an extremely high dV/dt, per-
haps 5 to even 10V/ns ! With real world lead inductances,
the VSW node can easily overshoot the V rail. This can
IN
result in poor RFI behavior and if the overshoot is severe
enough, damage the IC itself.
Selecting Bypass Capacitors
The basic topology as shown in Figure 1 uses two bypass
capacitors, one for the V input supply and one for the
IN
VOUT output supply.
User selection of an appropriate output capacitor is rela-
tivelyeasy,as this capacitorsees onlytheACripplecurrent
in the inductor. As the LT1676 is designed for Buck or
step-down applications, output voltage will nearly always
be compatible with tantalum type capacitors, which are
generally available in ratings up to 35V or so. These
tantalum types offer good volumetric efficiency and many
areavailablewithspecifiedESRperformance.Theproduct
ofinductorACripplecurrentandoutputcapacitorESRwill
manifestitselfas peak-to-peakvoltagerippleontheoutput
node. (Note: If this ripple becomes too large, heavier
control loop compensation, at least at the switching fre-
The final inductor selection is generally based on cost,
which usually translates into choosing the smallest physi-
cal size part that meets the desired inductance value,
resistance and current carrying capability. An additional
factor to consider is that of physical construction. Briefly
stated, “open” inductors built on a rod- or barrel-shaped
core generally offer the smallest physical size and lowest
cost. However their open construction does not contain
the resulting magnetic field, and they may not be accept-
able in RFI-sensitive applications. Toroidal style induc-
tors, many available in surface mount configuration, offer
improved RFI performance, generally at an increase in
cost and physical size. And although custom design is
always a possibility, most potential LT1676 applications
can be handled by the array of standard, off-the-shelf
inductor products offered by the major suppliers.
quency, may be required on the V pin.) The most
C
demanding applications, requiring very low output ripple,
may be best served not with a single extremely large
output capacitor, but instead by the common technique of
a separate L/C lowpass post filter in series with the output.
(In this case, “Two caps are better than one.”)
Selecting Freewheeling Diode
The input bypass capacitor is normally a more difficult
Highestefficiencyoperationrequires theuseofaSchottky
type diode. DC switching losses are minimized due to its
low forward voltage drop, and AC behavior is benign due
to its lack of a significant reverse recovery time. Schottky
diodes are generally available with reverse voltage ratings
of60Vandeven100V,andarepricecompetitivewithother
types.
choice. In a typical application e.g., 48V to 5VOUT
,
IN
relatively heavy V current is drawn by the power switch
IN
for only a small portion of the oscillator period (low ON
duty cycle). The resulting RMS ripple current, for which
the capacitor must be rated, is often several times the DC
average V current. Similarly, the “glitch” seen on the V
IN
IN
supply as the power switch turns on and off will be related
The use of so-called “ultrafast” recovery diodes is gener-
ally not recommended. When operating in continuous
mode, the reverse recovery time exhibited by “ultrafast”
diodes will result in a slingshot type effect. The power
to the product of capacitor ESR, and the relatively high
instantaneous current drawn by the switch. To compound
these problems is the fact that most of these applications
will be designed for a relatively high input voltage, for
8
LT1676
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APPLICATIONS INFORMATION
whichtantalumcapacitors aregenerallyunavailable.Rela-
tively bulky “high frequency” aluminum electrolytic types,
specifically constructed and rated for switching supply
applications, may be the only choice.
tON(MIN). When combined with the large ratio of V to
IN
(V + I • R), the diode forward voltage plus inductor I • R
F
voltage drop, the potential exists for a loss of control.
Expressed mathematically the requirement to maintain
control is:
Minimum Load Considerations
V +I•R
F
f• t
≤
As discussed previously, a lightly loaded LT1676 with V
ON
C
V
IN
pin control voltage below the boost threshold will operate
in low dV/dt mode. This affords greater controllability at
light loads, as minimum tON requirements are relaxed. In
many applications, it is possible to operate the LT1676
down to zero external load without “pulse skipping”!
In these cases, the LT1676’s modest VCC current
requirement of several milliamperes provides enough of a
load to avoid pulse skipping.
where:
f = switching frequency
ON = switch ON time
V = diode forward voltage
t
F
V = Input voltage
IN
I • R = inductor I • R voltage drop
If this condition is not observed, the current will not be
limited at IPK, but will cycle-by-cycle ratchet up to some
higher value. Using the nominal LT1676 clock frequency
However, some users may be indifferent to pulse skipping
behavior, but instead may be concerned with maintaining
maximum possible efficiency at light loads. This require-
mentcanbesatisfiedbyforcingthepartintoBurstModeTM
operation. The use of an external comparator whose
output controls the shutdown pin allows high efficiency at
light loads through Burst Mode operation behavior (see
Typical Applications and Figure 8).
of 100KHz, a V of 48V and a (V + I • R) of say 0.7V, the
IN
F
maximum tON to maintain control would be approximately
140ns, an unacceptably short time.
The solution to this dilemma is to slow down the oscillator
when the FB pin voltage is abnormally low thereby indicat-
ing some sort of short-circuit condition. Figure 2 shows
the typical response of Oscillator Frequency vs FB divider
Thevenin voltage and impedance. Oscillator frequency is
unaffecteduntilFBvoltagedrops toabout2/3ofits normal
value. Below this point the oscillator frequency decreases
roughly linearly down to a limit of about 25kHz. This lower
Maximum Load/Short-Circuit Considerations
The LT1676 is a current mode controller. It uses the V
C
node voltage as an input to a current comparator which
turns off the output switch on a cycle-by-cycle basis as
this peak current is reached. The internal clamp on the V
C
node, nominally 2V, then acts as an output switch peak
current limit. This action becomes the switch current limit
specification. The maximum available output power is
then determined by the switch current limit.
120
100
R
= 22k
TH
80
60
R
TH
= 10k
R
= 4.7k
TH
A potential controllability problem could occur under
short-circuit conditions. If the power supply output is
short circuited, the feedback amplifier responds to the low
40
20
0
LT1676
FB
R
TH
output voltage by raising the control voltage, V , to its
C
peak current limit value. Ideally, the output switch would
be turned on, and then turned off as its current exceeded
0
0.25
0.50
0.75
1.00
1.25
thevalueindicatedbyV .However,thereis finiteresponse
C
FB DIVIDER THEVENIN VOLTAGE (V)
time involved in both the current comparator and turnoff
of the output switch. These result in a minimum on time
1676 F02
Figure 2. Oscillator Frequency vs FB Divider
Thevenin Voltage and Impedance
Burst Mode is a trademark of Linear Technology Corporation.
9
LT1676
U
W U U
APPLICATIONS INFORMATION
oscillator frequency during short-circuit conditions can Power loss internal to the LT1676 related to actual output
current is composed of both DC and AC switching losses.
These can be roughly estimated as follows:
thenmaintaincontrolwiththeeffectiveminimumONtime.
A further potential problem with short-circuit operation
might occur if the user were operating the part with its
oscillator slaved to an external frequency source via the
SYNC pin. However, the LT1676 has circuitry that auto-
matically disables the sync function when the oscillator is
slowed down due to abnormally low FB voltage.
DC switching losses are dominated by output switch “ON
voltage”, i.e.,
P
DC = VON • IOUT • DC
VON = Output switch ON voltage, typically 1V at 500mA
IOUT = Output current
Feedback Divider Considerations
DC = ON duty cycle
AnLT1676applicationtypicallyincludes aresistivedivider AC switching losses are typically dominated by power lost
due to the finite rise time and fall time at the VSW node.
Assuming, for simplicity, a linear ramp up of both voltage
betweenVOUT andground, thecenternodeofwhichdrives
the FB pin to the reference voltage VREF. This establishes
a fixed ratio between the two resistors, but a second and current and a current rise/fall time equal to 15ns,
degree of freedom is offered by the overall impedance
PAC = 1/2 • V • IOUT • (tr + tf + 30ns) • f
IN
level of the resistor pair. The most obvious effect this has
is one of efficiency—a higher resistance feedback divider
will waste less power and offer somewhat higher effi-
ciency, especially at light load.
tr = (V /1.6)ns in high dV/dt mode
IN
(V /0.16)ns in low dV/dt mode
IN
tf = (V /1.6)ns (irrespective of dV/dt mode)
IN
f = switching frequency
However, remember that oscillator slowdown to achieve
short-circuit protection (discussed above) is dependent
on FB pin behavior, and this in turn, is sensitive to FB node
external impedance. Figure 2 shows the typical relation-
ship between FB divider Thevenin voltage and impedance,
and oscillator frequency. This shows that as feedback
network impedance increases beyond 10k, complete os-
cillator slowdown is not achieved, and short-circuit pro-
tection may be compromised. And as a practical matter,
the product of FB pin bias current and larger FB network
impedances will cause increasing output voltage error.
(Nominal cancellation for 10k of FB Thevenin impedance
is included internally.)
Total power dissipation of the die is simply the sum of
quiescent, DC and AC losses previously calculated.
PD(TOTAL) = PQ + PDC + PAC
Frequency Compensation
Loop frequency compensation is performed by connect-
ing a capacitor, or in most cases a series RC, from the
output of the error amplifier (V pin) to ground. Proper
C
loop compensation may be obtained by empirical meth-
ods as described in detail in Application Note 19. Briefly,
this involves applying a load transient and observing the
dynamic response over the expected range of V and
IN
ILOAD values.
Thermal Considerations
As a practical matter, a second small capacitor, directly
Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause exces-
sive die temperatures. The packages are rated at 110°C/W
for the 8-pin SO (S8) and 130°C/W for 8-pin PDIP (N8).
from the V pin to ground is generally recommended to
C
attenuate capacitive coupling from the VSW pin. A typical
value for this capacitor is 100pF. (See Switch Node Con-
siderations).
Quiescent power is given by:
Switch Node Considerations
PQ = IVIN • V + IVCC • VOUT
IN
For maximum efficiency, switch rise and fall times are
made as short as practical. To prevent radiation and high
(This assumes that the V pin is connected to VOUT.)
CC
10
LT1676
U
W
U U
APPLICATIONS INFORMATION
frequency resonance problems, proper layout of the com-
ponents connected to the IC is essential, especially the
i.e., SHDN, SYNC, V and FB. This can cause erratic
operation such as odd/even cycle behavior, pulse width
C
power path. B field (magnetic) radiation is minimized by “nervousness”, improper output voltage and/or prema-
keeping output diode, switch pin and intput bypass
capacitor leads as short as possible. E field radiation is
kept low by minimizing the length and area of all traces
ture current limit action.
As an example, assume that the capacitance between the
VSW node and a high impedance pin node is 0.1pF, and
connected to the switch pin (V ). A ground plane should
SW
further assume that the high impedance node in question
exhibits a capacitance of 1pF to ground. Due to the high
dV/dt, large excursion behavior of the VSW node, this will
couple a nearly 5V transient to the high impedance pin,
causing abnormal operation. (This assumes the “typical”
always be used under the switcher circuitry to prevent
interplane coupling.
Thehighspeedswitchingcurrentpathis shownschemati-
cally in Figure 3. Minimum lead length in these paths is
essential to ensure clean switching and minimal EMI. The
paths containing the input capacitor, output switch and
outputdiodearetheonlyones containingnanosecondrise
and fall times. Keep these paths as short as possible.
48V to 5VOUT application.) An explicit 100pF capacitor
IN
added to the node will reduce the amplitude of the distur-
bance to more like 50mV (although settling time will
increase).
Additionally, it is possible for the LT1676 to cause EMI Specific pin recommendations are as follows:
problems by “coupling to itself”. Specifically, this can
occur if the VSW pin is allowed to capacitively couple in an
uncontrolled manner to the part’s high impedance nodes,
SHDN: If unused, add a 100pF capacitor to ground.
SYNC: Ground if unused.
V : Add a capacitor directly to ground in addition to the
C
explicit compensation network. A value of one-tenth of
the main compensation capacitor is recommended, up
to a maximum of 100pF.
V
IN
+
V
IN
C1
LT1676
L1
V
SW
V
OUT
+
FB: Assuming the V pin is handled properly, this pin
C
D1
C2
usually requires no explicit capacitor of its own, but
keep this node physically small to minimize stray ca-
pacitance.
1676 F03
Figure 3. High Speed Current Switching Paths
U
TYPICAL APPLICATIONS
Minimum Component Count Application
User Programmable Undervoltage Lockout
Figure 4a shows a basic “minimum component count”
application. The circuit produces 5V at up to 500mA IOUT
with input voltages in the range of 12V to 48V. The typical
POUT/PIN efficiency is shown in Figure 4b. No pulse
skipping is observed down to zero external load. As
shown, the SHDN and SYNC pins are unused, however
either(orboth)canbeoptionallydrivenbyexternalsignals
as desired.
Figure 5 adds a resistor divider to the basic application.
This is asimple,cost-effectivewaytoaddauser-program-
mable undervoltage lockout (UVLO) function. Resistor R5
is chosen to have approximately 200µA through it at the
nominal SHDN pin lockout threshold of roughly 1.25V.
The somewhat arbitrary value of 200µA was chosen to be
significantlyabovetheSHDNpininputcurrenttominimize
its error contribution, but significantly below the typical
3.2mA the LT1676 draws in lockout mode. Resistor R4 is
then chosen to yield this same 200µA, less 2.5µA, with the
11
LT1676
TYPICAL APPLICATIONS
U
V
IN
12V TO
48V
5
V
IN
1
2
3
SHDN
V
CC
V
OUT
V
SW
5V
+
L1
220µH
+
C1
39µF
63V
C2
100µF
10V
0mA to 500mA
90
80
70
60
50
40
30
20
D1
R1
36.5k
1%
C5
100pF
LT1676
MBRS1100
7
8
FB
6
SYNC
V
C
C3
R2
12.1k
1%
2200pF
X7R
C4
100pF
R3
22k
5%
GND
4
V
= 12V
= 24V
IN
1676 F04a
V
IN
C1: PANASONIC HFQ
FOR 3.3V V
VERSION:
V
= 36V
= 48V
OUT
IN
C2: AVX D CASE TPSD107M010R0080
C4, C5: X7R OR COG/NPO
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
L1: COILCRAFT DO3316P-224
R1: 24.3K, R2: 14.7k
L1: 150µH, DO3316P-154
: 0mA TO 500mA
V
IN
I
OUT
1
10
100
1000
I
(mA)
LOAD
1676 F04b
Figure 4a. Minimum Component Count Application
Figure 4b. POUT/PIN Efficiency
V
IN
+
R4
5
C1
C5
210k
1%
V
IN
1
6
2
3
SHDN
V
CC
L1
V
SW
V
OUT
+
D1
LT1676
C2
R1
R5
6.19k
1%
7
8
FB
SYNC
V
C
C3
R3
R2
GND
4
C4
1676 F05
Figure 5. User Programmable Undervoltage Lockout
desired V UVLO voltage minus 1.25V applied across it.
the SHDN pin will drop to the shutdown threshold, and the
IN
(The 2.5µA factor is an allowance to minimize error due to
part will draw its shutdown current only from the V rail.
IN
SHDN pin input current.)
The resistive divider of R4 and R5 will continue to draw
power from V . (The user should be aware that while the
IN
Behavioris as follows:Normaloperationis observedatthe
nominal input voltage of 48V. As the input voltage is
decreasedtoroughly43V, switchingactionwillstop, VOUT
SHDN pin lockout threshold is relatively accurate includ-
ing temperature effects, the SHDN pin shutdown thresh-
old is more coarse, and exhibits considerably more
temperature drift. Nevertheless the shutdown threshold
will always be well below the lockout threshold.)
will drop to zero, and the LT1676 will draw its V and V
IN
CC
quiescent currents from the V supply. At a much lower
IN
input voltage, typically 18V or so at 25°C, the voltage on
12
LT1676
U
TYPICAL APPLICATIONS
A possible way to improve light load efficiency is in Burst
Mode operation.
Micropower Undervoltage Lockout
Certain applications may require very low current drain
when in undervoltage lockout mode. This can be accom-
plished with the addition of a few more external compo-
nents. Figure 6 shows an LTC®1440 micropower
comparator/reference added to control the LT1676 via its
SHDN pin. The extremely low input bias current of the
CMOS comparator allows the impedance of the resistor
divider R4/R5 to be increased, thereby minimizing power
drain. Hysteresis is externally programmable via resistor
divider R6/R7. The LTC1440 output directly controls the
LT1676 via its shutdown pin, driving it to either 5V (ON) or
0V (Full Shutdown). A simple linear voltage regulator to
power the LTC1440 is provided by Q1, Q2 and R7. Just
below the UVLO threshold, nominally 43V, total current
drain is typically 50µA.
Figure 7 shows the LT1676 configured for Burst Mode
operation. Output voltage regulation is now provided in a
“bang-bang” digital manner, via comparator U2, an
LTC1440. Resistor divider R3/R4 provides a scaled ver-
sionoftheoutputvoltage, whichis comparedagainstU2’s
internal reference. Intentional hysteresis is set by the R5/
R6 divider. As the output voltage falls below the regulation
range, the LT1676 is turned on. The output voltage rises,
and as it climbs above the regulation range, the LT1676 is
turned off. Efficiency is maximized, as the LT1676 is only
powered up while it is providing heavy output current.
Figure 7b shows that efficiency is typically maintained at
75% or better down to a load current of 10mA. Even at a
load of 1mA, efficiency is still a respectable 59% to 68%,
depending on V .
IN
Burst Mode Operation Configuration
Resistor divider R1/R2 is still present, but does not
directly influence output voltage. It is chosen to ensure
that the LT1676 delivers high output current throughout
the voltage regulation range. Its presence is also required
Figure 4b demonstrates that power supply efficiency de-
grades with lower output load current. This is not surpris-
ing,as theLT1676itselfrepresents afixedpoweroverhead.
V
IN
5
V
IN
2
3
6
1
SYNC
V
CC
R8
10M
V
5V
OUT
V
SW
+
L1
220µH
C1
39µF
63V
+
C2
100µF
10V
D1
R1
36.5
1%
U1
LT1676
MBRS1100
7
8
FB
Q1
SHDN
V
C
PN2484
R2
C3
12.1k
1%
2200pF
R3
22k
C4
100pF
GND
4
Q2
2N2369
NC
7
+
V
8
3
4
V
+
–
IN
OUT
IN
IN
C1: PANASONIC HFQ
C2: AVX D CASE TPSD107M010R0080
C4, C5: X7R OR COG/NPO
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
L1: COILCRAFT DO3316P-224
U2
LTC1440
R6
22k
R4
6.8M
6
5
REF
HYST
GND
–
V
R7
R5
2.4M
240k
2
1
1676 F06
Figure 6. Micropower Undervoltage Lockout
13
LT1676
U
TYPICAL APPLICATIONS
to maintain proper short-circuit protection. Transistors
Minimum Size Inductor Application
Q1, Q2 and resistor R7 form a high V , low quiescent
IN
Figure 4a employs power path parts that are capable of
delivering the full rated output capability of the LT1676.
Potential users with low output current requirements may
be interested in substituting a physically smaller and less
costly power inductor. The circuit shown on the last page
of this data sheet is topologically identical to the basic
application, but specifies a much smaller inductor, and, a
somewhat smaller input electrolytic capacitor. This circuit
is capableofdeliveringupto150mAat5V,or,upto200mA
at 3.3V. The only disadvantage is that due to the increased
resistance in the inductor, the circuit is no longer capable
of withstanding indefinite short circuits to ground. The
LT1676 will still current limit at its nominal ILIM value, but
this will overheat the inductor. Momentary short circuits
of a few seconds or less can still be tolerated.
current voltage regulator to power U2.
Burst Mode Operation Configuration with UVLO
Figure 7a uses an external comparator to control the
LT1676 via its SHDN pin. As such, the user’s ability to set
an undervoltage lockout (UVLO) threshold with a resistor
divider from V to SHDN pin to ground is lost. This ability
IN
is regained in the slightly more complicated circuit shown
in Figure 8.
A dual comparator, the LTC1442, replaces the previous
single comparator. The second comparator monitors a
resistive divider between V and ground to provide the
IN
(user-adjustable) UVLO function. The two comparator
outputs are logically combined in a CMOS NOR gate (U3)
to drive the LT1676 SHDN pin.
V
IN
5
+
C1
V
IN
6
1
2
3
R7
10M
SYNC
V
CC
L1
V
OUT
V
SW
5V
+
U1
LT1676
R1
39k
5%
D1
C2
Q1
PN2484
7
8
FB
90
80
70
60
50
40
30
20
SHDN
V
C
R2
10k
5%
R3
323k
1%
GND
4
Q2
2N2369
C3
100pF
V
IN
= 12V
V
= 48V
IN
NC
7
V
= 36V
IN
+
V
IN
= 24V
V
8
3
4
+
–
OUT
IN
IN
C1: PANASONIC HFQ 39µF AT 63V
C2: AVX D CASE 100µF 10V
TPSD107M010R0080
D1: MOTOROLA 100V, 1A,
SMD SCHOTTKY
U2
R5
22k
LTC1440
6
5
R4
100k
1%
REF
HYST
GND
MBRS1100 (T3)
L1: COILCRAFT DO3316-224
–
V
R6
1
10
100
1000
2
1
2.4M
I
(mA)
LOAD
1676 F07a
1676 F07b
(a)
(b)
Figure 7. Burst Mode Operation Configuration for High Efficiency at Light Load
14
LT1676
U
TYPICAL APPLICATIONS
V
IN
5
V
IN
6
1
2
3
SYNC
V
CC
+
R7
10M
L1
C1
V
OUT
5V
V
SW
+
U1
LT1676
R1
39k
D1
C2
Q1
PN2484
7
8
FB
SHDN
V
C
Q2
2N2369
R2
10k
GND
4
C3
V
IN
NC
R3
R8
6.8M
323k
1%
+
V
+
OUTA
INA
C1: PANASONIC HFQ 39µF AT 63V
C2: AVX D CASE 100µF 10V
TPSD107M010R0080
D1: MOTOROLA 100V, 1A,
SMD SCHOTTKY
5
3
–
INB
1
2
U2
LTC1442
R5
4
22k
R4
REF
U3
7S02
100k
1%
MBRS1100 (T3)
L1: COILCRAFT DO3316-224
OUTB
HYST
–
V
R9
R6
240k
2.4M
1676 F08
Figure 8. Burst Mode Operation Configuration with Micropower UVLO
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.400*
(10.160)
MAX
0.130 ± 0.005
0.300 – 0.325
0.045 – 0.065
(3.302 ± 0.127)
(1.143 – 1.651)
(7.620 – 8.255)
8
7
6
5
0.065
(1.651)
TYP
0.255 ± 0.015*
(6.477 ± 0.381)
0.009 – 0.015
0.125
(0.229 – 0.381)
0.020
(3.175)
MIN
+0.035
–0.015
(0.508)
MIN
1
2
4
3
0.325
0.100 ± 0.010
(2.540 ± 0.254)
0.018 ± 0.003
+0.889
–0.381
8.255
(
)
(0.457 ± 0.076)
N8 1197
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
0.010 – 0.020
(0.254 – 0.508)
7
5
8
6
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
0.008 – 0.010
(0.203 – 0.254)
(0.101 – 0.254)
0°– 8° TYP
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
0.016 – 0.050
0.406 – 1.270
0.050
(1.270)
TYP
0.014 – 0.019
(0.355 – 0.483)
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
SO8 0996
1
2
3
4
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe onexisting patent rights.
15
LT1676
TYPICAL APPLICATION
U
Minimum Inductor Size Application
V
IN
12V TO
48V
5
V
IN
1
2
3
SHDN
V
CC
V
5V
OUT
C5
100pF
V
SW
+
L1
220µH
C2
100µF
10V
0mA to 150mA
R1
36.5k
1%
D1
+
C1
12µF
63V
LT1676
7
8
FB
6
SYNC
V
C
C3
2200pF
X7R
R2
12.1k
1%
C4
100pF
R3
GND
4
22k
5%
1676 TA02
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY FOR 3.3V V
VERSION:
: 0mA TO 200mA
C1: PANASONIC HFQ
C2: AVX D CASE TPSD107M010R0080
C4, C5: X7R OR COG/NPO
OUT
MBRS1100 (T3)
I
OUT
L1: COILCRAFT DO1608C-224
L1: 150µH, DO1608C-154
R1: 24.3K, R2: 14.7k
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1676f LT/TP 0499 4K • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1998
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
16
●
●
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
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