LT6108IDCB-1#PBF_技术文档

LT6108-1/LT6108-2

High Side Current Sense

Ampliﬁer with Reference

and Comparator

FEATURES

DESCRIPTION

The LT^®6108 is a complete high side current sense device

that incorporates a precision current sense ampliﬁer, an

integrated voltage reference and a comparator. Two ver-

sions of the LT6108 are available. The LT6108-1 has a

latching comparator and the LT6108-2 has a non-latching

comparator. In addition, the current sense ampliﬁer and

comparatorinputsandoutputsaredirectlyaccessible.The

ampliﬁer gain and comparator trip point are conﬁgured

by external resistors. The open-drain comparator output

allows for easy interface to other system components.

n

Current Sense Ampliﬁer

– Fast Step Response: 500ns

– Low Offset Voltage: 125µV Maximum

– Low Gain Error: 0.2% Maximum

n

Internal 400mV Precision Reference

n

Internal Comparator

– Fast Response Time: 500ns

– Total Threshold Error: 1.25% Maximum

– Latching or Non-Latching Comparator Option

n

Wide Supply Range: 2.7V to 60V

n

Supply Current: 450µA

Low Shutdown Current: 5µA Maximum

TheoverallpropagationdelayoftheLT6108istypicallyonly

1.4µs, allowing for quick reaction to overcurrent condi-

tions. The 1MHz bandwidth allows the LT6108 to be used

for error detection in critical applications such as motor

control. The high threshold accuracy of the comparator,

combined with the ability to latch the comparator, ensures

the LT6108 can capture high speed events.

n

Speciﬁed for –40°C to 125°C Temperature Range

n

Available in 8-Lead MSOP and 8-Lead (2mm × 3mm)

DFN Packages

APPLICATIONS

n

Overcurrent and Fault Detection

The LT6108 is fully speciﬁed for operation from –40°C to

125°C, making it suitable for industrial and automotive

applications. The LT6108 is available in the small 8-lead

MSOP and 8-lead DFN packages.

L, LT, LTC, LTM, TimerBlox, Linear Technology and the Linear logo are registered trademarks

of Linear Technology Corporation. All other trademarks are the property of their respective

owners.

n

Current Shunt Measurement

n

Battery Monitoring

Motor Control

Automotive Monitoring and Control

Remote Sensing

Industrial Control

n

TYPICAL APPLICATION

Response to Overcurrent Event

Circuit Fault Protection with Very Fast Latching Load Disconnect

0.1Ω

IRF9640

12V

TO LOAD

V

LOAD

0.1µF

1k

100Ω

10V/DIV

6.2V*

0V

SENSEHI SENSELO

+

V

OUTA

OUT

I

3.3V

LOAD

LT6108-1

200mA/DIV

6.04k

1.6k

RESET

EN/RST

1k

10k

0mA

250mA DISCONNECT

V

OUTC

5V/DIV

250mA DISCONNECT

2N2700

OUTC

INC

–

0V

V

610812 TA01a

610812 TA01b

5µs/DIV

*CMH25234B

610812fa

1

LT6108-1/LT6108-2

ABSOLUTE MAXIMUM RATINGS ^{(Note 1)}

+

–

Total Supply Voltage (V to V ).................................60V

AmpliﬁerOutputShort-CircuitDuration(toV ).. Indeﬁnite

Maximum Voltage

Operating Temperature Range (Note 3)

+

(SENSELO, SENSEHI, OUTA)............................... V + 1V

LT6108I................................................–40°C to 85°C

LT6108H ............................................ –40°C to 125°C

Speciﬁed Temperature Range (Note 3)

+

Maximum V – (SENSELO or SENSEHI)....................33V

Maximum EN, EN/RST Voltage .................................60V

Maximum Comparator Input Voltage........................60V

Maximum Comparator Output Voltage......................60V

Input Current (Note 2)..........................................–10mA

SENSEHI, SENSELO Input Current ....................... 10mA

Differential SENSEHI or SENSELO Input Current .. 2.5mA

LT6108I................................................–40°C to 85°C

LT6108H ............................................ –40°C to 125°C

Maximum Junction Temperature .......................... 150°C

Storage Temperature Range .................. –65°C to 150°C

MSOP Lead Temperature (Soldering, 10 sec)........300°C

PIN CONFIGURATION

LT6108-1

LT6108-2

TOP VIEW

SENSELO

EN/RST

OUTC

1

2

3

4

8 SENSEHI

7 V

6 OUTA

5 INC

SENSELO

EN

1

2

3

4

8 SENSEHI

7 V

6 OUTA

5 INC

+

OUTC

–

V

MS8 PACKAGE

8-LEAD PLASTIC MSOP

MS8 PACKAGE

8-LEAD PLASTIC MSOP

θ

JA

= 163°C/W, θ = 45°C/W

θ

JA

= 163°C/W, θ = 45°C/W

JC

TOP VIEW

8

SENSEHI

SENSELO

EN/RST

OUTC

1

2

3

4

SENSEHI

SENSELO

EN

1

2

3

4

+

7

6

5

V

7

6

5

V

9

OUTA

INC

OUTC

OUTA

INC

–

V

DCB PACKAGE

8-LEAD (2mm × 3mm) PLASTIC DFN

DCB PACKAGE

8-LEAD (2mm × 3mm) PLASTIC DFN

θ

= 64°C/W, θ = 10°C/W

θ

= 64°C/W, θ = 10°C/W

JA

JC

JA

JC

–

EXPOSED PAD (PIN 9) IS V , PCB CONNECTION OPTIONAL

610812fa

2

LT6108-1/LT6108-2

ORDER INFORMATION

LEAD FREE FINISH

LT6108AIMS8-1#PBF

LT6108IMS8-1#PBF

LT6108AHMS8-1#PBF

LT6108HMS8-1#PBF

LT6108AIMS8-2#PBF

LT6108IMS8-2#PBF

LT6108AHMS8-2#PBF

LT6108HMS8-2#PBF

TAPE AND REEL

PART MARKING* PACKAGE DESCRIPTION

SPECIFIED TEMPERATURE RANGE

–40°C to 85°C

LT6108AIMS8-1#TRPBF

LT6108IMS8-1#TRPBF

LT6108AHMS8-1#TRPBF

LT6108HMS8-1#TRPBF

LT6108AIMS8-2#TRPBF

LT6108IMS8-2#TRPBF

LT6108AHMS8-2#TRPBF

LT6108HMS8-2#TRPBF

LTFND

LTFNG

8-Lead Plastic MSOP

–40°C to 85°C

–40°C to 125°C

–40°C to 85°C

–40°C to 125°C

Consult LTC Marketing for parts speciﬁed with wider operating temperature ranges. *The temperature grade is identiﬁed by a label on the shipping container.

Consult LTC Marketing for information on non-standard lead based ﬁnish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel speciﬁcations, go to: http://www.linear.com/tapeandreel/

Lead Free Finish

TAPE AND REEL (MINI)

TAPE AND REEL

PART MARKING*

LFNF

PACKAGE DESCRIPTION

SPECIFIED TEMPERATURE RANGE

–40°C to 85°C

LT6108IDCB-1#TRMPBF

LT6108IDCB-1#TRPBF

8-Lead (2mm × 3mm) Plastic DFN

LT6108HDCB-1#TRMPBF LT6108HDCB-1#TRPBF

LT6108IDCB-2#TRMPBF LT6108IDCB-2#TRPBF

LT6108HDCB-2#TRMPBF LT6108HDCB-2#TRPBF

LFNF

–40°C to 125°C

LFNH

–40°C to 85°C

LFNH

–40°C to 125°C

TRM = 500 pieces. *Temperature grades are identiﬁed by a label on the shipping container.

Consult LTC Marketing for parts speciﬁed with wider operating temperature ranges.

Consult LTC Marketing for information on lead based ﬁnish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel speciﬁcations, go to: http://www.linear.com/tapeandreel/

610812fa

3

LT6108-1/LT6108-2

ELECTRICAL CHARACTERISTICS ^Thel denotes the speciﬁcations which apply over the full operating

temperature range, otherwise speciﬁcations are at T_A= 25°C. V⁺= 12V, V_PULLUP= V⁺, V_EN= V_EN/RST= 2.7V, R_IN= 100Ω,

R_OUT= R1 + R2 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless otherwise noted. (See Figure 3)

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

V

+

l

V

Supply Voltage Range

Supply Current (Note 4)

2.7

60

+

I

S

V = 2.7V, R = 1k, V

= 5mV

450

550

µA

IN

SENSE

+

V = 60V, R = 1k, V

650

950

µA

IN

SENSE

l

+

Supply Current in Shutdown

V = 2.7V, V

= 0V, R = 1k, V

= 0.5V

3

7

5

7

µA

EN/RST

IN

SENSE

+

V = 60V, V

= 0V, R = 1k, V

11

13

µA

EN/RST

IN

SENSE

+

EN/RST Pin Current

EN Pin Current

V

= 0V, V = 60V (LT6108-1 Only)

–200

–100

nA

V

EN/RST

+

= 0V, V = 60V (LT6108-2 Only)

EN

+

l

V

IH

V

IL

V

IH

V

IL

EN/RST Pin Input High

EN/RST Pin Input Low

EN Pin Input High

EN Pin Input Low

V = 2.7V to 60V (LT6108-1 Only)

1.9

+

V = 2.7V to 60V (LT6108-1 Only)

0.8

V

+

V = 2.7V to 60V (LT6108-2 Only)

V

+

V = 2.7V to 60V (LT6108-2 Only)

V

Current Sense Ampliﬁer

V

OS

Input Offset Voltage

V

SENSE

V

SENSE

V

SENSE

V

SENSE

= 5mV, LT6108A

= 5mV, LT6108

= 5mV, LT6108A

= 5mV, LT6108

–125

–350

–250

–450

125

350

250

450

µV

l

Input Offset Voltage Drift

V

= 5mV

0.8

60

µV/°C

∆V /∆T

SENSE

+

OS

I

B

Input Bias Current

(SENSELO, SENSEHI)

V = 2.7V to 60V

300

350

nA

l

+

I

Input Offset Current

V = 2.7V to 60V

5

nA

OS

l

Output Current (Note 5)

1

mA

OUTA

+

PSRR

Power Supply Rejection Ratio

(Note 6)

V = 2.7V to 60V

120

114

127

dB

+

CMRR

Common Mode Rejection Ratio

V = 36V, V

= 5mV, V

= 2.7V to 36V

125

dB

SENSE

ICM

+

V = 60V, V

= 27V to 60V

110

103

dB

l

V

Full-Scale Input Sense Voltage

(Note 5)

R

IN

= 500Ω

500

mV

SENSE(MAX)

+

Gain Error (Note 7)

V = 2.7V to 12V

–0.08

%

l

V = 12V to 60V, V

= 5mV to 100mV, MS8 Package

= 5mV to 100mV, DFN Package

–0.2

–0.3

0

SENSE

+

V = 12V to 60V, V

+

l

SENSELO Voltage (Note 8)

V = 2.7V, V

= 100mV, R

= 100mV

= 2k

OUT

2.5

27

V

SENSE

V = 60V, V

+

l

Output Swing High (V to V

)

V = 2.7V, V

= 27mV

0.2

0.5

V

OUTA

SENSE

+

V = 12V, V

= 120mV

BW

Signal Bandwidth

I

= 1mA

1

MHz

kHz

OUT

+

= 100µA

140

t

Input Step Response (to 50% of V = 2.7V, V

Final Output Voltage)

= 24mV Step, Output Rising Edge

SENSE

500

ns

r

+

V = 12V to 60V, V

= 100mV Step, Output Rising Edge

SENSE

Settling Time to 1%

V

= 10mV to 100mV, R = 2k

OUT

2

µs

SETTLE

SENSE

610812fa

4

LT6108-1/LT6108-2

ELECTRICAL CHARACTERISTICS ^Thel denotes the speciﬁcations which apply over the full operating

temperature range, otherwise speciﬁcations are at T_A= 25°C. V⁺= 12V, V_PULLUP= V⁺, V_EN= V_EN/RST= 2.7V, R_IN= 100Ω,

R_OUT= R1 + R2 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless otherwise noted. (See Figure 3)

SYMBOL

Reference and Comparator

Rising Input Threshold Voltage

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

+

l

V

V = 2.7V to 60V, LT6108A

395

392

400

405

408

mV

TH(R)

(Note 9)

V = 2.7V to 60V, LT6108

+

V

= V

– V

TH(F)

V = 2.7V to 60V

3

10

15

mV

nA

HYS

OL

HYS

TH(R)

+

l

Comparator Input Bias Current

Output Low Voltage

V

INC

= 0V, V = 60V

–50

+

V

I

= 500µA, V = 2.7V

60

150

220

mV

OUTC

High to Low Propagation Delay

5mV Overdrive

100mV Overdrive

3

0.5

µs

Output Fall Time

Reset Time

0.08

0.5

µs

t

LT6108-1 Only

RESET

l

Valid RST Pulse Width

2

15

RPW

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 5: The full-scale input sense voltage and the maximum output

current must be considered to achieve the speciﬁed performance.

Note 6: Supply voltage and input common mode voltage are varied while

ampliﬁer input offset voltage is monitored.

Note 2: Input and output pins have ESD diodes connected to ground. The

SENSEHI and SENSELO pins have additional current handling capability

speciﬁed as SENSEHI, SENSELO Input Current.

Note 7: The speciﬁed gain error does not include the effect of external

resistors R and R . Although gain error is only guaranteed between

IN

OUT

+

12V and 60V, similar performance is expected for V < 12V, as well.

Note 3: The LT6108I is guaranteed to meet speciﬁed performance from

–40°C to 85°C. LT6108H is guaranteed to meet speciﬁed performance

from –40°C to 125°C.

Note 4: Supply current is speciﬁed with the comparator output high. When

the comparator output goes low the supply current will increase by 75µA

typically.

Note 8: Refer to SENSELO, SENSEHI Range in the Applications

Information section for more information.

Note 9: The input threshold voltage which causes the output voltage of the

comparator to transition from high to low is speciﬁed. The input voltage

which causes the comparator output to transition from low to high is

the magnitude of the difference between the speciﬁed threshold and the

hysteresis.

610812fa

5

LT6108-1/LT6108-2

Performance characteristics taken at T = 25°C,

TYPICAL PERFORMANCE CHARACTERISTICS

A

V⁺= 12V, V_PULLUP= V⁺, V_EN= V_EN/RST= 2.7V, R_IN= 100Ω, R_OUT= R1 + R2 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless

otherwise noted. (See Figure 3)

Supply Current vs Supply Voltage

Start-Up Supply Current

Enable/Disable Response

600

500

400

300

200

100

0

+

V

5V/DIV

V

EN/RST

2V/DIV

0V

I

S

I

S

500µA/DIV

0µA

610812 G02

0

20

30

40

50

60

10

10µs/DIV

100µs/DIV

SUPPLY VOLTAGE (V)

610812 G03

610812 G01

Input Offset Voltage

vs Temperature

Ampliﬁer Offset Voltage

vs Supply Voltage

Offset Voltage Drift Distribution

300

200

100

0

12

10

8

100

80

5 TYPICAL UNITS

60

40

20

0

6

–20

–40

–60

–80

–100

–200

–300

4

2

0

–2 –1.5 –1 –0.5

0

0.5

1

1.5

2

–40 –25 –10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

610812 G04

0

10

30

40

50

60

20

OFFSET VOLTAGE DRIFT (µV/°C)

SUPPLY VOLTAGE (V)

610812 G38

610812 G05

Ampliﬁer Gain Error

vs Temperature

Ampliﬁer Output Swing

vs Temperature

Ampliﬁer Gain Error Distribution

0.50

0.45

0.40

0.35

0.30

0.25

0.20

0.15

0.10

0.05

0

0.02

0

25

20

V

= 5mV TO 100mV

SENSE

+

–0.02

–0.04

–0.06

–0.08

–0.10

–0.12

–0.14

–016

–0.18

V

= 12V

R

= 1k

IN

V

= 120mV

SENSE

15

R

IN

= 100Ω

10

5

+

V

= 2.7V

V

= 27mV

SENSE

V

= 5mV TO 100mV

SENSE

–25

0

–50

0

25

50

75 100 125

–25

–50

0

25

50

75 100 125

–0.048 –0.052 –0.056 –0.060 –0.064 –0.068

GAIN ERROR (%)

TEMPERATURE (°C)

610812 G18

610812 G06

610812 G07

610812fa

6

LT6108-1/LT6108-2

Performance characteristics taken at T = 25°C,

TYPICAL PERFORMANCE CHARACTERISTICS

A

V⁺= 12V, V_PULLUP= V⁺, V_EN= V_EN/RST= 2.7V, R_IN= 100Ω, R_OUT= R1 + R2 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless

otherwise noted. (See Figure 3)

Common Mode Rejection Ratio

vs Frequency

LT6108-1 Step Response

Ampliﬁer Gain vs Frequency

140

120

100

80

46

40

34

28

22

16

V

SENSE

100mV/DIV

0V

G = 100, R

= 10k

OUT

V

OUTA

1V/DIV

0V

G = 50, R

G = 20, R

= 5k

OUT

V

OUTC

60

2V/DIV

= 2k

OUT

0V

40

V

EN/RST

5V/DIV

20

I

= 1mA

= 100µA

OUTA

R

= 2k

0V

OUT

100mV INC OVERDRIVE

0

1

10 100 1k 10k 100k 1M 10M

1k

10k

100k

FREQUENCY (Hz)

1M

10M

2µs/DIV

FREQUENCY (Hz)

610812 G11

610812 G10

610812 G09

Ampliﬁer Input Bias Current

vs Temperature

Ampliﬁer Step Response

(V_SENSE= 0mV to 100mV)

LT6108-2 Step Response

100

90

80

70

60

50

40

30

20

10

0

R

OUT

= 2k,100mV INC OVERDRIVE

R

= 100Ω

IN

V

SENSE

G = 100V/V

100mV/DIV

0V

V

OUTA

SENSEHI

V

OUTA

2V/DIV

1V/DIV

0V

SENSELO

0V

V

OUTC

V

SENSE

2V/DIV

50mV/DIV

0V

–40 –25 –10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

610812 G13

2µs/DIV

610812 G14

610812 G12

Ampliﬁer Step Response

(V_SENSE= 0mV to 100mV)

Ampliﬁer Step Response

(V_SENSE= 10mV to 100mV)

Ampliﬁer Step Response

(V_SENSE= 10mV to 100mV)

R

= 1k

OUT

R

= 1k

IN

R

= 100Ω

IN

R

= 20k

R

= 20k

G = 100V/V

OUT

G = 20V/V

V

OUTA

V

OUTA

1V/DIV

2V/DIV

0V

V

SENSE

V

100mV/DIV

0V

SENSE

100mV/DIV

0V

50mV/DIV

0V

2µs/DIV

610912 G17

610812 G16

610812 G15

610812fa

7

LT6108-1/LT6108-2

Performance characteristics taken at T = 25°C,

TYPICAL PERFORMANCE CHARACTERISTICS

A

V⁺= 12V, V_PULLUP= V⁺, V_EN= V_EN/RST= 2.7V, R_IN= 100Ω, R_OUT= R1 + R2 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless

otherwise noted. (See Figure 3)

Comparator Threshold

vs Temperature

Comparator Threshold

Distribution

Power Supply Rejection Ratio

vs Frequency

408

406

404

402

400

398

396

394

392

160

140

120

100

80

25

20

5 TYPICAL UNITS

15

10

5

60

40

20

0

–40 –25 –10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

610812 G20

1

10 100 1k 10k 100k 1M 10M

FREQUENCY (Hz)

396

397.6 399.2 400.8 402.8

COMPARATOR THRESHOLD (mV)

404

610812 G08

610812 G19

Hysteresis Distribution

Hysteresis vs Temperature

Hysteresis vs Supply Voltage

20

18

16

14

12

10

8

30

25

20

15

10

5

14

12

10

8

5 TYPICAL UNITS

–40°C

25°C

125°C

6

4

2

0

3

7.7 9.3 10.9 12.5 14.1 15.7 17.3

COMPARATOR HYSTERESIS (mV)

–40 –25 –10

5

20 35 50 65 80 95 110 125

TEMPERATURE (°C)

610812 G22

40

60

4.6 6.2

0

10

20

30

50

+

V

(V)

610812 G21

610812 G23

LT6108-1 EN/RST Current vs

Voltage

LT6108-2 EN Current

vs Voltage

50

0

50

0

–50

–100

–150

–200

–250

–100

–150

–200

–250

0

20

30

40

50

60

0

20

30

40

50

60

10

EN/RST VOLTAGE (V)

EN VOLTAGE (V)

610812 G24

610812 G25

610812fa

8

LT6108-1/LT6108-2

Performance characteristics taken at T = 25°C,

TYPICAL PERFORMANCE CHARACTERISTICS

A

V⁺= 12V, V_PULLUP= V⁺, V_EN= V_EN/RST= 2.7V, R_IN= 100Ω, R_OUT= R1 + R2 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless

otherwise noted. (See Figure 3)

Comparator Input Bias Current

vs Input Voltage

Comparator Output Low Voltage

vs Output Sink Current

Comparator Input Bias Current

vs Input Voltage

10

5

10

5

1.00

0.75

0.50

0.25

0

125°C

25°C

–40°C

0

–5

–10

–15

–20

–10

–15

–20

125°C

25°C

–40°C

0

0.2

0.4

0.6

0.8

1.0

0

20

40

60

0

1

2

3

COMPARATOR INPUT VOLTAGE (V)

I

(mA)

OUTC

610812 G28

610812 G27

610812 G29

Comparator Propagation Delay

vs Input Overdrive

Comparator Rise/Fall Time

vs Pull-Up Resistor

Comparator Output Leakage

Current vs Pull-Up Voltage

23

18

13

8

10000

1000

100

5.0

4.5

4.0

3.5

3.0

2.5

2.0

1.5

1.0

0.5

0

V

= 0.9 • V

= 0.1 • V

OH

OL

PULLUP

100mV INC OVERDRIVE

C

= 2pF

L

125°C

RISING INPUT

LT6108-1 AND LT6108-2

FALLING

INPUT

LT6108-2

FALLING INPUT

LT6108-2

RISING INPUT

LT6108-1 AND

LT6108-2

3

–40°C AND 25°C

10

–2

0

20

30

40

50

60

1

10

100

1000

10

0

40

80

120

160

200

R

PULL-UP RESISTOR (kΩ)

COMPARATOR OUTPUT PULL-UP VOLTAGE (V)

COMPARATOR INPUT OVERDRIVE (mV)

C

610812 G32

610812 G30

610812 G31

LT6108-1 Comparator Step

Response (100mV INC Overdrive)

LT6108-1 Comparator Step

Response (5mV INC Overdrive)

+

V

= 5V

V

= 5V

V

INC

V

INC

0.5V/DIV

0V

0.5V/DIV

0V

V

OUTC

V

OUTC

2V/DIV

0V

V

EN/RST

5V/DIV

0V

5µs/DIV

610812 G34

610812 G33

610812fa

9

LT6108-1/LT6108-2

Performance characteristics taken at T = 25°C,

TYPICAL PERFORMANCE CHARACTERISTICS

A

V⁺= 12V, V_PULLUP= V⁺, V_EN= V_EN/RST= 2.7V, R_IN= 100Ω, R_OUT= R1 + R2 = 10k, gain = 100, R_C= 25.5k, C_L= C_LC= 2pF, unless

otherwise noted. (See Figure 3)

LT6108-2 Comparator Step

Response (5mV INC Overdrive)

LT6108-2 Comparator Step

Response (100mV INC Overdrive)

LT6108-1 Comparator Reset

Response

+

V

= 5V

V

= 5V

V

INC

0.5V/DIV

0V

INC

0.5V/DIV

0V

V

OUTC

5V/DIV

0V

V

OUTC

1V/DIV

OUTC

1V/DIV

V

EN/RST

2V/DIV

0V

5µs/DIV

610812 G35

610812 G36

610812 G37

PIN FUNCTIONS

SENSELO (Pin 1): Sense Ampliﬁer Input. This pin must

be tied to the load end of the sense resistor.

OUTA (Pin 6): Current Output of the Sense Ampliﬁer. This

pin will source a current that is equal to the sense voltage

divided by the external gain setting resistor, R .

IN

EN/RST (Pin 2, LT6108-1 Only): Enable and Latch Reset

Input. When the EN/RST pin is pulled high the LT6108-1

is enabled. When the EN/RST pin is pulled low for longer

than typically 40µs, the LT6108-1 will enter the shutdown

mode. Pulsing this pin low for between 2µs and 15µs will

reset the comparator of the LT6108-1.

+

V (Pin 7): Positive Supply Pin. The V pin can be con-

nected directly to either side of the sense resistor, R

.

SENSE

+

When V is tied to the load end of the sense resistor, the

+

SENSEHI pin can go up to 0.2V above V . Supply current

is drawn through this pin.

EN (Pin 2, LT6108-2 Only): Enable Input. When the en-

able pin is pulled high the LT6108-2 is enabled. When the

enable pin is pulled low for longer than typically 40µs, the

LT6108-2 will enter the shutdown mode

SENSEHI (Pin 8): Sense Ampliﬁer Input. The internal

sense ampliﬁer will drive SENSEHI to the same potential

as SENSELO. A resistor (typically R ) tied from supply

IN

to SENSEHI sets the output current, I

= V

/R ,

SENSE

OUT

SENSE IN

where V

is the voltage developed across R

.

SENSE

OUTC (Pin 3): Open-Drain Comparator Output. Off-state

–

voltage may be as high as 60V above V , regardless of

ExposedPad(Pin9,DCBPackageOnly):V .Theexposed

+

–

V used.

pad may be left open or connected to device V . Connect-

–

ing the exposed pad to a V plane will improve thermal

–

V (Pin 4): Negative Supply Pin. This pin is normally con-

management in high voltage applications. The exposed

nected to ground.

–

pad should not be used as the primary connection for V .

INC (Pin 5): This is the inverting input of the comparator.

The other comparator input is internally connected to the

400mV reference.

610812fa

10

LT6108-1/LT6108-2

BLOCK DIAGRAMS

7

+

LT6108-1

V

100Ω

34V

6V

3k

SENSEHI

8

–

+

SENSELO

1

OUTA

6

–

V

–

V

+

–

V

200nA

EN/RST

ENABLE AND

RESET TIMING

2

RESET

+

V

–

+

INC

5

OVERCURRENT FLAG

OUTC

3

400mV

REFERENCE

–

V

–

V

4

610812 F01

Figure 1. LT6108-1 Block Diagram (Latching Comparator)

7

+

LT6108-2

V

100Ω

34V

6V

–

3k

SENSEHI

8

–

+

SENSELO

1

OUTA

6

5

–

V

+

–

V

100nA

EN

2

+

V

–

+

INC

OVERCURRENT FLAG

OUTC

3

400mV

REFERENCE

–

V

4

610812 F02

Figure 2. LT6108-2 Block Diagram (Non-Latching Comparator)

610812fa

11

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

Note that V

can be exceeded without damag-

The LT6108 high side current sense ampliﬁer provides

accuratemonitoringofcurrentsthroughanexternalsense

resistor. The input sense voltage is level-shifted from the

sensed power supply to a ground referenced output and

is ampliﬁed by a user-selected gain to the output. The

output voltage is directly proportional to the current ﬂow-

ing through the sense resistor.

SENSE(MAX)

ing the ampliﬁer, however, output accuracy will degrade

as V exceeds V , resulting in increased

SENSE

SENSE(MAX)

output current, I

.

OUTA

Selection of External Current Sense Resistor

Theexternalsenseresistor,R ,hasasigniﬁcanteffect

on the function of a current sensing system and must be

chosen with care.

SENSE

The LT6108 comparator has a threshold set with a built-in

400mV precision reference and has 10mV of hysteresis.

The open-drain output can be easily used to level shift to

digital supplies.

First, the power dissipation in the resistor should be

considered. The measured load current will cause power

dissipation as well as a voltage drop in R

result, the sense resistor should be as small as possible

while still providing the input dynamic range required by

the measurement. Note that the input dynamic range is

the difference between the maximum input signal and the

minimum accurately reproduced signal, and is limited

primarily by input DC offset of the internal sense ampli-

ﬁer of the LT6108. To ensure the speciﬁed performance,

. As a

SENSE

Ampliﬁer Theory of Operation

An internal sense ampliﬁer loop forces SENSEHI to have

the same potential as SENSELO as shown in Figure 3.

Connecting an external resistor, R , between SENSEHI

IN

and V

forces a potential, V

, across R . A

SUPPLY

corresponding current, I

SENSE IN

, equal to V

/R , will

SENSE IN

OUTA

ﬂow through R . The high impedance inputs of the sense

IN

ampliﬁer do not load this current, so it will ﬂow through

R

should be small enough that V

SENSE(MAX)

does not

SENSE

exceed V

SENSE

an internal MOSFET to the output pin, OUTA.

under peak load conditions. As an

example, an application may require the maximum sense

voltage be 100mV. If this application is expected to draw

The output current can be transformed back into a voltage

–

by adding a resistor from OUTA to V (typically ground).

2A at peak load, R

should be set to 50mΩ.

SENSE

The output voltage is then:

Once the maximum R

value is determined, the mini-

–

SENSE

V

= V + I

• R

OUT

OUTA OUT

mum sense resistor value will be set by the resolution or

dynamic range required. The minimum signal that can be

accuratelyrepresentedbythissenseampliﬁerislimitedby

theinputoffset.Asanexample,theLT6108hasamaximum

input offset of 125µV. If the minimum current is 20mA, a

where R

= R1 + R2 as shown in Figure 3.

OUT

Table 1. Example Gain Conﬁgurations

GAIN

20

R

IN

R

V

FOR V

= 5V

I

AT V

= 5V

OUT

SENSE

OUT

OUTA

499Ω

200Ω

100Ω

10k

250mV

100mV

50mV

500µA

sense resistor of 6.25mΩ will set V

to 125µV. This is

SENSE

50

the same value as the input offset. A larger sense resistor

100

will reduce the error due to offset by increasing the sense

voltage for a given load current. Choosing a 50mΩ R

SENSE

Useful Equations

Input Voltage: V_SENSE= I_SENSE•R_SENSE

will maximize the dynamic range and provide a system

that has 100mV across the sense resistor at peak load

(2A), while input offset causes an error equivalent to only

2.5mA of load current.

V_OUT

V_SENSER_IN

R_OUT

Voltage Gain:

Current Gain:

=

In the previous example, the peak dissipation in R

I_OUTAR_SENSE

I_SENSE

SENSE

=

is 200mW. If a 5mΩ sense resistor is employed, then

the effective current error is 25mA, while the peak sense

voltage is reduced to 10mV at 2A, dissipating only 20mW.

R_IN

610812fa

12

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

The low offset and corresponding large dynamic range of

theLT6108makeitmoreﬂexiblethanothersolutionsinthis

respect.The125µVmaximumoffsetgives72dBofdynamic

range for a sense voltage that is limited to 500mV max.

Selection of External Input Gain Resistor, R

IN

R

should be chosen to allow the required speed and

IN

resolution while limiting the output current to 1mA. The

maximum value for R is 1k to maintain good loop sta-

IN

SENSE

bility. For a given V

, larger values of R will lower

IN

Sense Resistor Connection

power dissipation in the LT6108 due to the reduction

Kelvin connection of the SENSEHI and SENSELO inputs

to the sense resistor should be used in all but the lowest

power applications. Solder connections and PC board

interconnections that carry high currents can cause sig-

niﬁcant error in measurement due to their relatively large

resistances.One10mm× 10mmsquaretraceof1ozcopper

is approximately 0.5mΩ. A 1mV error can be caused by as

little as 2A ﬂowing through this small interconnect. This

in I

while smaller values of R will result in faster

OUT

IN

response time due to the increase in I . If low sense

OUT

currents must be resolved accurately in a system that has

a very wide dynamic range, a smaller R may be used

IN

if the maximum I

such as with a Schottky diode across R

current is limited in another way,

OUTA

(Figure 4).

SENSE

This will reduce the high current measurement accuracy

by limiting the result, while increasing the low current

measurement resolution.

will cause a 1% error for a full-scale V

of 100mV.

SENSE

A 10A load current in the same interconnect will cause

a 5% error for the same 100mV signal. By isolating the

sense traces from the high current paths, this error can

be reduced by orders of magnitude. A sense resistor with

integratedKelvinsenseterminalswillgivethebestresults.

Figure3illustratestherecommendedmethodforconnect-

ing the SENSEHI and SENSELO pins to the sense resistor.

+

V

R

D

SENSE

610812 F04

LOAD

Figure 4. Shunt Diode Limits Maximum Input Voltage to Allow

Better Low Input Resolution Without Overranging

V

SUPPLY

+

R

IN

R

SENSE

V

SENSE

LT6108-1

–

SENSEHI

8

7

1

SENSELO

LOAD

+

–

V

SENSE

R

SENSE

C1

I

=

–

+

SENSE

V

+

V

2

3

EN/RST

V

RESET

OUTA

INC

6

5

V

OUT

V

+

PULLUP

V

I

OUTA

R2*

R1*

C

L

–

+

R

C

OUTC

OVERCURRENT

FLAG

400mV

C

LC

REFERENCE

–

V

610812 F03

4

*R

OUT

= R1 + R2

Figure 3. LT6108-1 Typical Connection

610812fa

13

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

This approach can be helpful in cases where occasional

bursts of high currents can be ignored.

Ampliﬁer Error Sources

The current sense system uses an ampliﬁer and resistors

to apply gain and level-shift the result. Consequently, the

output is dependent on the characteristics of the ampliﬁer,

suchasgainerrorandinputoffset, aswellasthematching

of the external resistors.

Care should be taken when designing the board layout for

R , especially for small R values. All trace and inter-

IN

connect resistances will increase the effective R value,

IN

causing a gain error.

The power dissipated in the sense resistor can create a

thermal gradient across a printed circuit board and con-

Ideally, the circuit output is:

R

R_IN

^OUT; V_SENSE= R_SENSE•I_SENSE

sequently a gain error if R and R

are placed such

V_OUT= V_SENSE

•

IN

OUT

that they operate at different temperatures. If signiﬁcant

power is being dissipated in the sense resistor then care

In this case, the only error is due to external resistor

mismatch, which provides an error in gain only. However,

offset voltage, input bias current and ﬁnite gain in the

ampliﬁer can cause additional errors:

should be taken to place R and R

such that the gain

OUT

IN

error due to the thermal gradient is minimized.

Selection of External Output Gain Resistor, R

OUT

Output Voltage Error, ∆V

, Due to the Ampliﬁer

OUT(VOS)

The output resistor, R , determines how the output cur-

OUT

DC Offset Voltage, V

OS

rent is converted to voltage. V

is simply I

• R

.

OUT

OUTA

OUT

Typically, R

is a combination of resistors conﬁgured

OUT

R_OUT

R_IN

∆V_OUT(VOS)= V_OS•

as a resistor divider which has a voltage tap going to the

comparator input to set the comparator threshold.

The DC offset voltage of the ampliﬁer adds directly to the

In choosing an output resistor, the maximum output volt-

age must ﬁrst be considered. If the subsequent circuit is a

valueofthesensevoltage, V . AsV isincreased,

SENSE

accuracyimproves.Thisisthedominanterrorofthesystem

and it limits the available dynamic range.

buffer or ADC with limited input range, then R

must be

OUT

chosen so that I

• R

is less than the allowed

OUTA(MAX)

maximum input range of this circuit.

OUT

Output Voltage Error, ∆V

, Due to the Bias

OUT(IBIAS)

+

–

In addition, the output impedance is determined by R

.

Currents I and I

OUT

B

If another circuit is being driven, then the input impedance

ofthatcircuitmustbeconsidered.Ifthesubsequentcircuit

has high enough input impedance, then almost any use-

ful output impedance will be acceptable. However, if the

subsequent circuit has relatively low input impedance, or

draws spikes of current such as an ADC load, then a lower

outputimpedancemayberequiredtopreservetheaccuracy

oftheoutput. MoreinformationcanbefoundintheOutput

Filtering section. As an example, if the input impedance of

+

The ampliﬁer bias current I ﬂows into the SENSELO pin

B

–

while I ﬂows into the SENSEHI pin. The error due to I

B

is the following:





R_SENSE

R_IN

∆V_OUT(IBIAS)= R_OUTI_B⁺•

–I_B

–









+

–

Since I ≈ I = I

, if R

<< R then,

B

BIAS

SENSE IN

∆V

= –R

(I

)

OUT(IBIAS)

OUT BIAS

the driven circuit, R

, is 100 times R , then the

IN(DRIVEN)

OUT

It is useful to refer the error to the input:

∆V = –R (I

accuracy of V

will be reduced by 1% since:

R_OUT•R_IN(DRIVEN)

R_OUT+ R_IN(DRIVEN)

100

OUT

)

IN BIAS

VIN(IBIAS)

V_OUT= I_OUTA

•

For instance, if I

is 100nA and R is 1k, the input re-

IN

BIAS

ferred error is 100µV. This error becomes less signiﬁcant

= I_OUTA•R_OUT

•

= 0.99•I_OUTA•R_OUT

as the value of R decreases. The bias current error can

IN

101

610812fa

14

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

+

be reduced if an external resistor, R , is connected as

Output Current Limitations Due to Power Dissipation

IN

shown in Figure 5, the error is then reduced to:

The LT6108 can deliver a continuous current of 1mA to the

+

–

V

= R

• I ; I = I – I

OUTA pin. This current ﬂows through R and enters the

OUT(IBIAS)

OUT OS OS

B

IN

current sense ampliﬁer via the SENSEHI pin. The power

Minimizing low current errors will maximize the dynamic

range of the circuit.

dissipated in the LT6108 due to the output signal is:

P

= (V

– V

) • I

OUT

SENSEHI

OUTA

+

V

+

7

+

Since V

≈ V , P

≈ (V – V

) • I

OUTA OUTA

SENSEHI

OUTA

V

LT6108

V

BATT

There is also power dissipated due to the quiescent power

supply current:

R

IN

–

+

8

1

SENSEHI

+

R

SENSE

P = I • V

S

OUTA

6

SENSELO

V

OUT

+

The comparator output current ﬂows into the comparator

R

IN

R

OUT

–

V

4

–

I

SENSE

output pin and out of the V pin. The power dissipated in

610812 F05

the LT6108 due to the comparator is often insigniﬁcant

and can be calculated as follows:

Figure 5. R_IN⁺Reduces Error Due to I_B

–

P

OUTC

= (V

– V ) • I

OUTC

Output Voltage Error, ∆V

, Due to

OUT(GAIN ERROR)

The total power dissipated is the sum of these

dissipations:

External Resistors

The LT6108 exhibits a very low gain error. As a result,

the gain error is only signiﬁcant when low tolerance

resistors are used to set the gain. Note the gain error is

systematically negative. For instance, if 0.1% resistors

P

TOTAL

= P

+ P + P

OUTC S

OUTA

At maximum supply and maximum output currents, the

totalpowerdissipationcanexceed150mW.Thiswillcause

signiﬁcant heating of the LT6108 die. In order to prevent

damage to the LT6108, the maximum expected dissipa-

tion in each application should be calculated. This number

are used for R and R

then the resulting worst-case

IN

OUT

gain error is –0.4% with R = 100Ω. Figure 6 is a graph

IN

of the maximum gain error which can be expected versus

the external resistor tolerance.

can be multiplied by the θ value, 163°C/W for the MS8

JA

package or 64°C/W for the DFN, to ﬁnd the maximum

expecteddietemperature.Properheatsinkingandthermal

relief should be used to ensure that the die temperature

does not exceed the maximum rating.

10

1

R

IN

= 100Ω

Output Filtering

R

= 1k

IN

The AC output voltage, V , is simply I

• Z . This

OUT

OUTA

0.1

makes ﬁltering straightforward. Any circuit may be used

which generates the required Z to get the desired ﬁlter

OUT

response. For example, a capacitor in parallel with R

OUT

0.01

0.1

1

10

will give a lowpass response. This will reduce noise at the

output, and may also be useful as a charge reservoir to

keep the output steady while driving a switching circuit

RESISTOR TOLERANCE (%)

610812 F06

Figure 6. Gain Error vs Resistor Tolerance

610812fa

15

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

such as a MUX or ADC. This output capacitor in parallel

60

50

40

with R

will create an output pole at:

OUT

1

f_–3dB

=

2•π •R_OUT•C_L

40.2V

SENSELO, SENSEHI Range

The difference between V

VALID SENSELO/

SENSEHI RANGE

+

30

27

(see Figure 7) and V , as

SENSE

BATT

well as the maximum value of V

, must be considered

20.2V

20

to ensure that the SENSELO pin doesn’t exceed the range

listed in the Electrical Characteristics table. The SENSELO

and SENSEHI pins of the LT6108 can function from 0.2V

above the positive supply to 33V below it. These operat-

ing voltages are limited by internal diode clamps shown

in Figures 1 and 2. On supplies less than 35.5V, the lower

10

2.8

2.5

2.7

10

20

30 35.5 40

V (V)

50

60

610812 F08

+

–

range is limited by V + 2.5V. This allows the monitored

Figure 8. Allowable SENSELO, SENSEHI Voltage Range

supply, V

, to be separate from the LT6108 positive

BATT

supply as shown in Figure 7. Figure 8 shows the range of

operating voltages for the SENSELO and SENSEHI inputs,

7

+

for different supply voltage inputs (V ). The SENSELO and

V

LT6108

V

BATT

SENSEHI range has been designed to allow the LT6108 to

monitor its own supply current (in addition to the load),

R

IN

–

+

8

1

SENSEHI

as long as V

Figure 9.

is less than 200mV. This is shown in

R

SENSE

6

OUTA

SENSELO

V

OUT

R

OUT

–

V

4

Minimum Output Voltage

I

SENSE

610812 F09

The output of the LT6108 current sense ampliﬁer can

produceanon-zerooutputvoltagewhenthesensevoltage

Figure 9. LT6108 Supply Current Monitored with Load

is zero. This is a result of the sense ampliﬁer V being

OS

forced across R as discussed in the Output Voltage Er-

IN

ror, ∆V

section. Figure 10 shows the effect of the

OUT(VOS)

120

G = 100

input offset voltage on the transfer function for parts at

100

80

the V limits. With a negative offset voltage, zero input

OS

+

V

= –125µV

OS

V

60

7

+

V

LT6108

V

BATT

40

20

0

V

= 125µV

OS

R

IN

–

+

8

1

SENSEHI

R

SENSE

6

OUTA

SENSELO

V

OUT

0

100 200 300 400 500 600 700 800 900 1000

INPUT SENSE VOLTAGE (µV)

R

OUT

I

–

SENSE

V

4

610812 F10

610812 F07

Figure 10. Ampliﬁer Output Voltage vs Input Sense Voltage

Figure 7. V⁺Powered Separately from Load Supply (V_BATT

)

610812fa

16

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

sense voltage produces an output voltage. With a positive

offset voltage, the output voltage is zero until the input

sense voltage exceeds the input offset voltage. Neglect-

overdrive on the comparator input being determined by

the speed of the ampliﬁer output.

Internal Reference and Comparator

ing V , the output circuit is not limited by saturation of

OS

pull-down circuitry and can reach 0V.

The integrated precision reference and comparator com-

bined with the high precision current sense allow for rapid

andeasydetectionofabnormalloadcurrents. Thisisoften

critical in systems that require high levels of safety and

reliability. The LT6108-1 comparator is optimized for fault

detectionandisdesignedwithalatchingoutput.Thelatch-

ing output prevents faults from clearing themselves and

requires a separate system or user to reset the output. In

applications where the comparator output can intervene

and disconnect loads from the supply, a latched output

is required to avoid oscillation. The latching output is

also useful for detecting problems that are intermittent.

The comparator output on the LT6108-2 is non-latching

and can be used in applications where a latching output

is not desired.

Response Time

The LT6108 ampliﬁer is designed to exhibit fast response

to inputs for the purpose of circuit protection or current

monitoring. This response time will be affected by the

external components in two ways, delay and speed.

If the output current is very low and an input transient

occurs, there may be an increased delay before the

outputvoltagebeginstochange. TheTypicalPerformance

Characteristics show that this delay is short and it can

be improved by increasing the minimum output current,

either by increasing R

or decreasing R . Note that

SENSE

IN

the Typical Performance Characteristics are labeled with

respect to the initial sense voltage.

The comparator has one input available externally. The

other comparator input is connected internally to the

400mV precision reference. The input threshold (the

voltage which causes the output to transition from high

to low) is designed to be equal to that of the reference.

The reference voltage is established with respect to the

The speed is also affected by the external components.

Using a larger R

will decrease the response time, since

OUT

V

= I

OUT

• Z

where Z

is the parallel combination

OUT OUTA OUT

OUT

of R

and any parasitic and/or load capacitance. Note

that reducing R or increasing R

will both have the

IN

OUT

effect of increasing the voltage gain of the circuit. If the

–

device V connection.

output capacitance is limiting the speed of the system, R

IN

and R

can be decreased together in order to maintain

OUT

Comparator Input

the desired gain and provide more current to charge the

output capacitance.

–

ThecomparatorinputcanswingfromV to60Vregardless

of the supply voltage used. The input current for inputs

well above the threshold is just a few pAs. With decreas-

ing input voltage, a small bias current begins to be drawn

out of the input near the threshold, reaching 50nA max

when at ground potential. Note that this change in input

bias current can cause a small nonlinearity in the OUTA

transfer function if the comparator input is coupled to

the ampliﬁer output with a voltage divider. For example,

if the maximum comparator input current is 50nA, and

the resistance seen looking out of the comparator input is

1k, then a change in output voltage of 50µV will be seen

on the analog output when the comparator input voltage

passes through its threshold.

The response time of the comparator is the sum of the

propagation delay and the fall time. The propagation delay

is a function of the overdrive voltage on the input of the

comparator.Alargeroverdrivewillresultinalowerpropaga-

tion delay. This helps achieve a fast system response time

to fault events. The fall time is affected by the load on the

output of the comparator as well as the pull-up voltage.

The LT6108 ampliﬁer has a typical response time of 500ns

andthecomparatorshaveatypicalresponsetimeof500ns.

When conﬁgured as a system, the ampliﬁer output drives

the comparator input causing a total system response

time which is typically greater than that implied by the

individually speciﬁed response times. This is due to the

610812fa

17

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

Setting Comparator Threshold

As shown in Figure 12, R2 can be used to increase the

gain from V

to V

without changing V

.

SENSE

OUT

SENSE(TRIP)

Thecomparatorhasaninternal400mVprecisionreference.

In order to set the trip point of the LT6108 comparator as

conﬁgured in Figure 11, the input sense voltage at which

As before, R1 can be easily calculated:

400mV

R1= R_IN

the comparator will trip, V

must be calculated:

SENSE(TRIP)

V_SENSE(TRIP)

V

= I

• R

SENSE(TRIP) SENSE

SENSE(TRIP)

The gain is now:

The selection of R is discussed in the Selection of Exter-

nal Input Gain Resistor R section. Once R is selected,

OUT

IN

R1+ R2

A_V=

IN

R

can be calculated:

R_IN

400mV

V_SENSE(TRIP)

This gain equation can be easily solved for R2:

R2 = A • R – R1

R_OUT= R_IN

V

IN

Since the ampliﬁer output is connected directly to the

IftheconﬁgurationofFigure11givestoomuchgain,R2can

comparator input, the gain from V

to V

is:

be used to reduce the gain without changing V

SENSE

OUT

SENSE(TRIP)

as shown in Figure 13. A can be easily calculated:

V

400mV

A_V=

R1

A_V=

V_SENSE(TRIP)

R_IN

V

SUPPLY

+

R

IN

R

SENSE

V

SENSE

LT6108-1

–

SENSEHI

8

7

1

SENSELO

LOAD

+

–

V

SENSE

R

SENSE

C1

I

=

–

+

SENSE

V

+

V

2

3

EN/RST

V

RESET

OUTA

INC

6

5

V

OUT

V

+

PULLUP

V

I

OUTA

C

L

–

+

R

C

OUTC

OVERCURRENT

FLAG

R

OUT

400mV

C

LC

REFERENCE

–

V

610812 F11

4

Figure 11. Basic Comparator Conﬁguration

610812fa

18

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

V

SUPPLY

+

R

IN

R

SENSE

V

SENSE

LT6108-1

–

SENSEHI

8

1

SENSELO

LOAD

+

–

V

SENSE

R

SENSE

C1

I

=

–

+

SENSE

V

7

+

V

2

3

EN/RST

V

RESET

OUTA

INC

6

V

OUT

V

+

PULLUP

V

I

OUTA

R2

R1

C

L

5

–

+

R

C

OUTC

OVERCURRENT

FLAG

400mV

C

LC

REFERENCE

–

V

610812 F12

4

Figure 12: Comparator Conﬁguration with Increased A_V

V

SUPPLY

+

R

IN

R

SENSE

V

SENSE

LT6108-1

–

SENSEHI

8

7

1

SENSELO

LOAD

+

–

V

SENSE

R

SENSE

C1

I

=

–

+

SENSE

V

+

V

2

3

EN/RST

V

RESET

OUTA

INC

6

5

+

V

I

V

C

L

OUTA

PULLUP

–

+

R

C

OUTC

OVERCURRENT

FLAG

R2

400mV

REFERENCE

C

LC

V

OUT

R1

–

610812 F13

V

4

Figure 13: Comparator Conﬁguration with Reduced A_V

610812fa

19

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

This gain equation can be easily solved for R1:

circuitry will have an effect on both the rising and fall-

ing input thresholds, V (the actual internal threshold

TH

R1 = A • R

V

IN

remains unaffected).

The value of R2 can be calculated:

Figure 15 shows how to add additional hysteresis to the

comparator.

400mV •R_IN– V_SENSE(TRIP)•R1

R2=

V_SENSE(TRIP)

R5canbecalculatedfromtheampliﬁeroutputcurrentwhich

is required to cause the comparator output to trip, I

.

OVER

Hysteresis

400mV

I_OVER

R5=

, Assuming R1+ R2 >> R5

(

)

The comparator has a typical built-in hysteresis of 10mV

to simplify design, ensure stable operation in the pres-

ence of noise at the input, and to reject supply noise that

might be induced by state change load transients. The

hysteresis is designed such that the threshold voltage is

altered when the output is transitioning from low to high

as is shown in Figure 14.

To ensure (R1 + R2) >> R5, R1 should be chosen such

that R1 >> R5 so that V does not change signiﬁcantly

OUTA

when the comparator trips.

R3 should be chosen to allow sufﬁcient V and compara-

tor output rise time due to capacitive loading.

OL

R2 can be calculated:

External positive feedback circuitry can be employed

to increase the effective hysteresis if desired, but such





V_DD– 390mV

V_HYS(EXTRA)

R2 = R1•





INCREASING

OUTC

V

INC





Note that the hysteresis being added, V

, is in

HYS(EXTRA)

610812 F14

V

additiontothetypical10mVofbuilt-inhysteresis. Forvery

large values of R2 PCB related leakage may become an

issue. A tee network can be implemented to reduce the

required resistor values.

HYS

V

TH

Figure 14. Comparator Output Transfer Characteristics

+

V

7

+

V

LT6108-1

+

–

V

R

IN

–

+

8

1

SENSEHI

SENSELO

R

SENSE

OUTA

INC

6

5

I

LOAD

+

V

+

R4

R5

V

R1

–

+

VTH

R3

3

OUTC

400mV

REFERENCE

–

V

4

V

DD

R2

610812 F15

Figure 15. Inverting Comparator with Added Hysteresis

610812fa

20

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

The approximate total hysteresis is:

EN/RST Pin (LT6108-1 Only)

The EN/RST pin performs the two functions of resetting

the latch on the comparator as well as shutting down the

LT6108-1. When this pin is pulled high the LT6108-1 is

enabled. After powering on the LT6108-1, the comparator

mustberesetinordertoguaranteeavalidstateatitsoutput.

V –390mV





DD

V_HYS= 10mV + R1•









R2

For example, to achieve I

= 900µA with 50mV of total

OVER

hysteresis, R5 = 442Ω. Choosing R1 = 4.42k, R3 = 10k

and V = 5V results in R2 = 513k.

DD

Applying a pulse to the EN/RST pin will reset the compara-

tor from its tripped low state as long as the input on the

comparator is below the threshold and hysteresis. For

The analog output voltage will also be affected when the

comparator trips due to the current injected into R5 by

the positive feedback. Because of this, it is desirable to

have (R1 + R2) >> R5. The maximum V

by this can be calculated as:

example, if V is pulled higher than 400mV and latches

INC

thecomparator,aresetpulsewillnotresetthatcomparator

unless its input is held below the threshold by a voltage

greater than the 10mV typical hysteresis. The comparator

output typically unlatches in 0.5µs with 2pF of capacitive

load. Increased capacitive loading on the comparator

output will cause an increased unlatch time.

error caused

OUTA

R5





∆V_OUTA= V_DD

•









R1+ R2+ R5

In the previous example, this is an error of 4.3mV at the

output of the ampliﬁer or 43µV at the input of the ampliﬁer

assuming a gain of 100.

Figure 16 shows the reset functionality of the EN/RST

pin. The width of the pulse applied to reset the compara-

tor must be greater than t

RPW(MAX)

(2µs) but less than

RPW(MIN)

When using the comparator with its input decoupled from

the output of the ampliﬁer it may be driven directly by a

voltage source. It is useful to know the threshold voltage

equationswithadditionalhysteresis. Theinputrisingedge

threshold which causes the output to transition from high

to low is:

t

(15µs). Applying a pulse that is longer than

40µs typically (or tying the pin low) will cause the part

to enter shutdown. Once the part has entered shutdown,

the supply current will be reduced to 3µA typically and the

ampliﬁer, comparator and reference will cease to function

until the EN/RST pin is transitioned high. When the part

is disabled, both the ampliﬁer and comparator outputs

are high impedance.

R1

R2





V_{TH R}= 400mV • 1+







( )



RESET PULSE WIDTH LIMITS

COMPARATOR

The input falling edge threshold which causes the output

to transition from low to high is:

EN/RST

RESET

t

RPW(MIN)

2µs

R1

R2

R1

R2











t

RPW(MAX)

15µs

V_{TH F}= 390mV 1+

– V











^DD

( )



610812 F16

OUTC

Comparator Output

t

RESET

0.5µs (TYPICAL)

The comparator output can maintain a logic-low level of

150mV while sinking 500µA. The output can sink higher

Figure 16. Comparator Reset Functionality

currents at elevated V levels as shown in the Typical

OL

PerformanceCharacteristics.Loadcurrentsareconducted

When the EN/RST pin is transitioned from low to high

to enable the part, the ampliﬁer output PMOS can turn

on momentarily causing typically 1mA of current to ﬂow

into the SENSEHI pin and out of the OUTA pin. Once

–

to the V pin. The output off-state voltage may range

–

between 0V and 60V with respect to V , regardless of the

supply voltage used.

the ampliﬁer is fully on, the output will go to the correct

610812fa

21

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

current.Figure17showsthisbehaviorandtheimpactithas

Power Up

on V

. Circuitry connected to OUTA can be protected

OUTA

After powering on the LT6108-1, the comparator must

be reset in order to guarantee a valid state at its output.

Fast supply ramps may cause a supply current transient

during start-up as shown in the Typical Performance

Characteristics. This current can be lowered by reducing

the edge speed of the supply.

from these transients by using an external diode to clamp

, or a capacitor to ﬁlter V

V

.

OUTA

+

V

= 60V

R

= 100Ω

OUT

IN

= 10k

V

EN/RST

2V/DIV

Reverse-Supply Protection

0V

The LT6108 is not protected internally from external rever-

sal of supply polarity. To prevent damage that may occur

during this condition, a Schottky diode should be added

V

OUTA

2V/DIV

–

in series with V (Figure 18). This will limit the reverse

0V

current through the LT6108. Note that this diode will limit

the low voltage operation of the LT6108 by effectively

50µs/DIV

610812 F17

reducing the supply voltage to the part by V .

D

Figure 17. Ampliﬁer Enable Response

Also note that the comparator reference, comparator

–

output and EN/RST input are referenced to the V pin. In

EN Pin (LT6108-2)

order to preserve the precision of the reference and to

–

Whenthispinispulledhigh,theLT6108-2isenabled.When

the enable pin is pulled low for longer than 40µs typically,

the LT6108-2 will enter the shutdown mode.

avoid driving the comparator inputs below V , R2 must

–

connect to the V pin. This will shift the ampliﬁer output

voltage up by V . V

can be accurately measured

D

OUTA

+

V

7

+

V

LT6108-1

+

–

V

R

IN

–

+

8

1

SENSEHI

SENSELO

R

SENSE

OUTA

INC

6

5

V

DD

I

+

LOAD

+

V

R1

DD

R3

–

+

3

2

OUTC

V

OUTA

R2

–

400mV

REFERENCE

V

DD

EN/RST

–

V

4

610812 F18

+

V

D

–

Figure 18. Schottky Prevents Damage During Supply Reversal

610812fa

22

LT6108-1/LT6108-2

APPLICATIONS INFORMATION

differentially across R1 and R2. The comparator output

valid input levels to the LT6108 and avoid driving EN/RST

–

low voltage will also be shifted up by V . The EN/RST pin

below V the negative supply of the driving circuit should

D

–

threshold is referenced to the V pin. In order to provide

be tied to V of the LT6108.

TYPICAL APPLICATIONS

Overcurrent Battery Fault Protection

12 LITHIUM

40V CELL STACK

IRF9640

TO LOAD

0.1Ω

+

10µF

100k

6.2V*

R10

100Ω

8

1

6

SENSEHI SENSELO

+

7

V

OUTA

V

OUT

0.8A

OVERCURRENT

DETECTION

5V

LT6108-1

+

9.53k

475Ω

5

2

3

100k

10k

RESET

EN/RST

INC

–

OUTC

V

2N7000

4

610812 TA02

*CMH25234B

MCU Interfacing with Hardware Interrupts

0.1Ω

TO LOAD

Example:

+

V

5V

0V

OUTC GOES LOW

100Ω

8

1

SENSEHI SENSELO

+

6

7

V

OUT

V

OUTA

ADC IN

MCU INTERUPT

AtMega1280

PB0

LT6108-1

5

6

7

2

3

1

2

3

RESET

8.66k

1.33k

EN/RST

PB1

PCINT2

PCINT3

ADC2

OVERCURRENT ROUTINE

RESET COMPARATOR

5

5V

OUTC

INC

–

V

/ADC IN

OUT

10k

4

PB5

6108 TA03

610812 TA03b

The comparator is set to have a 300mA overcurrent

threshold. The MCU will receive the comparator output as

a hardware interrupt and immediately run an appropriate

fault routine.

610812fa

23

LT6108-1/LT6108-2

TYPICAL APPLICATIONS

Simpliﬁed DC Motor Torque Control

V

MOTOR

100µF

1k

0.1Ω

8

7

1

6

SENSEHI SENSELO

+

CURRENT SET POINT (0V TO 5V)

BRUSHED

DC MOTOR

(0A TO 5A)

MABUCHI

RS-540SH

V

OUTA

V

OUT

1µF

0.47µF

1N5818

100k

LT6108-1

9k

1k

5V

2

3

5

EN/RST

INC

RESET

5

–

2

4

7

+

V

1

3

6

MOD OUT

IRF640

OUTC

³+

–

LTC6246

V

LTC6992-1

4

100k

4

78.7k

SET DIV

GND

5V

280k

1M

2

610812 TA04

The ﬁgure above shows a simpliﬁed DC motor control

circuit. The circuit controls motor current, which is pro-

portional to motor torque; the LT6108 is used to provide

current feedback to an integrator that servos the motor

current to the current set point. The LTC^®6992 is used to

convert the output of the difference amp to the motors

PWM control signal.

Power-On Reset or Disconnect Using TimerBlox^®Circuit

5V

7

+

V

LT6108-1

SENSEHI

+

R

V

IN

100Ω

8

1

–

+

R7

10k

R

SENSE

SENSELO

OUTA

INC

6

5

I

–

+

LOAD

V

R1

9.53k

V

5V

–

+

3

2

OUTC

R2

499Ω

R6

30k

CREATES A DELAYED

10µs RESET PULSE

ON START-UP

C1

0.1µF

Q1

2N2222

400mV

REFERENCE

EN/RST

TRIG

GND

SET

OUT

OPTIONAL:

LTC6993-3

R4

1M

DISCHARGES C1

WHEN SUPPLY

+

V

–

V

IS DISCONNECTED

610812 TA06

4

DIV

R5

487k

TheLTC6993-3providesa10µsresetpulsetotheLT6108-1.

TheresetpulseisdelayedbyR4andC1whosetimeconstant

must be greater than 10ms and longer than the supply

turn-on time. Optional components R6 and Q1 discharge

capacitor C1 when the supply and/or ground are discon-

nected. This ensures that when the power supply and/or

ground are restored, capacitor C1 can fully recharge and

triggertheLTC6993-3toproduceanothercomparatorreset

pulse. These optional components are particularly useful

if the power and/or ground connections are intermittent,

as can occur when PCB are plugged into a connector.

610812fa

24

LT6108-1/LT6108-2

TYPICAL APPLICATIONS

LT6108-2 with External Latch and Power-On Reset or Disconnect

5V

7

+

V

LT6108-2

SENSEHI

+

R

V

IN

100Ω

8

1

–

+

R3

10k

R

SENSE

SENSELO

OUTA

INC

6

5

I

–

+

LOAD

V

R7

9.53k

R1

24.9k

V

VTH

–

+

3

OUTC

R8

499Ω

400mV

REFERENCE

–

V

4

R5*

R9*

30k

V

DD

100k

R2

200k

Q1*

2N2222

C1

0.1µF

R4*

3.4k

610812 TA06

*OPTIONAL COMPONENT

R6

1M

The input rising edge threshold which causes the output

to transition from high to low is:

An external latch is implemented with positive feedback.

R6 and C1 provide a reset pulse on power-up. The time

constant formed by R6 and C1 should be set slower than

that of the supply. Optional components R9 and Q1 dis-

charge capacitor C1 when the supply and/or ground are

disconnected. This ensures that when the power supply

and/orgroundarerestored,capacitorC1canfullyrecharge.

While C1 is charging, the NOR gate output is low, ensuring

that the comparator powers up in the correct state. These

optional components are particularly useful if the power

and/or ground connections are intermittent, as can occur

when PCB are plugged into a connector. R4 and R5 are

optional and minimize the movement of the rising input

threshold voltage.













400mV

V_{TH R}= 400mV if R4 = R5 •

( )

V – 400mV

DD

The input falling edge which causes the output to transi-

tion from low to high is:







 







1

V_DD•R1

R2+ R4||R5

V_{TH F}= 390mV • R1 •

+

–

( )

 

 

R1 R2+ R4||R5

610812fa

25

LT6108-1/LT6108-2

TYPICAL APPLICATIONS

Precision Power-On Reset Using TimerBlox Circuit

5V

7

+

V

LT6108-1

+

R

V

IN

100Ω

8

1

SENSEHI

SENSELO

–

+

R3

10k

R

SENSE

OUTA

INC

6

5

I

–

+

LOAD

V

R1

9.53k

V

R8

–

+

100k

3

2

OUTC

R2

499Ω

1 SECOND DELAY

ON START-UP

10µs RESET PULSE

GENERATOR

400mV

REFERENCE

EN/RST

TRIG

OUT

TRIG

OUT

C1

0.1µF

LTC6994-1

LTC6993-1

+

GND

V

GND

V

–

V

C2

0.1µF

R6

1M

610812 TA08

4

SET

DIV

SET

DIV

R7

191k

R5

681k

R4

487k

610812fa

26

LT6108-1/LT6108-2

PACKAGE DESCRIPTION

Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.

MS8 Package

8-Lead Plastic MSOP

(Reference LTC DWG # 05-08-ꢀꢂꢂ0 Rev F)

0.889 0.ꢀꢁ7

(.035 .005)

5.ꢁ3

3.ꢁ0 – 3.45

(.ꢁ0ꢂ)

(.ꢀꢁꢂ – .ꢀ3ꢂ)

MIN

3.00 0.ꢀ0ꢁ

(.ꢀꢀ8 .004)

(NOTE 3)

0.5ꢁ

(.0ꢁ05)

REF

0.ꢂ5

(.0ꢁ5ꢂ)

BSC

0.4ꢁ 0.038

(.0ꢀꢂ5 .00ꢀ5)

TYP

8

7 ꢂ 5

RECOMMENDED SOLDER PAD LAYOUT

3.00 0.ꢀ0ꢁ

(.ꢀꢀ8 .004)

(NOTE 4)

4.90 0.ꢀ5ꢁ

(.ꢀ93 .00ꢂ)

DETAIL “A”

0.ꢁ54

(.0ꢀ0)

0° – ꢂ° TYP

GAUGE PLANE

ꢀ

ꢁ

3

4

0.53 0.ꢀ5ꢁ

(.0ꢁꢀ .00ꢂ)

ꢀ.ꢀ0

(.043)

MAX

0.8ꢂ

(.034)

REF

DETAIL “A”

0.ꢀ8

(.007)

SEATING

PLANE

0.ꢁꢁ – 0.38

0.ꢀ0ꢀꢂ 0.0508

(.009 – .0ꢀ5)

(.004 .00ꢁ)

0.ꢂ5

(.0ꢁ5ꢂ)

BSC

TYP

MSOP (MS8) 0307 REV F

NOTE:

ꢀ. DIMENSIONS IN MILLIMETER/(INCH)

ꢁ. DRAWING NOT TO SCALE

3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.

MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.ꢀ5ꢁmm (.00ꢂ") PER SIDE

4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.

INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.ꢀ5ꢁmm (.00ꢂ") PER SIDE

5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.ꢀ0ꢁmm (.004") MAX

610812fa

27

LT6108-1/LT6108-2

PACKAGE DESCRIPTION

Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.

DCB Package

8-Lead Plastic DFN (2mm × 3mm)

(Reference LTC DWG # 05-08-ꢀ7ꢀ8 Rev A)

0.70 0.05

ꢀ.35 0.05

3.50 0.05

ꢀ.65 0.05

2.ꢀ0 0.05

PACKAGE

OUTLINE

0.25 0.05

0.45 BSC

ꢀ.35 REF

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED

R = 0.ꢀꢀ5

2.00 0.ꢀ0

(2 SIDES)

0.40 0.ꢀ0

TYP

5

R = 0.05

TYP

8

ꢀ.35 0.ꢀ0

ꢀ.65 0.ꢀ0

3.00 0.ꢀ0

(2 SIDES)

PIN ꢀ NOTCH

PIN ꢀ BAR

TOP MARK

(SEE NOTE 6)

R = 0.20 OR 0.25

× 45° CHAMFER

(DCB8) DFN 0ꢀ06 REV A

4

ꢀ

0.23 0.05

0.45 BSC

0.75 0.05

0.200 REF

ꢀ.35 REF

BOTTOM VIEW—EXPOSED PAD

0.00 – 0.05

NOTE:

ꢀ. DRAWING IS NOT A JEDEC PACKAGE OUTLINE

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE

MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.ꢀ5mm ON ANY SIDE

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN ꢀ LOCATION ON THE

TOP AND BOTTOM OF PACKAGE

610812fa

28

LT6108-1/LT6108-2

REVISION HISTORY

REV

DATE

DESCRIPTION

PAGE NUMBER

A

12/12 Addition of A-grade Performance and Electrical Characteristics

Addition of A-grade Order Information

1, 3, 4, 5, 12, 13, 16 (Fig10), 28

2

Clariﬁcation to Absolute Maximum Short Circuit Duration

Clariﬁcation to nomenclature used in Typical Performance Characteristics

Clariﬁcation to Description of EN/RST Pin Function

2

6, 7, 9

10

Internal Reference Block redrawn for consistency

11, 13, 18, 19

Additional information provided to Reverse Supply Protection

Correction to Overcurrent Battery Fault Protection diagram

22

23

610812fa

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

29

LT6108-1/LT6108-2

TYPICAL APPLICATION

ADC Driving Application

SENSE

HIGH

SENSE

LOW

0.1Ω

IN

OUT

0.1µF

V

CC

V

REF

100Ω

COMP

8

7

1

SENSEHI SENSELO

+

6

+

V

OUTA

IN

LTC2470

TO

MCU

LT6108-1

V

CC

0.1µF

2

3

RESET

8.66k

1.33k

EN/RST

10k

5

OUTC

INC

–

V

4

OVERCURRENT

6108 TA05

The low sampling current of the LTC2470 16-bit delta

sigma ADC is ideal for the LT6108.

RELATED PARTS

PART NUMBER DESCRIPTION

COMMENTS

LT1787

LTC4150

LT6100

LTC6101

LTC6102

LTC6103

LTC6104

LT6105

LT6106

LT6107

LT6109

Bidirectional High Side Current Sense Ampliﬁer

Coulomb Counter/Battery Gas Gauge

2.7V to 60V, 75µV Offset, 60µA Quiescent, 8V/V Gain

Indicates Charge Quantity and Polarity

Gain-Selectable High Side Current Sense Ampliﬁer

High Voltage High Side Current Sense Ampliﬁer

Zero Drift High Side Current Sense Ampliﬁer

Dual High Side Current Sense Ampliﬁer

4.1V to 48V, Gain Settings: 10, 12.5, 20, 25, 40, 50V/V

Up to 100V, Resistor Set Gain, 300µV Offset, SOT-23

Up to 100V, Resistor Set Gain, 10µV Offset, MSOP8/DFN

4V to 60V, Resistor Set Gain, 2 Independent Amps, MSOP8

4V to 60V, Separate Gain Control for Each Direction, MSOP8

–0.3V to 44V Input Range, 300µV Offset, 1% Gain Error

2.7V to 36V, 250µV Offset, Resistor Set Gain, SOT-23

2.7V to 36V, –55°C to 150°C, Fully Tested: –55°C, 25°C, 150°C

2.7V to 60V, 125µV, Resistor Set Gain, 1.25% Threshold Error

Bidirectional High Side Current Sense Ampliﬁer

Precision Rail-to-Rail Input Current Sense Amplifer

Low Cost High Side Current Sense Ampliﬁer

High Temperature High Side Current Sense Ampliﬁer

High Side Current Sense Ampliﬁer with Reference and

Comparators

LT6700

Dual Comparator with 400mV Reference

1.4V to 18V, 6.5µA Supply Current

610812fa

LT 1212 REV A • PRINTED IN USA

30 ^{LinearTechnology Corporation}

1630 McCarthy Blvd., Milpitas, CA 95035-7417

●

 LINEAR TECHNOLOGY CORPORATION 2011

(408) 432-1900 FAX: (408) 434-0507 www.linear.com


型号：	LT6108IDCB-1#PBF
厂家：	Linear
描述：	LT6108 - High Side Current Sense Amplifier with Reference and Comparator; Package: DFN; Pins: 8; Temperature Range: -40°C to 85°C 光电二极管
文件：	总30页 (文件大小：398K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LT6108IDCB-1#PBF [Linear]

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