LT1373HVCS8 [Linear]
250kHz Low Supply Current High Efficiency 1.5A Switching Regulator; 250kHz的低电源电流高效率1.5A开关稳压器
| 型号： | LT1373HVCS8 |
| 厂家： | 
Linear |
| 描述： | 250kHz Low Supply Current High Efficiency 1.5A Switching Regulator |
| 文件： | 总12页 (文件大小：184K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT1373
250kHz Low Supply Current
High Efficiency
1.5A Switching Regulator
U
FEATURES
DESCRIPTIO
The LT®1373 is a low supply current high frequency
current mode switching regulator. It can be operated in all
standard switching configurations including boost, buck,
flyback, forward, inverting and “Cuk.” A 1.5A high effi-
ciency switch is included on the die, along with all oscilla-
tor, control and protection circuitry. All functions of the
LT1373 are integrated into 8-pin SO/PDIP packages.
■
1mA IQ at 250kHz
■
Uses Small Inductors: 15µH
■
All Surface Mount Components
■
Only 0.6 Square Inch of Board Space
■
Low Minimum Supply Voltage: 2.7V
■
Constant Frequency Current Mode
■
Current Limited Power Switch: 1.5A
■
Regulates Positive or Negative Outputs
Compared to the 500kHz LT1372, which draws 4mA of
quiescent current, the LT1373 switches at 250kHz, typi-
cally consumes only 1mA and has higher efficiency. High
frequencyswitchingallowsforsmallinductorstobeused.
All surface mount components consume less than 0.6
square inch of board space.
■
Shutdown Supply Current: 12µA Typ
■
Easy External Synchronization
■
8-Pin SO or PDIP Packages
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APPLICATIO S
New design techniques increase flexibility and maintain
ease of use. Switching is easily synchronized to an exter-
nal logic level source. A logic low on the shutdown pin
reduces supply current to 12µA. Unique error amplifier
circuitry can regulate positive or negative output voltage
while maintaining simple frequency compensation tech-
niques. Nonlinear error amplifier transconductance re-
duces output overshoot on start-up or overload recovery.
Oscillator frequency shifting protects external compo-
nents during overload conditions.
■
Boost Regulators
■
CCFL Backlight Driver
■
Laptop Computer Supplies
■
Multiple Output Flyback Supplies
■
Inverting Supplies
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATIO
12V Output Efficiency
5V-to-12V Boost Converter
100
V
= 5V
IN
f = 250kHz
D1
5V
L1*
22µH
MBRS120T3
†
V
90
80
70
60
50
OUT
12V
5
R1
215k
1%
V
IN
4
8
2
ON
V
S/S
SW
FB
OFF
+
C4**
22µF
LT1373
+
C1**
22µF
†MAX I
L1
OUT
GND
V
C
1
I
OUT
15µH 0.3A
22µH 0.35A
R2
6, 7
24.9k
1%
C2
0.01µF
*SUMIDA CD75-220KC (22µH) OR
COILCRAFT D03316-153 (15µH)
AVX TPSD226M025R0200
R3
5k
1
10
100
1000
**
OUTPUT CURRENT (mA)
LT1373 • TA01
LT1373 • TA02
1
LT1373
W W U W
U W
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ABSOLUTE AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
(Note 1)
Supply Voltage ....................................................... 30V
Switch Voltage
ORDER PART
TOP VIEW
NUMBER
V
1
2
3
4
V
SW
8
7
6
5
LT1373 ............................................................... 35V
LT1373HV .......................................................... 42V
S/S Pin Voltage....................................................... 30V
Feedback Pin Voltage (Transient, 10ms) .............. ±10V
Feedback Pin Current........................................... 10mA
Negative Feedback Pin Voltage
(Transient, 10ms)............................................. ±10V
Operating Junction Temperature Range
Commercial ........................................ 0°C to 125°C*
Industrial ......................................... –40°C to 125°C
Short Circuit ......................................... 0°C to 150°C
Storage Temperature Range ................ –65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
C
LT1373CN8
LT1373IN8
FB
NFB
S/S
GND
LT1373HVCN8
LT1373CS8
LT1373HVIN8
LT1373IS8
GND S
V
IN
LT1373HVCS8
LT1373HVIS8
N8 PACKAGE
8-LEAD PDIP
S8 PACKAGE
8-LEAD PLASTIC SO
S8 PART MARKING
TJMAX = 125°C, θJA = 100°C/ W (N8)
TJMAX = 125°C, θJA = 120°C/ W (S8)
1373
1373H
1373I 1373HI
Consult factory for Military grade parts.
*Units shipped prior to Date Code 9552 are rated at 100°C maximum
operating temperature.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
SYMBOL PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
V
Reference Voltage
Measured at Feedback Pin
V = 0.8V
C
1.230
1.225
1.245
1.245
1.260
1.265
V
V
REF
●
I
Feedback Input Current
V
= V
REF
50
150
275
nA
nA
FB
FB
●
●
Reference Voltage Line Regulation
Negative Feedback Reference Voltage
2.7V ≤ V ≤ 25V, V = 0.8V
0.01
0.03
%/V
IN
C
V
Measured at Negative Feedback Pin
Feedback Pin Open, V = 0.8V
–2.51
–2.55
–2.45
–2.45
–2.39
–2.35
V
V
NFB
●
●
●
C
I
Negative Feedback Input Current
V
= V
–12
–7
–2
µA
NFB
NFB
NFR
Negative Feedback Reference Voltage
Line Regulation
2.7V ≤ V ≤ 25V, V = 0.8V
0.01
0.05
%/V
IN
C
g
Error Amplifier Transconductance
∆I = ±5µA
250
150
375
500
600
µmho
µmho
m
C
●
●
●
Error Amplifier Source Current
Error Amplifier Sink Current
Error Amplifier Clamp Voltage
V
V
= V
– 150mV, V = 1.5V
25
50
90
µA
µA
FB
FB
REF
C
= V + 150mV, V = 1.5V
850
1500
REF
C
High Clamp, V = 1V
Low Clamp, V = 1.5V
1.70
0.25
1.95
0.40
2.30
0.52
V
V
FB
FB
A
f
Error Amplifier Voltage Gain
250
1
V/V
V
V
V Pin Threshold
C
Duty Cycle = 0%
0.8
1.25
Switching Frequency
2.7V ≤ V ≤ 25V
225
210
200
250
250
275
290
290
kHz
kHz
kHz
IN
0°C ≤ T ≤ 125°C
●
●
J
–40°C ≤ T ≤ 0°C (I Grade)
J
Maximum Switch Duty Cycle
90
95
%
Switch Current Limit Blanking Time
340
500
ns
2
LT1373
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted.
SYMBOL PARAMETER
BV Output Switch Breakdown Voltage
CONDITIONS
MIN
TYP
MAX
UNITS
LT1373
LT1373HV
●
●
35
47
V
0°C ≤ T ≤ 125°C
42
40
47
V
V
J
–40°C ≤ T ≤ 0°C (I Grade)
J
V
SAT
Output Switch “On” Resistance
I
= 1A
SW
●
0.5
0.85
Ω
I
Switch Current Limit
Duty Cycle = 50%
Duty Cycle = 80% (Note 2)
●
●
1.5
1.3
1.9
1.7
2.7
2.5
A
A
LIM
∆I
IN
Supply Current Increase During Switch On-Time
10
20
mA/A
∆I
SW
Control Voltage to Switch Current
Transconductance
2
A/V
Minimum Input Voltage
Supply Current
●
●
2.4
1
2.7
1.5
V
I
2.7V ≤ V ≤ 25V
mA
Q
IN
Shutdown Supply Current
2.7V ≤ V ≤ 25V, V ≤ 0.6V
IN S/S
0°C ≤ T ≤ 125°C
●
12
30
50
µA
µA
J
–40°C ≤ T ≤ 0°C (I Grade)
J
Shutdown Threshold
2.7V ≤ V ≤ 25V
●
●
●
●
0.6
5
1.3
12
2
V
µs
IN
Shutdown Delay
100
15
S/S Pin Input Current
Synchronization Frequency Range
0V ≤ V ≤ 5V
–10
300
µA
S/S
340
kHz
Note 1: Absolute Maximum Ratings are those values beyond which the life
of the device may be impaired.
Note 2: For duty cycles (DC) between 50% and 90%, minimum
guaranteed switch current is given by I = 0.667 (2.75 – DC).
LIM
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TYPICAL PERFOR A CE CHARACTERISTICS
Switch Saturation Voltage
vs Switch Current
Switch Current Limit
vs Duty Cycle
Minimum Input Voltage
vs Temperature
3.0
2.5
2.0
1.5
1.0
0.5
0
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
3.0
2.8
2.6
2.4
2.2
2.0
1.8
150°C
100°C
25°C
25°C AND
125°C
–55°C
–55°C
0
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
SWITCH CURRENT (A)
75 100
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
–50 –25
0
25 50
125 150
TEMPERATURE (°C)
LT1373 • G01
LT1373 • G02
LT1373 • G03
3
LT1373
TYPICAL PERFOR A CE CHARACTERISTICS
U W
Shutdown Delay and Threshold
vs Temperature
Minimum Synchronization
Voltage vs Temperature
Error Amplifier Output Current
vs Feedback Pin Voltage
20
18
16
14
12
10
8
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
100
75
3.0
2.5
2.0
1.5
1.0
0.5
0
f
= 330kHz
SYNC
25°C
–55°C
SHUTDOWN
THRESHOLD
50
125°C
25
SHUTDOWN
DELAY
0
6
–25
–50
–75
4
2
0
–50
50
100 125
150
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
–0.3
–0.2
–0.1
V
0.1
–25
0
25
75
REF
TEMPERATURE (°C)
FEEDBACK PIN VOLTAGE (V)
LT1373 • G04
LT1373 • G05
LT1373 • G06
S/S Pin Input Current
vs Voltage
Switching Frequency
Error Amplifier Transconductance
vs Temperature
vs Feedback Pin Voltage
500
400
300
200
100
0
5
4
110
100
90
80
70
60
50
40
30
20
10
V
= 5V
∆I (V )
∆V (FB)
IN
C
g
=
m
3
2
1
0
–1
–2
–3
–4
–5
–50
50
100 125
150
–25
0
25
75
–1
0
1
2
3
4
5
6
7
8
9
0
0.1 0.2 0.3
0.6 0.7 0.8 0.9 1.0
0.4 0.5
TEMPERATURE (°C)
S/S PIN VOLTAGE (V)
FEEDBACK PIN VOLTAGE (V)
LT1373 • G09
LT1373 • G07
LT1373 • G08
VC Pin Threshold and High
Clamp Voltage vs Temperature
Feedback Input Current
vs Temperature
Negative Feedback Input
Current vs Temperature
2.4
2.2
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0
–2
400
350
300
250
200
150
100
50
V
NFB
= V
NFR
V
FB
= V
REF
V
C
HIGH CLAMP
–4
–6
–8
–10
–12
–14
–16
–18
–20
V
C
THRESHOLD
0
–50
50
100 125
–50
50
100 125
150
–25
0
25
75
150
–25
0
25
75
50 75
25
TEMPERATURE (°C)
–50 –25
0
100 125 150
TEMPERATURE (°C)
TEMPERATURE (°C)
LT1373 • G10
LT1373 • G12
LT1373 • G11
4
LT1373
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PI FU CTIO S
VC (Pin 1): Compensation Pin. The VC pin is used for
frequency compensation, current limiting and soft start. It
is the output of the error amplifier and the input of the
currentcomparator. Loopfrequencycompensationcanbe
performed with an RC network connected from the VC pin
to ground.
floating. To synchronize switching, drive the S/S pin be-
tween 300kHz and 340kHz.
VIN (Pin 5): Input Supply Pin. Bypass VIN with 10µF or
more. The part goes into undervoltage lockout when VIN
drops below 2.5V. Undervoltage lockout stops switching
and pulls the VC pin low.
FB (Pin 2): The feedback pin is used for positive output
voltage sensing and oscillator frequency shifting. It is the
inverting input to the error amplifier. The noninverting
input of this amplifier is internally tied to a 1.245V
reference. Load on the FB pin should not exceed 100µA
when the NFB pin is used. See Applications Information.
GND S (Pin 6): The ground sense pin is a “clean” ground.
The internal reference, error amplifier and negative feed-
back amplifier are referred to the ground sense pin. Con-
nect it to ground. Keep the ground path connection to the
output resistor divider and the VC compensation network
free of large ground currents.
NFB(Pin3):Thenegativefeedbackpinisusedfornegative
output voltage sensing. It is connected to the inverting
input of the negative feedback amplifier through a 400k
source resistor.
GND (Pin 7): The ground pin is the emitter connection of
thepowerswitchandhaslargecurrentsflowingthroughit.
It should be connected directly to a good quality ground
plane.
S/S (Pin 4): Shutdown and Synchronization Pin. The S/S
pin is logic level compatible. Shutdown is active low and
the shutdown threshold is typically 1.3V. For normal
operation, pull the S/S pin high, tie it to VIN or leave it
V
SW (Pin 8): The switch pin is the collector of the power
switch and has large currents flowing through it. Keep the
traces to the switching components as short as possible to
minimize radiation and voltage spikes.
W
BLOCK DIAGRA
V
SW
IN
SHUTDOWN
DELAY AND RESET
LOW DROPOUT
2.3V REG
S/S
ANTI-SAT
250kHz
LOGIC
DRIVER
SWITCH
SYNC
OSC
5:1 FREQUENCY
SHIFT
+
NEGATIVE
FEEDBACK
400k
200k
AMP
NFB
FB
–
COMP
–
+
–
ERROR
CURRENT
AMP
0.08Ω
AMP
+
A
≈ 6
V
V
C
1.245V
REF
GND LT1373 • BD
GND SENSE
5
LT1373
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OPERATIO
The LT1373 is a current mode switcher. This means that
switch duty cycle is directly controlled by switch current
rather than by output voltage. Referring to the Block
Diagram, the switch is turned “On” at the start of each
oscillator cycle. It is turned “Off” when switch current
reaches a predetermined level. Control of output voltage
is obtained by using the output of a voltage sensing error
amplifier to set current trip level. This technique has
several advantages. First, it has immediate response to
input voltage variations, unlike voltage mode switchers
which have notoriously poor line transient response.
Second,itreducesthe90°phaseshiftatmid-frequencies
in the energy storage inductor. This greatly simplifies
closed-loop frequency compensation under widely vary-
ing input voltage or output load conditions. Finally, it
allows simple pulse-by-pulse current limiting to provide
maximum switch protection under output overload or
short conditions. A low dropout internal regulator pro-
vides a 2.3V supply for all internal circuitry. This low
dropout design allows input voltage to vary from 2.7V to
25V with virtually no change in device performance. A
250kHz oscillator is the basic clock for all internal timing.
It turns “On” the output switch via the logic and driver
circuitry. Special adaptive anti-sat circuitry detects onset
of saturation in the power switch and adjusts driver
current instantaneously to limit switch saturation. This
minimizes driver dissipation and provides very rapid
turn-off of the switch.
put overshoot on start-up or overload recovery. When
the feedback voltage exceeds the reference by 40mV,
error amplifier transconductance increases ten times,
whichreducesoutputovershoot.Thefeedbackinputalso
invokes oscillator frequency shifting, which helps pro-
tect components during overload conditions. When the
feedback voltage drops below 0.6V, the oscillator fre-
quencyisreduced5:1.Lowerswitchingfrequencyallows
full control of switch current limit by reducing minimum
switch duty cycle.
Unique error amplifier circuitry allows the LT1373 to
directly regulate negative output voltages. The negative
feedback amplifier’s 400k source resistor is brought out
fornegativeoutputvoltagesensing. TheNFBpinregulates
at –2.45V while the amplifier output internally drives the
FB pin to 1.245V. This architecture, which uses the same
main error amplifier, prevents duplicating functions and
maintains ease of use. (Consult Linear Technology Mar-
keting for units that can regulate down to –1.25V.)
The error signal developed at the amplifier output is
brought out externally. This pin (VC) has three different
functions. Itisusedforfrequencycompensation, current
limit adjustment and soft starting. During normal regula-
tor operation this pin sits at a voltage between 1V (low
outputcurrent)and 1.9V(highoutputcurrent). Theerror
amplifierisacurrentoutput(gm)type, sothisvoltagecan
be externally clamped for lowering current limit. Like-
wise, acapacitorcoupledexternalclampwillprovidesoft
start. Switch duty cycle goes to zero if the VC pin is pulled
below the control pin threshold, placing the LT1373 in an
idle mode.
A 1.245V bandgap reference biases the positive input of
the error amplifier. The negative input of the amplifier is
broughtoutforpositiveoutputvoltagesensing.Theerror
amplifier has nonlinear transconductance to reduce out-
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APPLICATIO S I FOR ATIO
Positive Output Voltage Setting
V
OUT
The LT1373 develops a 1.245V reference (VREF) from the
FB pin to ground. Output voltage is set by connecting the
FB pin to an output resistor divider (Figure 1). The FB pin
bias current represents a small error and can usually be
ignoredforvaluesofR2upto25k.Thesuggestedvaluefor
R2 is 24.9k. The NFB pin is normally left open for positive
output applications.
R1
R2
V
= V
1 +
R1
OUT
REF
(
)
FB
PIN
V
OUT
1.245
– 1
R1 = R2
(
)
R2
V
REF
LT1373 • F01
Figure 1. Positive Output Resistor Divider
6
LT1373
W U U
APPLICATIO S I FOR ATIO
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A logic low on the S/S pin activates shutdown, reducing
the part’s supply current to 12µA. Typical synchronization
range is from 1.05 and 1.8 times the part’s natural switch-
ingfrequency,butisonlyguaranteedbetween300kHzand
340kHz. A 12µs resetable shutdown delay network guar-
antees the part will not go into shutdown while receiving
a synchronization signal.
Negative Output Voltage Setting
The LT1373 develops a –2.45V reference (VNFR) from the
NFB pin to ground. Output voltage is set by connecting the
NFB pin to an output resistor divider (Figure 2). The –7µA
NFBpinbiascurrent(INFB)cancauseoutputvoltageerrors
and should not be ignored. This has been accounted for in
the formula in Figure 2. The suggested value for R2 is
2.49k. The FB pin is normally left open for negative output
applications. SeeDualPolarityOutputVoltageSensingfor
limitations of FB pin loading when using the NFB pin.
Caution should be used when synchronizing above
330kHz because at higher sync frequencies the ampli-
tude of the internal slope compensation used to prevent
subharmonic switching is reduced. This type of
subharmonic switching only occurs when the duty cycle
of the switch is above 50%. Higher inductor values will
tend to eliminate problems.
–V
OUT
R1
R1
R2
–V
= V
V
1 +
+ I (R1)
NFB
OUT
NFB
(
)
I
NFB
NFB
PIN
– 2.45
OUT
R1 =
2.45
R2
–6
R2
+ (7 • 10
)
(
)
V
Thermal Considerations
NFR
LT1373 • F02
Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause exces-
sive die temperatures. The packages are rated at 120°C/W
for SO (S8) and 130°C/W for PDIP (N8).
Figure 2. Negative Output Resistor Divider
Dual Polarity Output Voltage Sensing
Certain applications benefit from sensing both positive
and negative output voltages. One example is the Dual
Output Flyback Converter with Overvoltage Protection
circuit shown in the Typical Applications section. Each
output voltage resistor divider is individually set as de-
scribed above. When both the FB and NFB pins are used,
the LT1373 acts to prevent either output from going
beyond its set output voltage. For example in this applica-
tion, if the positive output were more heavily loaded than
the negative, the negative output would be greater and
would regulate at the desired set-point voltage. The posi-
tive output would sag slightly below its set-point voltage.
This technique prevents either output from going unregu-
lated high at no load. Please note that the load on the FB
pin should not exceed 100µA when the NFB pin is used.
This situation occurs when the resistor dividers are used
at both FB and NFB. True load on FB is not the full divider
current unless the positive output is shorted to ground.
See Dual Output Flyback Converter application.
Average supply current (including driver current) is:
IIN = 1mA + DC (ISW/60 + ISW • 0.004)
ISW = switch current
DC = switch duty cycle
Switch power dissipation is given by:
PSW = (ISW)2 • RSW • DC
RSW = output switch “On” resistance
Total power dissipation of the die is the sum of supply
current times supply voltage plus switch power:
P
D(TOTAL) = (IIN • VIN) + PSW
Choosing the Inductor
For most applications the inductor will fall in the range of
10µHto50µH.Lowervaluesarechosentoreducephysical
size of the inductor. Higher values allow more output
current because they reduce peak current seen by the
power switch which has a 1.5A limit. Higher values also
reduce input ripple voltage, and reduce core loss.
Shutdown and Synchronization
The dual function S/S pin provides easy shutdown and
synchronization. It is logic level compatible and can be
pulledhigh, tiedtoVIN orleftfloatingfornormaloperation.
When choosing an inductor you might have to consider
maximum load current, core and copper losses, allowable
7
LT1373
W U U
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APPLICATIO S I FOR ATIO
inductor gets too hot, wire insulation will melt and cause
turn-to-turn shorts). Keep in mind that all good things
like high efficiency, low profile and high temperature
operation will increase cost, sometimes dramatically.
component height, output voltage ripple, EMI, fault cur-
rent in the inductor, saturation, and of course, cost. The
following procedure is suggested as a way of handling
thesesomewhatcomplicatedandconflictingrequirements.
1. Assume that the average inductor current (for a boost
converter) is equal to load current times VOUT/VIN and
decide whether or not the inductor must withstand
continuous overload conditions. If average inductor
current at maximum load current is 0.5A, for instance,
a 0.5A inductor may not survive a continuous 1.5A
overload condition. Also, be aware that boost convert-
ers are not short-circuit protected, and that under
outputshortconditions, inductorcurrentislimitedonly
by the available current of the input supply.
5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the Linear Technology application
department if you feel uncertain about the final choice.
They have experience with a wide range of inductor
types and can tell you about the latest developments in
low profile, surface mounting, etc.
Output Capacitor
The output capacitor is normally chosen by its effective
series resistance (ESR), because this is what determines
output ripple voltage. At 500kHz, any polarized capacitor
is essentially resistive. To get low ESR takes volume, so
physically smaller capacitors have high ESR. The ESR
range for typical LT1373 applications is 0.05Ω to 0.5Ω. A
typical output capacitor is an AVX type TPS, 22µF at 25V,
with a guaranteed ESR less than 0.2Ω. This is a “D” size
surface mount solid tantalum capacitor. TPS capacitors
are specially constructed and tested for low ESR, so they
give the lowest ESR for a given volume. To further reduce
ESR, multiple output capacitors can be used in parallel.
The value in microfarads is not particularly critical and
values from 22µF to greater than 500µF work well, but you
cannot cheat mother nature on ESR. If you find a tiny 22µF
solid tantalum capacitor, it will have high ESR and output
ripple voltage will be terrible. Table 1 shows some typical
solid tantalum surface mount capacitors.
2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, espe-
cially with smaller inductors and lighter loads, so don’t
omit this step. Powered iron cores are forgiving be-
cause they saturate softly, whereas ferrite cores satu-
rate abruptly. Other core materials fall in between
somewhere. The following formula assumes continu-
ous mode operation, but it errors only slightly on the
high side for discontinuous mode, so it can be used for
all conditions.
V
V
V (V
2(f)(L)(V
– V )
OUT
IN OUT
IN
I
= I
•
+
PEAK
OUT
)
IN
OUT
V = minimum input voltage
IN
f = 250kHz switching frequency
3. Decide if the design can tolerate an “open” core geom-
etry like a rod or barrel, which have high magnetic field
radiation, or whether it needs a closed core like a toroid
to prevent EMI problems. One would not want an open
core next to a magnetic storage media for instance!
This is a tough decision because the rods or barrels are
temptingly cheap and small, and there are no helpful
guidelines to calculate when the magnetic field radia-
tion will be a problem.
Table 1. Surface Mount Solid Tantalum Capacitor
ESR and Ripple Current
E CASE SIZE
ESR (MAX Ω)
RIPPLE CURRENT (A)
AVX TPS, Sprague 593D
AVX TAJ
0.1 to 0.3
0.7 to 0.9
0.7 to 1.1
0.4
D CASE SIZE
AVX TPS, Sprague 593D
AVX TAJ
0.1 to 0.3
0.9 to 2.0
0.7 to 1.1
0.36 to 0.24
C CASE SIZE
AVX TPS
AVX TAJ
0.2 (Typ)
1.8 to 3.0
0.5 (Typ)
4. Start shopping for an inductor which meets the require-
ments of core shape, peak current (to avoid saturation),
averagecurrent(tolimitheating),andfaultcurrent,(ifthe
0.22 to 0.17
B CASE SIZE
AVX TAJ
2.5 to 10
0.16 to 0.08
8
LT1373
W U U
APPLICATIO S I FOR ATIO
U
Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true and type TPS capacitors are
speciallytestedforsurgecapability,butsurgeruggedness
is not a critical issue with the output capacitor. Solid
tantalum capacitors fail during very high turn-on surges,
which do not occur at the output of regulators. High
discharge surges, such as when the regulator output is
dead shorted, do not harm the capacitors.
aluminum electrolytic capacitors may also be used and
have a high tolerance to turn-on surges.
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becomingavailableinsmallercasesizes.Thesearetempt-
ing for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generatesaloop“zero”at5kHzto50kHzthatisinstrumen-
tal in giving acceptable loop phase margin. Ceramic ca-
pacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
They are appropriate for input bypassing because of their
highripplecurrentratingsandtoleranceofturn-onsurges.
Linear Technology plans to issue a Design Note on the use
of ceramic capacitors in the near future.
Single inductor boost regulators have large RMS ripple
current in the output capacitor, which must be rated to
handle the current. The formula to calculate this is:
Output Capacitor Ripple Current (RMS)
DC
I
(RMS) = I
= I
1 – DC
RIPPLE
OUT
V
– V
IN
IN
OUT
OUT
Output Diode
V
The suggested output diode (D1) is a 1N5818 Schottky or
its Motorola equivalent, MBR130. It is rated at 1A average
forward current and 30V reverse voltage. Typical forward
voltage is 0.42V at 1A. The diode conducts current only
during switch-off time. Peak reverse voltage for boost
converters is equal to regulator output voltage. Average
forward current in normal operation is equal to output
current.
Input Capacitors
The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular,
and does not contain large squarewave currents as is
found in the output capacitor. Capacitors in the range of
10µF to 100µF with an ESR (effective series resistance) of
0.3Ω or less work well up to a full 1.5A switch current.
Higher ESR capacitors may be acceptable at low switch
currents. Input capacitor ripple current for boost con-
verter is:
Frequency Compensation
Loop frequency compensation is performed on the output
oftheerroramplifier(VC pin)withaseriesRC network.The
main pole is formed by the series capacitor and the output
impedance (≈1MΩ) of the error amplifier. The pole falls in
the range of 5Hz to 30Hz. The series resistor creates a
“zero”at2kHzto10kHz, whichimprovesloopstabilityand
transientresponse.Asecondcapacitor,typicallyonetenth
thesizeofthemaincompensationcapacitor,issometimes
used to reduce the switching frequency ripple on the VC
pin. VC pin ripple is caused by output voltage ripple
attenuatedbytheoutputdividerandmultipliedbytheerror
amplifier. Without the second capacitor, VC pin ripple is:
0.3(V )(V
– V )
IN
IN OUT
(f)(L)(V
I
=
RIPPLE
)
OUT
f = 250kHz switching frequency
Theinputcapacitorcanseeaveryhighsurgecurrentwhen
a battery or high capacitance source is connected “live”,
andsolidtantalumcapacitorscanfailunderthiscondition.
Several manufacturers have developed a line of solid
tantalum capacitors specially tested for surge capability
(AVX TPS series, for instance), but even these units may
fail if the input voltage approaches the maximum voltage
rating of the capacitor. AVX recommends derating capaci-
torvoltageby2:1forhighsurgeapplications. Ceramicand
1.245(V
)(g )(R )
m C
OUT
RIPPLE
V
V Pin Ripple =
C
9
LT1373
APPLICATIO S I FOR ATIO
W U U
U
VRIPPLE = output ripple (VP-P
)
Thehighspeedswitchingcurrentpathisshownschemati-
cally in Figure 3. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, output diode and output capacitor is
the only one containing nanosecond rise and fall times.
Keep this path as short as possible.
gm = error amplifier transconductance (≈375µmho)
RC = series resistor on VC pin
VOUT = DC output voltage
To prevent irregular switching, VC pin ripple should be
kept below 50mVP-P. Worst-case VC pin ripple occurs at
maximum output load current and will also be increased if
poor quality (high ESR) output capacitors are used. The
addition of a 0.001µF capacitor on the VC pin reduces
switching frequency ripple to only a few millivolts. A low
value for RC will also reduce VC pin ripple, but loop phase
margin may be inadequate.
SWITCH
NODE
L1
V
OUT
HIGH
FREQUENCY
CIRCULATING
PATH
V
IN
LOAD
Switch Node Considerations
LT1373 • F03
Formaximumefficiency,switchriseandfalltimearemade
as short as possible. To prevent radiation and high fre-
quency resonance problems, proper layout of the compo-
nents connected to the switch node is essential. B field
(magnetic) radiation is minimized by keeping output di-
ode, switchpinandoutputbypasscapacitorleadsasshort
as possible. E field radiation is kept low by minimizing the
length and area of all traces connected to the switch pin.
A ground plane should always be used under the switcher
circuitry to prevent interplane coupling.
Figure 3
More Help
For more detailed information on switching regulator
circuits, please see AN19. Linear Technology also offers a
computer software program, SwitcherCADTM, to assist in
designing switching converters. In addition, our applica-
tions department is always ready to lend a helping hand.
SwitcherCAD is a trademark of Linear Technology Corporation.
U
TYPICAL APPLICATIONS N
Positive-to-Negative Converter with Direct Feedback
Dual Output Flyback Converter with Overvoltage Protection
V
R2
275k
1%
R1
302.6k
1%
IN
2.7V TO 16V
T1*
+
2
4
3
C1
D2
+
•
V
IN
22µF
C3
P6KE-15A
D3
MBRS140T3
T1*
4.75V TO 13V
5
47µF
V
OUT
•
V
IN
1N4148
+
15V
†
2, 3
5
1
4
8
3
–V
C1
100µF
ON
+
OUT
V
S/S
SW
•
P6KE-20A
OFF
C3
–5V
R2
2.55k
1%
D1
MBRS130LT3
47µF
2
5
LT1373
4
8
1N4148
•
•
FB
S/S
V
IN
V
NFB
4
8
3
6, 7
ON
+
R3
2.49k
1%
SW
OFF
C4
47µF
†MAX I
V
GND
C
OUT
V
0.3A 3V
0.5A 5V
0.75A 9V
LT1373
1
6, 7
1
I
–V
OUT
IN
OUT
NFB
–15V
C2
0.01µF
R4
12.4k
1%
MBRS140T3
V
GND
C
R1
5k
1
6, 7
R5
2.49k
1%
C2
*COILTRONICS CTX20-2 (407) 241-7876
LT1373 • TA03
0.01µF
R3
5k
*DALE LPE-4841-100MB (605) 665-9301
LT1373 • TA04
10
LT1373
U
TYPICAL APPLICATIO S
Low Ripple 5V to –3V “Cuk”† Converter
V
OUT
L1*
V
IN
–3V
5V
2
1
3
4
250mA
•
•
R1
1k
1%
C2
47µF
16V
5
4
7
6
8
V
V
SW
IN
+
+
C1
22µF
10V
C6
0.1µF
S/S
LT1373
3
1
GND
NFB
GND S
V
C
C3
47µF
16V
D1**
+
R4
5k
R2
5.49k
1%
C4
0.01µF
*SUMIDA CLS62-100L
**MOTOROLA MBR0520LT3
†PATENTS MAY APPLY
LT1373 • TA05
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.400*
(10.160)
MAX
0.130 ± 0.005
0.300 – 0.325
0.045 – 0.065
(3.302 ± 0.127)
(1.143 – 1.651)
(7.620 – 8.255)
8
1
7
6
5
4
0.065
(1.651)
TYP
0.255 ± 0.015*
(6.477 ± 0.381)
0.009 – 0.015
(0.229 – 0.381)
0.125
0.020
(0.508)
MIN
(3.175)
MIN
+0.035
–0.015
2
3
0.325
0.018 ± 0.003
0.100
(2.54)
BSC
N8 1098
+0.889
8.255
(0.457 ± 0.076)
(
)
–0.381
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
0.010 – 0.020
(0.254 – 0.508)
7
5
8
6
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0°– 8° TYP
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
0.016 – 0.050
(0.406 – 1.270)
0.050
(1.270)
BSC
0.014 – 0.019
(0.355 – 0.483)
TYP
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
SO8 1298
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
1
2
3
4
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of circuits as described herein will not infringe on existing patent rights.
11
LT1373
U
TYPICAL APPLICATIO S
90% Efficient CCFL Supply
5mA MAX
Two Li-Ion Cells to 5V SEPIC Conveter
V
IN
4V TO 9V
LAMP
C2
27pF
D1
1N4148
10
T1
V
L1A
33µH
IN
5
4
3
2
1
5
C2
2.2µF
D1
4.5V
MBRS130LT3
+
V
TO 30V
IN
•
†
4
8
2
V
OUT
5V
10µF
ON
C1
0.1µF
V
S/S
SW
FB
OFF
C1
•
R2
LT1373
+
L1B
33µH
75k
1%
33µF
330Ω
1N5818
20V
+
Q1
Q2
C3
100µF
GND
6, 7
V
C
1
10V
R3
24.9k
1%
R1
5k
C4
0.01µF
2.7V TO
5.5V
D2
1N4148
L1
100µH
+
2.2µF
5
V
562Ω*
†MAX I
IN
10k
OUT
V
4
8
ON
S/S
V
SW
I
OFF
OUT
IN
C1 = AVX TPSD 336M020R0200
C2 = TOKIN 1E225ZY5U-C203-F
C3 = AVX TPSD 107M010R0100
L1 = COILTRONICS CTX33-2, SINGLE
INDUCTOR WITH TWO WINDINGS
20k
DIMMING
0.45A 4V
0.55A 5V
0.65A 7V
0.72A 9V
LT1373
2
V
FB
LT1373 • TA07
GND
V
C
0.1µF
22k
1N4148
6, 7
1
+
2µF
OPTIONAL REMOTE
DIMMING
LT1372 • TA06
C1 = WIMA MKP-20
L1 = COILCRAFT D03316-104
CCFL BACKLIGHT APPLICATION CIRCUITS
CONTAINED IN THIS DATA SHEET ARE
Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001 COVERED BY U.S. PATENT NUMBER 5408162
T1 = COILTRONICS CTX 110609
* = 1% FILM RESISTOR
AND OTHER PATENTS PENDING
DO NOT SUBSTITUTE COMPONENTS
COILTRONICS (407) 241-7876
COILCRAFT (708) 639-6400
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100kHz 1.25A Boost Switching Regulator
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Also for Flyback, Buck and Inverting Configurations
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Converts 2V to 5V at 600mA
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600kHz, 1A Switch PWM DC/DC Converter
Handles Up to 25V Inputs
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Only 1MHz Integrated Switching Regulator Available
5V at 200mA from 4.4V Input
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20µA I , 36V, 350mA Switch
Q
LT1949
1.1A, 0.5Ω, 30V Internal Switch, V as Low as 1.5V
IN
1373fb LT/TP 0200 2K REV B • PRINTED IN THE USA
LINEAR TECHNOLOGY CORPORATION 1995
12 LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
●
●
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
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