OPA692ID [BB]
Wideband, Fixed Gain Video BUFFER AMPLIFIER With Disable; 宽带固定增益视频缓冲放大器禁用
| 型号： | OPA692ID |
| 厂家： | 
BURR-BROWN CORPORATION |
| 描述： | Wideband, Fixed Gain Video BUFFER AMPLIFIER With Disable |
| 文件： | 总26页 (文件大小：499K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
OPA692
O
P
A
6
9
2
O
P
A
6
9
2
SBOS236C – MARCH 2002 – REVISED JANUARY 2003
Wideband, Fixed Gain
Video BUFFER AMPLIFIER With Disable
FEATURES
● FLEXIBLE SUPPLY RANGE:
+5V to +12V Single Supply
±2.5V to ±6V Dual Supplies
DESCRIPTION
The OPA692 provides an easy to use, broadband fixed gain
video buffer amplifier. Depending on the external connec-
tions, the internal resistor network may be used to provide
either a fixed gain of +2 video buffer or a gain of +1 or –1
voltage buffer. Operating on a very low 5.1mA supply cur-
rent, the OPA692 offers a slew rate and output power
normally associated with a much higher supply current. A
new output stage architecture delivers high output current
with minimal headroom and crossover distortion. This gives
exceptional single-supply operation. Using a single +5V
supply, the OPA692 can deliver a 1V to 4V output swing with
over 120mA drive current and > 200MHz bandwidth. This
combination of features makes the OPA692 an ideal RGB
line driver or single-supply Analog-to-Digital Converter (ADC)
input driver.
● INTERNALLY FIXED GAIN: +2 or ±1
● HIGH BANDWIDTH (G = +2): 225MHz
● LOW SUPPLY CURRENT: 5.1mA
● LOW DISABLED CURRENT: 150µA
● HIGH OUTPUT CURRENT: 190mA
● OUTPUT VOLTAGE SWING: ±4.0V
● SOT23-6 AVAILABLE
APPLICATIONS
● BROADBAND VIDEO LINE DRIVERS
● MULTIPLE LINE VIDEO DA
● PORTABLE INSTRUMENTS
● ADC BUFFERS
The low 5.1mA supply current for the OPA692 is precisely
trimmed at +25°C. This trim, along with low drift over tem-
perature, ensures a lower maximum supply current than
competing products that report only a room temperature
nominal supply current. System power may be further re-
duced by using the optional disable control pin. Leaving this
disable pin open, or holding it HIGH, gives normal operation.
If pulled LOW, the OPA692 supply current drops to less than
150µA while the I/O pins go into a high-impedance state.
● ACTIVE FILTERS
OPA692 RELATED PRODUCTS
SINGLES
OPA690
OPA691
OPA682
DUALS
OPA2690
OPA2691
OPA2682
TRIPLES
OPA3690
OPA3691
OPA3692
Voltage-Feedback
Current-Feedback
Fixed Gain
75Ω
Video
OPA692
Out
RG-59
RG-59
RG-59
RG-59
75Ω
75Ω
75Ω
75Ω
1
2
3
4
8
7
6
5
DIS
75Ω
75Ω
75Ω
+5V
Video
In
–5V
75Ω
SO-8
G = +2
225MHz, 4-Output Component Video DA
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Copyright © 2002-2003, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
www.ti.com
ABSOLUTE MAXIMUM RATINGS(1)
ELECTROSTATIC
DISCHARGE SENSITIVITY
This integrated circuit can be damaged by ESD. Texas Instru-
ments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
Power Supply ............................................................................... ±6.5VDC
Internal Power Dissipation(2) ............................ See Thermal Information
Differential Input Voltage(3) ............................................................... ±1.2V
Input Voltage Range............................................................................ ±VS
Storage Temperature Range: D, DVB ...........................–40°C to +125°C
Lead Temperature (soldering, 10s).............................................. +300°C
Junction Temperature (TJ ) ........................................................... +175°C
ESD Resistance: HBM ........................................................................ 2kV
MM ........................................................................ 200V
ESD damage can range from subtle performance degrada-
tion to complete device failure. Precision integrated circuits
may be more susceptible to damage because very small
parametric changes could cause the device not to meet its
published specifications.
NOTES: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability. These are stress ratings only, and functional operation of the
device at these or any other conditions beyond those specified is not implied.
(2) Packages must be derated based on specified θJA. Maximum TJ must be
observed. (3) Noninverting input to internal inverting node.
PACKAGE/ORDERING INFORMATION
SPECIFIED
PACKAGE
DESIGNATOR(1)
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
PRODUCT
PACKAGE-LEAD
OPA692ID
SO-8 Surface-Mount
D
"
–40°C to +85°C
OPA692
OPA692ID
OPA692IDR
Rails, 100
"
"
"
"
OAGI
"
Tape and Reel, 2500
Tape and Reel, 250
Tape and Reel, 3000
OPA692IDBV
SOT23-6
DBV
–40°C to +85°C
OPA692IDBVT
OPA692IDBVR
"
"
"
"
NOTES: (1) For the most current specifications and package information, refer to our web site at www.ti.com.
PIN CONFIGURATION
Top View
SO
Top View
SOT
Output
–VS
1
2
3
6
5
4
+VS
RF
402Ω
DIS
R
F
NC
–IN
+IN
–VS
1
2
3
4
8
7
6
5
DIS
RG
402Ω
R
G
402Ω
402Ω
+IN
–IN
+VS
Output
NC
6
5
4
NC: No Connection
OAGI
1
2
3
Pin Orientation/Package Marking
OPA692
2
SBOS236C
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = ±5V
Boldface limits are tested at +25°C.
G = +2 (–IN grounded) and RL = 100Ω (see Figure 1 for AC performance only), unless otherwise noted.
OPA692ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
0
°
C to
–40
+85°C
°
C to
MIN/
TEST
MAX LEVEL(2 )
PARAMETER
CONDITIONS
+25°C
+25°C(1)
70°C
UNITS
AC PERFORMANCE (see Figure 1)
Small-Signal Bandwidth (VO < 0.5Vp-p)
G = +1
G = +2
280
225
220
120
0.2
220
2000
1.6
1.9
12
MHz
MHz
MHz
MHz
dB
typ
min
typ
C
B
C
B
B
C
B
C
C
C
C
185
180
170
G = –1
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +1
Large-Signal Bandwidth
Slew Rate
G = +2, VO < 0.5Vp-p
VO < 0.5Vp-p
40
1
35
30
2
min
max
typ
1.5
G = +2, VO = 5Vp-p
G = +2, 4V Step
G = +2, VO = 0.5V Step
G = +2, VO = 5V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, f = 5MHz, VO = 2Vp-p
RL = 100Ω
MHz
V/µs
ns
1400
1375
1350
min
typ
Rise-and-Fall Time
ns
typ
Settling Time to 0.02%
0.1%
ns
typ
8
ns
typ
Harmonic Distortion
2nd-Harmonic
–69
–79
–76
–94
1.7
–62
–70
–72
–87
2.5
14
–59
–67
–70
–82
2.9
15
–57
–65
–68
–78
3.1
15
dBc
dBc
max
max
max
max
max
max
max
typ
B
B
B
B
B
B
B
C
C
C
C
RL ≥ 500Ω
3rd-Harmonic
RL = 100Ω
dBc
RL ≥ 500Ω
dBc
Input Voltage Noise
f > 1MHz
nV/√Hz
pA/√Hz
pA/√Hz
%
Noninverting Input Current Noise
Inverting Input Current Noise
Differential Gain
f > 1MHz
12
f > 1MHz
15
17
18
19
NTSC, RL = 150Ω
NTSC, RL = 37.5Ω
NTSC, RL = 150Ω
NTSC, RL = 37.5Ω
0.07
0.17
0.02
0.07
%
typ
Differential Phase
deg
typ
deg
typ
DC PERFORMANCE(3)
Gain Error
G = +1
G = +2
G = –1
±0.2
±0.3
±0.2
%
%
%
typ
C
A
B
±1.5
±1.5
±1.6
±1.6
±1.7
±1.7
max
max
Internal RF and RG
Maximum
402
402
457
347
462
342
0.13
±3.2
±12
+43
–300
±30
±90
464
340
Ω
Ω
max
min
A
A
B
A
B
A
B
A
B
Minimum
Average Drift
0.13
±2.5
0.13
±3.9
±20
%/C°
mV
max
max
max
max
max
max
max
Input Offset Voltage
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
±0.5
+15
±5
Average Offset Voltage Drift
Noninverting Input Bias Current
Average Noninverting Input Bias Current Drift
Inverting Input Bias Current
Average Inverting Input Bias Current Drift
µV/°C
µA
+35
+45
–300
±40
nA/°C
µA
±25
±200
nA°C
INPUT
Common-Mode Input Range
Noninverting Input Impedance
±3.5
±3.4
±3.3
±3.2
V
min
typ
B
C
100 || 2
kΩ || pF
OUTPUT
Voltage Output Swing
No Load
±4.0
±3.9
+190
–190
±250
0.12
±3.8
±3.7
+160
–160
±3.7
±3.6
+140
–140
±3.6
±3.3
+100
–100
V
V
min
min
min
min
typ
A
A
A
A
C
C
100Ω Load
Current Output, Sourcing
Sinking
mA
mA
mA
Ω
Short-Circuit Current
Closed-Loop Output Impedance
VO = 0
G = +2, f = 100kHz
typ
NOTES: (1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C
at high temperature limit specifications. (2) Test Levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by
characterization and simulation. (C) Typical value only for information. (3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
OPA692
SBOS236C
3
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = ±5V (Cont.)
Boldface limits are tested at +25°C.
G = +2 (–IN grounded) and RL = 100Ω (see Figure 1 for AC performance only), unless otherwise noted.
OPA692ID, IDBV
MIN/MAX OVER TEMPERATURE
C to –40 C to
TYP
0°
°
MIN/
TEST
MAX LEVEL(2 )
PARAMETER
CONDITIONS
+25°C
+25°C(1)
70°C
+85°C
UNITS
DISABLE/POWER DOWN (DIS Pin)
Power-Down Supply Current (+VS)
Disable Time
VDIS = 0
–150
1
–300
–350
–400
µA
µs
ns
dB
pF
mV
mV
V
max
typ
A
C
C
C
C
C
C
A
A
A
VIN = +1VDC
VIN = +1VDC
G = +2, 5MHz
Enable Time
25
typ
Off Isolation
70
typ
Output Capacitance in Disable
Turn-On Glitch
4
typ
G = +2, RL = 150Ω
G = +2, RL= 150Ω
±50
±20
3.3
1.8
75
typ
Turn-Off Glitch
typ
Enable Voltage
3.5
1.7
130
3.6
1.6
150
3.7
1.5
160
min
max
max
Disable Voltage
V
Control Pin Input Bias Current
VDIS = 0
µA
POWER SUPPLY
Specified Operating Voltage
Maximum Operating Voltage Range
Maximum Quiescent Current
Minimum Quiescent Current
Power-Supply Rejection Ratio (–PSRR)
±5
V
typ
max
max
min
min
C
A
A
A
A
±6
5.3
4.9
52
±6
5.5
4.5
50
±6
5.8
4.25
49
V
VS = ±5V
VS = ±5V
5.1
5.1
58
mA
mA
dB
Input Referred
TEMPERATURE RANGE
Specification: D, DBV
–40 to +85
°C
typ
C
Thermal Resistance, θJA
D
SO-8
125
150
°C/W
°C/W
typ
typ
C
C
DBV SOT23-6
NOTES: (1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C
at high temperature limit specifications. (2) Test Levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by
characterization and simulation. (C) Typical value only for information. (3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
OPA692
4
SBOS236C
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = +5V
Boldface limits are tested at +25°C.
G = +2 (–IN grounded though 0.1µF) and RL = 100Ω to VS/2 (see Figure 2 for AC performance only), unless otherwise noted.
OPA692ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
0
°
C to –40 C to
°
MIN/
TEST
MAX LEVEL(2 )
PARAMETER
CONDITIONS
+25°C
+25°C(1)
70°C
+85°C
UNITS
AC PERFORMANCE (see Figure 2)
Small-Signal Bandwidth (VO < 0.5Vp-p)
G = +1
G = +2
240
190
195
90
MHz
MHz
MHz
MHz
dB
typ
min
typ
C
B
C
B
B
C
B
C
C
C
C
168
160
140
G = –1
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +1
Large-Signal Bandwidth
Slew Rate
G = +2, VO < 0.5Vp-p
VO < 0.5Vp-p
40
1
30
25
3
min
max
typ
0.2
210
830
2.0
2.3
14
2.5
G = +2, VO = 2Vp-p
G = +2, 2V Step
G = +2, VO = 0.5V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, f = 5MHz, VO = 2Vp-p
RL = 100Ω to VS/2
RL ≥ 500Ω to VS/2
RL = 100Ω to VS/2
RL ≥ 500Ω to VS /2
f > 1MHz
MHz
V/µs
ns
600
575
550
min
typ
Rise-and-Fall Time
ns
typ
Settling Time to 0.02%
0.1%
ns
typ
10
ns
typ
Harmonic Distortion
2nd-Harmonic
–66
–73
–72
–77
1.7
12
–58
–65
–68
–72
2.5
14
–57
–63
–67
–70
2.9
15
–56
–62
–65
–69
3.1
15
dBc
dBc
max
max
max
max
max
max
max
B
B
B
B
B
B
B
3rd-Harmonic
dBc
dBc
Input Voltage Noise
nV/√Hz
pA/√Hz
pA/√Hz
Noninverting Input Current Noise
Inverting Input Current Noise
f > 1MHz
f > 1MHz
15
17
18
19
DC PERFORMANCE(3)
Gain Error
G = +1
G = +2
G = –1
±0.2
±0.3
±0.2
%
%
%
typ
C
A
B
±1.5
±1.5
±1.6
±1.6
±1.7
±1.7
max
max
Internal RF and RG
Maximum
402
402
457
347
0.13
±3
462
342
464
340
Ω
Ω
max
min
B
B
B
A
B
A
B
A
B
Minimum
Average Drift
0.13
±3.6
±12
0.13
±4.3
±20
%/C°
mV
max
max
max
max
max
max
max
Input Offset Voltage
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
VCM = 2.5V
±0.5
+20
±5
Average Offset Voltage Drift
Noninverting Input Bias Current
Average Noninverting Input Bias Current Drift
Inverting Input Bias Current
Average Inverting Input Bias Current Drift
µV/°C
µA
+40
+46
+56
–250
±30
–250
±40
nA/°C
µA
±25
±112
±200
nA°C
INPUT
Least Positive Input Voltage
Most Positive Input Voltage
Noninverting Input Impedance
1.5
3.5
1.6
3.4
1.7
3.3
1.8
3.2
V
V
max
min
typ
B
B
C
100 || 2
kΩ || pF
OUTPUT
Most Positive Output Voltage
No Load
RL = 100Ω
No Load
4.0
3.9
3.8
3.7
3.7
3.6
3.5
3.4
1.5
1.6
+80
–80
V
V
min
min
max
max
min
min
typ
A
A
A
A
A
A
C
Least Positive Output Voltage
1.0
1.2
1.3
V
RL = 100Ω
1.1
1.3
1.4
V
Current Output, Sourcing
Sinking
+160
–160
±250
+120
–120
+100
–100
mA
mA
mA
Short-Circuit Current
VO = VS/2
Output Impedance
G = +2, f = 100kHz
0.12
Ω
typ
C
NOTES: (1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C
at high temperature limit specifications. (2) Test Levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by
characterization and simulation. (C) Typical value only for information. (3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
OPA692
SBOS236C
5
www.ti.com
ELECTRICAL CHARACTERISTICS: VS = +5V (Cont.)
Boldface limits are tested at +25°C.
G = +2 (–IN grounded though 0.1µF) and RL = 100Ω to VS/2 (see Figure 2 for AC performance only), unless otherwise noted.
OPA692ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
0
°
C to –40 C to
°
MIN/
TEST
MAX LEVEL(2 )
PARAMETER
CONDITIONS
+25°C(1)
+25°C
–300
70°C
+85°C
UNITS
DISABLE/POWER DOWN (DIS Pin)
Power-Down Supply Current (+VS)
Off Isolation
VDIS = 0
–150
65
–350
–400
µA
dB
pF
mV
mV
V
typ
typ
C
C
C
B
B
B
B
C
G = +2, 5MHz
Output Capacitance in Disable
Turn-On Glitch
4
typ
G = +2, RL = 150Ω, VIN = 2.5V
G = +2, RL = 150Ω, VIN = 2.5V
±50
±20
3.3
1.8
75
typ
Turn-Off Glitch
typ
Enable Voltage
3.5
1.7
130
3.6
1.6
150
3.7
1.5
160
min
max
typ
Disable Voltage
V
Control Pin Input Bias Current (DIS
)
VDIS = 0
µA
POWER SUPPLY
Specified Single-Supply Operating Voltage
Maximum Single-Supply Operating Voltage
Maximum Quiescent Current
5
V
V
typ
max
max
min
typ
C
A
A
A
C
12
4.8
4.1
12
5.0
3.8
12
5.2
3.7
VS = +5V
VS = +5V
4.5
4.5
55
mA
mA
dB
Minimum Quiescent Current
Power-Supply Rejection Ratio (+PSRR)
Input Referred
TEMPERATURE RANGE
Specification: D, DBV
–40 to +85
°C
typ
C
Thermal Resistance, θJA
D
SO-8
125
150
°C/W
°C/W
typ
typ
C
C
DBV SOT23-6
NOTES: (1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C
at high temperature limit specifications. (2) Test Levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by
characterization and simulation. (C) Typical value only for information. (3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
OPA692
6
SBOS236C
www.ti.com
TYPICAL CHARACTERISTICS: VS = ±5V
TA = +25°C, G = +2, and RL = 100Ω (see Figure 1 for DC performance only), unless otherwise noted.
SMALL-SIGNAL FREQUENCY RESPONSE
LARGE-SIGNAL FREQUENCY RESPONSE
1
0
7
6
5
4
3
2
1
0
VO = 1Vp-p
VO = 2Vp-p
G = +1
G = –1
–1
–2
–3
–4
–5
–6
–7
–8
VO = 4Vp-p
VO = 7Vp-p
G = +2
0
250MHz
500MHz
0
125MHz
250MHz
Frequency (50MHz/div)
Frequency (25MHz/div)
SMALL-SIGNAL PULSE RESPONSE
LARGE-SIGNAL PULSE RESPONSE
4
3
400
300
VO = 5Vp-p
G = +2
VO = 0.5Vp-p
G = +2
2
200
1
100
0
0
–1
–2
–3
–4
–100
–200
–300
–400
Time (5ns/div)
Time (5ns/div)
COMPOSITE VIDEO dG/dP
DISABLED FEEDTHROUGH vs FREQUENCY
VDIS = 0
0.20
0.18
0.16
0.14
0.12
0.10
0.08
0.06
0.04
0.02
0
–50
–55
–60
–65
–70
–75
–80
–85
–90
–95
+5V
DIS
No Pull-Down
With 1.3kΩ Pull-Down
Video In
Video Loads
OPA692
dG
Optional
1.3kΩ
Pull-Down
–5V
dG
Reverse
Forward
dP
dP
1
2
3
4
0.5
1
10
100
Frequency (MHz)
Number of 150Ω Loads
OPA692
SBOS236C
7
www.ti.com
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +2, and RL = 100Ω (see Figure 1 for DC performance only), unless otherwise noted.
HARMONIC DISTORTION vs LOAD RESISTANCE
5MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE
–60
–65
–50
–55
–60
–65
–70
–75
–80
–85
–90
VO = 2Vp-p
f = 5MHz
VO = 2Vp-p
RL = 100Ω
f = 5MHz
–70
2nd-Harmonic
3rd-Harmonic
–75
2nd-Harmonic
–80
–85
–90
–95
–100
–105
–110
3rd-Harmonic
100
1000
2
2.5
3
3.5
4
4.5
5
5.5
6
Load Resistance (Ω)
Supply Voltage (±VS)
HARMONIC DISTORTION vs OUTPUT VOLTAGE
HARMONIC DISTORTION vs FREQUENCY (G = +2)
VO = 2Vp-p
–65
–70
–75
–80
–85
–50
–60
RL = 100Ω
f = 5MHz
dBc = dB Below Carrier
2nd-Harmonic
RL = 100Ω
2nd-Harmonic
–70
–80
3rd-Harmonic
3rd-Harmonic
–90
–100
0.1
1
5
0.1
1
10
20
Output Voltage Swing (Vp-p)
Frequency (MHz)
HARMONIC DISTORTION vs FREQUENCY (G = –1)
HARMONIC DISTORTION vs FREQUENCY (G = +1)
VO = 2Vp-p
–50
–60
–50
–60
VO = 2Vp-p
dBc = dB Below Carrier
dBc = dB Below Carrier
RL = 100Ω
RL = 100Ω
2nd-Harmonic
–70
–70
3rd-Harmonic
–80
–80
3rd-Harmonic
–90
–90
2nd-Harmonic
–100
–100
0.1
1
10
20
0.1
1
10
20
Frequency (MHz)
Frequency (MHz)
OPA692
8
SBOS236C
www.ti.com
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +2, and RL = 100Ω (see Figure 1 for DC performance only), unless otherwise noted.
2-TONE, 3RD-ORDER
INPUT VOLTAGE AND CURRENT NOISE DENSITY
INTERMODULATION SPURIOUS
100
10
1
–30
–40
–50
–60
–70
–80
–90
dBc = dB below carriers
50MHz
Inverting Input Current Noise (15pA/√Hz)
Noninverting Current Noise (12pA/√Hz)
Voltage Noise (1.7nV/√Hz)
20MHz
10MHz
Load Power at Matched 50Ω Load
100
1k
10k
100k
1M
10M
–8
–6
–4
–2
0
2
4
6
8
10
Frequency (Hz)
Single-Tone Load Power (dBm)
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
60
50
40
30
20
10
0
9
6
CL = 10pF
3
CL = 47pF
0
CL = 22pF
–3
–6
–9
VIN
CL = 100pF
RS
VO
OPA692
402Ω
CL
1kΩ
402Ω
1kΩ is optional.
1
10
100
1k
0
125MHz
Frequency (25MHz/div)
250MHz
Capacitive Load (pF)
PSRR vs FREQUENCY
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
65
60
55
50
45
40
35
30
25
20
10
8
250
200
150
100
50
Sourcing Output Current
Sinking Output Current
+PSRR
–PSRR
6
4
Quiescent Supply Current
2
0
0
1k
10k
100k
1M
10M
100M
–50
–25
0
25
50
75
100
125
Frequency (Hz)
Ambient Temperature (°C)
OPA692
SBOS236C
9
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TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +2, and RL = 100Ω (see Figure 1 for DC performance only), unless otherwise noted.
TYPICAL DC DRIFT OVER TEMPERATURE
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
2
1.5
1
40
5
4
Output Current Limited
1W Internal
Power Limit
Noninverting Input Bias Current
30
3
20
2
0.5
0
10
1
Inverting Input Bias Current
25Ω
Load Line
0
0
–1
–2
–3
–4
–5
–0.5
–1
–10
–20
–30
–40
50Ω Load Line
100Ω Load Line
1W Internal
Input Offset Voltage
–1.5
–2
Power Limit
Output Current Limit
–50
–25
0
25
50
75
100
125
–300 –250 –200 –150 –100 –50
0
50 100 150 200 250 300
Ambient Temperature (°C)
IO (mA)
LARGE-SIGNAL DISABLE/ENABLE RESPONSE
VDIS
DISABLE/ENABLE GLITCH
VDIS
6.0
4.0
2.0
0
6.0
4.0
2.0
0
2.0
1.6
1.2
0.8
0.4
0
30
20
Output Voltage
(0V Input)
Output Voltage
10
0
–10
–20
VIN = +1V
Time (200ns/div)
Time (20ns/div)
CLOSED-LOOP OUTPUT IMPEDANCE
10
+5V
OPA692
50Ω
ZO
–5V
402Ω
1
402Ω
0.1
10k
100k
1M
10M
100M
Frequency (Hz)
OPA692
10
SBOS236C
www.ti.com
TYPICAL CHARACTERISTICS: VS = +5V
TA = +25°C, G = +2, and RL = 100Ω (see Figure 2 for AC performance only), unless otherwise noted.
SMALL-SIGNAL FREQUENCY RESPONSE
LARGE-SIGNAL FREQUENCY RESPONSE
VO = 0.5Vp-p
1
0
7
6
5
4
3
2
1
0
G = +1
–1
–2
–3
–4
–5
–6
–7
–8
VO = 1Vp-p
G = –1
VO = 2Vp-p
G = +2
RL = 100Ω to 2.5V
0
250M
500M
0
125M
Frequency (Hz)
250M
Frequency (Hz)
SMALL-SIGNAL PULSE RESPONSE
LARGE-SIGNAL PULSE RESPONSE
2.9
2.8
2.7
2.6
2.5
2.4
2.3
2.2
2.1
4.1
3.7
3.3
2.9
2.5
2.1
1.7
1.3
0.9
G = +2
O = 0.5Vp-p
G = +2
VO = 2Vp-p
V
Time (5ns/div)
Time (5ns/div)
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
CL = 10pF
70
60
50
40
30
20
10
0
9
6
CL = 22pF
3
CL = 47pF
0
+5V
–3
–6
–9
806Ω
806Ω
0.1µF
57.6Ω
V
IN
R
CL = 100pF
S
V
O
OPA692
C
1kΩ
L
402Ω
402Ω
0.1µF
(1kΩ is optional)
1
10
100
1k
0
125MHz
Frequency (25MHz/div)
250MHz
Capacitive Load (pF)
OPA692
SBOS236C
11
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TYPICAL CHARACTERISTICS: VS = +5V (Cont.)
TA = +25°C, G = +2, and RL = 100Ω (see Figure 2 for AC performance only), unless otherwise noted.
HARMONIC DISTORTION vs FREQUENCY
HARMONIC DISTORTION vs LOAD RESISTANCE
–60
–65
–70
–75
–80
–50
–60
–70
–80
–90
VO = 2Vp-p
f = 5MHz
VO = 2Vp-p
L = 100Ω to 2.5V
R
2nd-Harmonic
2nd-Harmonic
3rd-Harmonic
3rd-Harmonic
0.1
1
10
20
100
1k
Frequency (MHz)
Load Resistance (Ω)
2-TONE, 3RD-ORDER
INTERMODULATION SPURIOUS
HARMONIC DISTORTION vs OUTPUT VOLTAGE
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
–60
–65
–70
–75
–80
dBc = dB Below Carriers
RL = 100Ω to 2.5V
f = 5MHz
50MHz
2nd-Harmonic
20MHz
10MHz
3rd-Harmonic
Load Power at Matched 50Ω Load
–6 –4 –2
Single-Tone Load Power (dBm)
–14
–12
–10
–8
0
2
0.1
1
2
3
Output Voltage Swing (Vp-p)
OPA692
12
SBOS236C
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Figure 2 shows the AC-coupled, gain of +2, single-supply
circuit configuration used as the basis of the +5V Electrical
and Typical Characteristics. Though not a rail-to-rail design,
the OPA692 requires minimal input and output voltage head-
room compared to other very wideband current-feedback op
amps. It will deliver a 3Vp-p output swing on a single +5V
supply with greater than 150MHz bandwidth. The key re-
quirement of broadband single-supply operation is to main-
tain input and output signal swings within the usable voltage
ranges at both the input and the output. The circuit of Figure
2 establishes an input midpoint bias using a simple resistive
divider from the +5V supply (two 806Ω resistors). The input
signal is then AC-coupled into this midpoint voltage bias. The
input voltage can swing to within 1.5V of either supply pin,
giving a 2Vp-p input signal range centered between the
supply pins. The input impedance matching resistor (57.6Ω)
used for testing is adjusted to give a 50Ω input match when
the parallel combination of the biasing divider network is
included. The gain resistor (RG) is AC-coupled, giving the
circuit a DC gain of +1—which puts the input DC bias voltage
(2.5V) on the output as well. Again, on a single +5V supply,
the output voltage can swing to within 1V of either supply pin
while delivering more than 120mA output current. A demand-
ing 100Ω load to a midpoint bias is used in this characteriza-
tion circuit. The new output stage used in the OPA692 can
deliver large bipolar output currents into this midpoint load
with minimal crossover distortion, as shown by the +5V
supply, 3rd-harmonic distortion typical characteristics.
APPLICATIONS INFORMATION
WIDEBAND BUFFER OPERATION
The OPA692 gives the exceptional AC performance of a
wideband current-feedback op amp with a highly linear, high-
power output stage. It features internal RF and RG resistors
that make it easy to select a gain of +2, +1, or –1 without any
external resistors. Requiring only 5.1mA quiescent current, the
OPA692 will swing to within 1V of either supply rail and deliver
in excess of 160mA at room temperature. This low output
headroom requirement, along with supply voltage indepen-
dent biasing, gives remarkable single (+5V) supply operation.
The OPA692 will deliver greater than 200MHz bandwidth
driving a 2Vp-p output into 100Ω on a single +5V supply.
Previous boosted output stage amplifiers have typically suf-
fered from very poor crossover distortion as the output current
goes through zero. The OPA692 achieves a comparable
power gain with much better linearity. The primary advantage
of a current-feedback op amp over a voltage-feedback op amp
is that AC performance (bandwidth and distortion) is relatively
independent of signal gain.
Figure 1 shows the DC-coupled, gain of +2, dual power-supply
circuit configuration used as the basis of the ±5V Electrical and
Typical Characteristics. For test purposes, the input imped-
ance is set to 50Ω with a resistor to ground and the output
impedance is set to 50Ω with a series output resistor. Voltage
swings reported in the specifications are taken directly at the
input and output pins while load powers (dBm) are defined at
a matched 50Ω load. For the circuit of Figure 1, the total
effective load will be 100Ω || 804Ω = 89Ω. The disable control
line (DIS) is typically left open to ensure normal amplifier
operation. In addition to the usual power-supply decoupling
capacitors to ground, a 0.1µF capacitor can be included
between the two power-supply pins. This optional added
capacitor will typically improve the 2nd-harmonic distortion
performance by 3dB to 6dB.
+VS
+5V
+
0.1µF
6.8µF
50Ω Source
0.1µF
806Ω
806Ω
DIS
VIN
VO 100Ω
57.6Ω
+5V
VS/2
OPA692
DIS
RF
402Ω
+
0.1µF
6.8µF
50Ω Source
RG
402Ω
VIN
50Ω Load
50Ω
50Ω
0.1µF
OPA692
RF
402Ω
FIGURE 2. AC-Coupled, G = +2, Single-Supply Specification
and Test Circuit.
RG
SINGLE-SUPPLY ADC INTERFACE
402Ω
0.1µF
6.8µF
Most modern, high-performance ADCs (such as the Texas
Instruments ADS8xx and ADS9xx series) operate on a single
+5V (or lower) power supply. It has been a considerable
challenge for single-supply op amps to deliver a low-distor-
tion input signal at the ADC input for signal frequencies
+
–5V
FIGURE 1. DC-Coupled, G = +2, Bipolar Supply, Specifica-
tion and Test Circuit.
OPA692
SBOS236C
13
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exceeding 5MHz. The high slew rate, exceptional output
swing, and high linearity of the OPA692 make it an ideal
single-supply ADC driver. Figure 3 shows an example input
interface to a very high performance 10-bit, 60MSPS CMOS
converter.
the output, centering the output voltage swing as well. Tested
performance at a 20MHz analog input frequency and a 60MSPS
clock rate on the converter gives > 58dBc SFDR.
WIDEBAND VIDEO MULTIPLEXING
The OPA692 in the circuit of Figure 3 provides 190MHz
bandwidth operating at a signal gain of +2 with a 2Vp-p output
swing. The noninverting input bias voltage is referenced to the
midpoint of the ADC signal range by dividing off the top and
bottom of the internal ADC reference ladder. With the gain
resistor (RG) AC-coupled, this bias voltage has a gain of +1 to
One common application for video speed amplifiers that
include a disable pin is to wire multiple amplifier outputs
together, then select which one of several possible video
inputs to source onto a single line. This simple wired-OR
video multiplexer can be easily implemented using the
OPA692, as shown in Figure 4.
+5V
+5V
RF
402Ω
RG
ADS826
10-Bit
60MSPS
0.1µF
Clock
402Ω
50Ω
Input
OPA692
2Vp-p
DIS
22pF
Input
1Vp-p
0.1µF
CM
2kΩ
+3.5V
REFT
0.1µF
+2.5V DC Bias
2kΩ
+1.5V
REFB
0.1µF
FIGURE 3. Wideband, AC-Coupled, Single-Supply ADC Driver.
+5V
2kΩ
|VOUT| < 2.6V
VDIS
+5V
Video 1
DIS
OPA692
75Ω
68.1Ω
–5V
402Ω
402Ω
402Ω
402Ω
75Ω Cable
RG-59
VOUT
+5V
68.1Ω
OPA692
Video 2
DIS
75Ω
–5V
2kΩ
FIGURE 4. 2-Channel Video Multiplexer.
OPA692
14
SBOS236C
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Typically, channel switching is performed either on sync or
retrace time in the video signal. The two inputs are approxi-
mately equal at this time. The make-before-break disable
characteristic of the OPA692 ensures that there is always
one amplifier controlling the line when using a wired-OR
circuit (see Figure 4). Since both inputs may be on for a short
period during the transition between channels, the outputs
are combined through the output impedance matching resis-
tors (68.1Ω in this case). When one channel is disabled, its
feedback network forms part of the output impedance and
slightly attenuates the signal in getting out onto the cable.
The matching resistors have been set to get a signal gain of
+1 at the load while providing > 20dB return loss at the load.
4-CHANNEL FREQUENCY CHANNELIZER
The circuit of Figure 5 is a 4-channel multiplexer. In this
circuit the OPA691 provides the drive for all four channels.
Each channel includes a bandpass filter and each bandpass
filter is set for a different frequency band. This allows the
channelizing part of this circuit. The role of the OPA692 is to
provide impedance isolation. This is done through the use of
four matching resistances (59Ω in this case). These match-
ing resistors ensure that the signals will combine during the
transition between channels. They have been used to get a
gain of +1 at the load.
This circuit may be used with a different number of channels.
Its limitation comes from the drive requirement for each
channel, as well as the minimum acceptable return loss.
The video multiplexer connection (see Figure 4) also insures
that the maximum differential voltage across the inputs of the
unselected channel do not exceed the rated ±1.2V maximum
for standard video signal levels. In any case, VOUT must be
< ±2.6Vp-p in order to not exceed the absolute maximum
differential input voltage (±1.2V) on the disabled channel.
The output resistor value (RO) to keep a gain of +1 at the
load, depends on the number of channels. For the OPA692,
Equation 1 gives:
(1)
The Disable Operation section shows the turn-on and turn-off
switching glitches using a grounded input for a single chan-
nel is typically less than ±50mV. Where two outputs are
switched (see Figure 4), the output line is always under the
control of one amplifier or the other due to the make-before-
break disable timing. In this case, the switching glitches for
two 0V inputs drops to < 20mV.
75Ω • n – 2 + 804Ω
(
)
[
]
241200Ω
RO
=
•
1+
–1
2
2
75Ω • n − 2 + 804Ω
(
)
[
]
Where n = number of devices in multiplexer.
+5V
DIS 1
DIS 2
DIS 3
DIS 4
75Ω
RO
59Ω
#1
OPA692
75Ω
–5V
+5V
75Ω
RO
59Ω
#2
OPA692
+5V
75Ω
75Ω Cable
–5V
VOUT
OPA691
+5V
RG-59
75Ω
RO
75Ω Load
59Ω
#3
–5V
OPA692
75Ω
–5V
+5V
75Ω
RO
59Ω
#4
OPA692
75Ω
–5V
G = +2 Stages
FIGURE 5. 4-Channel Frequency Channelizer.
OPA692
SBOS236C
15
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DELAY-EQUALIZED LOW-PASS FILTER
The circuit in Figure 6 realizes a 5th-order Butterworth low-
pass filter with a –3dB bandwidth of 20MHz and group delay
equalization. This filter is based on the KRC active filter
topology using amplifiers with a fixed positive gain ≥ 1.
VIN
200Ω
VOUT
OPA692
+5V
The OPA692 makes a good amplifier for this type of filter. The
first stage is the group delay equalizer, which is based on a
gain of –1. The second stage has a high-Q pole, uses a gain
of +2 for minimum component sensitivity, and also produces a
real pole. The last stage has a low-Q pole, and uses a gain of
+1 for minimum component sensitivity.
80.6kΩ
200Ω
402Ω
402Ω
2.7nF
OPA227
–5V
2.7nF
The component values have been predistorted to compensate
for the op amps parasitic effects. The low-Q pole section was
placed last to minimize noise peaking in the passband, while
maintaining good dynamic range performance.
FIGURE 7. Precision Wideband, Unity-Gain Buffer.
PRECISION VOLTAGE BUFFER
DESIGN-IN TOOLS
The precision buffer in Figure 7 combines the DC precision
and low 1/f noise of the OPA227 with the high-speed perfor-
mance of the OPA692. The 80.6kΩ resistor makes the high-
frequency and low-frequency nominal gains equal. The
OPA692 takes over from the OPA227 at approximately 32kHz.
DEMONSTRATION BOARDS
Two PC boards are available to assist in the initial evaluation
of circuit performance using the OPA692 in its two package
styles. All of these are available free as an unpopulated PC
56pF
402Ω
402Ω
49.9Ω
105Ω
226Ω
VIN
OPA692
220pF
27pF
115Ω
OPA692
402Ω
100pF
402Ω
68pF
95.3Ω
226Ω
VOUT
OPA692
39pF
402Ω
402Ω
(Open)
FIGURE 6. Butterworth LP Filter with Delay Equalization.
OPA692
16
SBOS236C
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board delivered with descriptive documentation. The sum-
mary information for these boards is shown in the table
below.
over-temperature specifications because the output stage
junction temperatures are higher than the minimum specified
operating ambient.
DRIVING CAPACITIVE LOADS
BOARD
PART
NUMBER
LITERATURE
REQUEST
NUMBER
One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADC—including additional
external capacitance which may be recommended to improve
ADC linearity. A high-speed amplifier like the OPA692 can be
very susceptible to decreased stability and frequency re-
sponse peaking when a capacitive load is placed directly on
the output pin. When the amplifier’s open-loop output resis-
tance is considered, this capacitive load introduces an addi-
tional pole in the signal path that can decrease the phase
margin. Several external solutions to this problem have been
suggested. When the primary considerations are frequency
response flatness, pulse response fidelity, and/or distortion,
the simplest and most effective solution is to isolate the
capacitive load from the feedback loop by inserting a series
isolation resistor between the amplifier output and the capaci-
tive load. This does not eliminate the pole from the loop
response, but rather shifts it and adds a zero at a higher
frequency. The additional zero acts to cancel the phase lag
from the capacitive load pole, thus increasing the phase
margin and improving stability.
PRODUCT
PACKAGE
OPA692ID
OPA692IDBV
SO-8
SOT23-6
DEM-OPA68xU
DEM-OPA6xxN
SBOU009
SBOU010
To request any of these boards, check the Texas Instruments
web site at www.ti.com.
OPERATING SUGGESTIONS
GAIN SETTING
Setting the gain with the OPA692 is very easy. For a gain of
+2, ground the –IN pin and drive the +IN pin with the signal.
For a gain of +1, leave the –IN pin open and drive the +IN pin
with the signal. For a gain of –1, ground the +IN pin and drive
the –IN pin with the signal. As the internal resistor values (not
their ratio) change over temperature and process, external
resistors should not be used to modify the gain.
OUTPUT CURRENT AND VOLTAGE
The OPA692 provides output voltage and current capabilities
that are unsurpassed in a low-cost monolithic op amp. Under
no-load conditions at +25°C, the output voltage typically
swings closer than 1V to either supply rail; the tested swing
limit is within 1.2V of either rail. Into a 15Ω load (the minimum
tested load), it is specified to deliver more than ±160mA.
The Typical Characteristics show the recommended “RS vs
Capacitive Load” and the resulting frequency response at the
load. Parasitic capacitive loads greater than 2pF can begin to
degrade the performance of the OPA692. Long PC board
traces, unmatched cables, and connections to multiple de-
vices can easily cause this value to be exceeded. Always
consider this effect carefully, and add the recommended
series resistor as close as possible to the OPA692 output pin
(see the Board Layout Guidelines section).
The specifications described previously, though familiar in
the industry, consider voltage and current limits separately. In
many applications, it is the voltage times current, or V-I
product, which is more relevant to circuit operation. Refer to
the “Output Voltage and Current Limitations” plot in the
Typical Characteristics. The X- and Y-axes of this graph show
the zero-voltage output current limit and the zero-current
output voltage limit, respectively. The four quadrants give a
more detailed view of the OPA692 output drive capabilities,
noting that the graph is bounded by a safe operating area of
1W maximum internal power dissipation. Superimposing
resistor load lines onto the plot shows that the OPA692 can
drive ±2.5V into 25Ω, or ±3.5V into 50Ω without exceeding
the output capabilities or the 1W dissipation limit. A 100Ω
load line (the standard test circuit load) shows the full ±3.9V
output swing capability (see the Electrical Characteristics).
DISTORTION PERFORMANCE
The OPA692 provides good distortion performance into a
100Ω load on ±5V supplies. Relative to alternative solutions, it
provides exceptional performance into lighter loads and/or
operating on a single +5V supply. Generally, until the funda-
mental signal reaches very high-frequency or power levels, the
2nd-harmonic will dominate the distortion with a negligible 3rd-
harmonic component. Focusing then on the 2nd-harmonic,
increasing the load impedance improves distortion directly.
Remember that the total load includes the feedback network—
in the noninverting configuration (see Figure 1) this is the sum
of RF + RG, while in the inverting configuration, it is just RF.
Also, providing an additional supply decoupling capacitor
(0.1µF) between the supply pins (for bipolar operation) im-
proves the 2nd-order distortion slightly (3dB to 6dB).
The minimum specified output voltage and current over
temperature are set by worst-case simulations at the cold
temperature extreme. Only at cold startup will the output
current and voltage decrease to the numbers shown in the
Electrical Characteristics. As the output transistors deliver
power, their junction temperatures increase, decreasing their
In most op amps, increasing the output voltage swing increases
harmonic distortion directly. The Typical Characteristics show the
2nd-harmonic increasing at a little less than the expected 2x rate
while the 3rd-harmonic increases at a much lower rate than the
expected 3x. Where the test power doubles, the difference
between it and the 2nd-harmonic decreases less than the
V
BEs (increasing the available output voltage swing), and
increasing their current gains (increasing the available output
current). In steady-state operation, the available output volt-
age and current is always greater than that shown in the
OPA692
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17
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expected 6dB, while the difference between it and the 3rd
decreases by less than the expected 12dB. This also shows up
in the 2-tone, 3rd-order intermodulation spurious (IM3) response
curves. The 3rd-order spurious levels are extremely low at low
output power levels. The output stage continues to hold them low
even as the fundamental power reaches very high levels. As the
Typical Characteristics show, the spurious intermodulation pow-
ers do not increase as predicted by a traditional intercept model.
As the fundamental power level increases, the dynamic range
does not decrease significantly. For two tones centered at
20MHz, with 10dBm/tone into a matched 50Ω load (i.e., 2Vp-p
for each tone at the load, which requires 8Vp-p for the overall
2-tone envelope at the output pin), the Typical Characteristics
show 58dBc difference between the test-tone power and the 3rd-
order intermodulation spurious levels. This exceptional perfor-
mance improves further when operating at lower frequencies.
square root of the sum of all squared output noise voltage
contributors. Equation 2 shows the general form for the output
noise voltage using the terms shown in Figure 8.
(2)
2
2
2
EO
=
ENI + IBNRS + 4kTRS NG2 + I R
+ 4kTRFNG
(
)
(
)
BI
F
Dividing this expression by the noise gain (NG = (1 + RF/RG))
will give the equivalent input-referred spot noise voltage at the
noninverting input, as shown in Equation 3.
(3)
2
IBIRF
NG
4kTRF
NG
2
2
EN
=
ENI + IBNRS + 4kTRS
+
+
(
)
Evaluating these two equations for the OPA692 circuit and
component values (see Figure 1) will give a total output spot
noise voltage of 8.2nV/√Hz and a total equivalent input spot
noise voltage of 4.1nV/√Hz. This total input-referred spot
noise voltage is higher than the 1.7nV/√Hz specification for
the op amp voltage noise alone. This reflects the noise added
to the output by the inverting current noise times the feed-
back resistor.
NOISE PERFORMANCE
The OPA692 offers an excellent balance between voltage and
current noise terms to achieve low output noise. The inverting
current noise (15pA/√Hz) is significantly lower than earlier
solutions while the input voltage noise (1.7nV√Hz) is lower
than most unity-gain stable, wideband, voltage-feedback op
amps. This low input voltage noise was achieved at the price
of higher noninverting input current noise (12pA/√Hz). As long
as the AC source impedance looking out of the noninverting
node is less than 100Ω, this current noise will not contribute
significantly to the total output noise. The op amp input voltage
noise and the two input current noise terms combine to give
low output noise for the gain settings, available using the
OPA692. Figure 8 shows the op amp noise analysis model
with all the noise terms included. In this model, all noise terms
are taken to be noise voltage or current density terms in either
DC ACCURACY
The OPA692 provides exceptional bandwidth in high gains,
giving fast pulse settling but only moderate DC accuracy. The
Electrical Characteristics show an input offset voltage com-
parable to high-speed voltage-feedback amplifiers. However,
the two input bias currents are somewhat higher and are
unmatched. Bias current cancellation techniques will not
reduce the output DC offset for OPA692. As the two input
bias currents are unrelated in both magnitude and polarity,
matching the source impedance looking out of each input to
reduce their error contribution to the output is ineffective.
Evaluating the configuration of Figure 1, using worst-case
+25°C input offset voltage and the two input bias currents,
gives a worst-case output offset range equal to:
nV/√Hz or pA/√Hz
.
The total output spot noise voltage can be computed as the
±(NG • VOS(max)) + (IBN • RS/2 • NG) ± (IBI • RF)
where NG = noninverting signal gain
= ±(2 • 2.5mV) + (35µA • 25Ω • 2) ± (402Ω • 25µA)
= ±5mV + 1.75mV ± 10.05mV
ENI
EO
OPA692
RS
IBN
= –13.3mV → +16.80mV
Minimizing the resistance seen by the noninverting input will
give the best DC offset performance.
ERS
RF
√4kTRS
For significantly improved DC accuracy, consider the preci-
sion buffer circuit (see Figure 7).
√4kTRF
IBI
RG
4kT
RG
4kT = 1.6E –20J
at 290°K
DISABLE OPERATION
The OPA692 provides an optional disable feature that may be
used either to reduce system power or to implement a simple
channel multiplexing operation. If the DIS control pin is left
unconnected, the OPA692 will operate normally. To disable,
FIGURE 8. Noise Model.
OPA692
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the control pin must be asserted LOW. Figure 9 shows a
simplified internal circuit for the disable control feature.
THERMAL ANALYSIS
Due to the high output power capability of the OPA692,
heatsinking or forced airflow may be required under extreme
operating conditions. Maximum desired junction temperature
will set the maximum allowed internal power dissipation, as
described below. In no case should the maximum junction
temperature be allowed to exceed 175°C.
In normal operation, base current to Q1 is provided through
+VS
Operating junction temperature (TJ) is given by TA + PD • θJA.
The total internal power dissipation (PD) is the sum of
quiescent power (PDQ) and additional power dissipated in the
output stage (PDL) to deliver load power. Quiescent power is
simply the specified no-load supply current times the total
supply voltage across the part. PDL depends on the required
output signal and load but would, for a grounded resistive
load, be at a maximum when the output is fixed at a voltage
equal to 1/2 either supply voltage (for equal bipolar supplies).
15kΩ
Q1
110kΩ
25kΩ
–VS
IS
2
VDIS
Under this condition PDL = VS /(4 • RL), where RL includes
Control
feedback network loading.
Note that it is the power in the output stage and not in the
load that determines internal power dissipation.
FIGURE 9. Simplified Disable Control Circuit.
As a worst-case example, compute the maximum TJ using an
OPA692IDBV (SOT23-6 package) in the circuit of Figure 1
operating at the maximum specified ambient temperature of
the 110kΩ resistor while the emitter current through the 15kΩ
resistor sets up a voltage drop that is inadequate to turn on
the two diodes in Q1’s emitter. As VDIS is pulled LOW,
additional current is pulled through the 15kΩ resistor eventu-
ally turning on these two diodes (≈ 75µA). At this point, any
further current pulled out of VDIS goes through those diodes
holding the emitter-base voltage of Q1 at approximately 0V.
This shuts off the collector current out of Q1, turning the
amplifier off. The supply current in the disable mode is only
that required to operate the circuit of Figure 8. Additional
circuitry ensures that turn-on time occurs faster than turn-off
time (make-before-break).
+85°C and driving a grounded 20Ω load to +2.5VDC
:
PD = 10V • 5.8mA + 52/(4 • (20Ω || 800Ω)) = 378mW
Maximum TJ = +85°C + (0.39W • 150°C/W) = 142°C
Although this is still well below the specified maximum
junction temperature, system reliability considerations may
require lower junction temperatures. Remember, this is a
worst-case internal power dissipation—use your actual sig-
nal and load to compute PDL. The highest possible internal
dissipation occurs if the load requires current to be forced
into the output for positive output voltages or sourced from
the output for negative output voltages. This puts a high
current through a large internal voltage drop in the output
transistors. The “Output Voltage and Current Limitations” plot
shown in the Typical Characteristics include a boundary for
1W maximum internal power dissipation under these condi-
tions.
When disabled, the output and input nodes go to a high-
impedance state. If the OPA692 is operating in a gain of +1,
this will show a very high impedance (4pF || 1MΩ) at the
output and exceptional signal isolation. If operating at a gain
of +2, the total feedback network resistance (RF + RG) will
appear as the impedance looking back into the output, but
the circuit will still show very high forward and reverse
isolation. If configured at a gain of –1, the input and output
will be connected through the feedback network resistance
(RF + RG) giving relatively poor input to output isolation.
BOARD LAYOUT GUIDELINES
Achieving optimum performance with a high-frequency am-
plifier like the OPA692 requires careful attention to board
layout parasitics and external component types. Recommen-
dations that will optimize performance include:
One key parameter in disable operation is the output glitch
when switching in and out of the disabled mode. The Typical
Characteristics show these glitches for the circuit of Figure 1
with the input signal set to 0V. The glitch waveform at the
output pin is plotted along with the DIS pin voltage.
a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the
output pin can cause instability: on the noninverting input, it
can react with the source impedance to cause unintentional
bandlimiting. To reduce unwanted capacitance, a window
around the signal I/O pins should be opened in all of the
ground and power planes around those pins. Otherwise,
ground and power planes should be unbroken elsewhere on
the board.
The transition edge rate (dV/dt) of the DIS control line will
influence this glitch. Slowing this edge can be achieved by
adding a simple RC filter into the VDIS pin from a higher
speed logic line. If extremely fast transition logic is used, a
2kΩ series resistor between the logic gate and the DIS input
pin will provide adequate bandlimiting using just the parasitic
input capacitance on the DIS pin while still ensuring an
adequate logic level swing.
OPA692
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b) Minimize the distance (< 0.25") from the power-supply
pins to high-frequency 0.1µF decoupling capacitors. At
the device pins, the ground and power-plane layout should
not be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power-supply
connections (on pins 4 and 7) should always be decoupled
with these capacitors. An optional supply decoupling capaci-
tor across the two power supplies (for bipolar operation) will
improve 2nd-harmonic distortion performance. Larger (2.2µF
to 6.8µF) decoupling capacitors, effective at lower frequen-
cies, should also be used on the main supply pins. These
may be placed somewhat further from the device and may be
shared among several devices in the same area of the PC
board.
and the input impedance of the destination device; this total
effective impedance should be set to match the trace imped-
ance. The high output voltage and current capability of the
OPA692 allows multiple destination devices to be handled as
separate transmission lines, each with their own series and
shunt terminations. If the 6dB attenuation of a doubly-termi-
nated transmission line is unacceptable, a long trace can be
series-terminated at the source end only. Treat the trace as
a capacitive load in this case and set the series resistor value
as shown in the plot of “RS vs Capacitive Load.” This will not
preserve signal integrity as well as a doubly-terminated line.
If the input impedance of the destination device is low, there
will be some signal attenuation due to the voltage divider
formed by the series output into the terminating impedance.
e) Socketing a high-speed part like the OPA692 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an ex-
tremely troublesome parasitic network which can make it
almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the OPA692
onto the board.
c) Careful selection and placement of external compo-
nents will preserve the high-frequency performance of
the OPA692. Any external resistors should be a very low
reactance type. Surface-mount resistors work best and allow
a tighter overall layout. Metal-film and carbon composition,
axially-leaded resistors can also provide good high-frequency
performance. Again, keep their leads and PC-board trace
length as short as possible. Never use wirewound type
resistors in a high-frequency application. All external compo-
nents should also be placed close to the package.
INPUT AND ESD PROTECTION
The OPA692 is built using a very high-speed complementary
bipolar process. The internal junction breakdown voltages
are relatively low for these very small geometry devices.
These breakdowns are reflected in the Absolute Maximum
Ratings table. All device pins have limited ESD protection
using internal diodes to the power supplies, as shown in
Figure 10.
d) Connections to other wideband devices on the board
may be made with short direct traces or through onboard
transmission lines. For short connections, consider the
trace and the input to the next device as a lumped capacitive
load. Relatively wide traces (50mils to 100mils) should be
used, preferably with ground and power planes opened up
around them. Estimate the total capacitive load and set RS
from the plot of recommended “RS vs Capacitive Load.” Low
parasitic capacitive loads (< 5pF) may not need an RS
because the OPA692 is nominally compensated to operate
with a 2pF parasitic load. If a long trace is required, and the
6dB signal loss intrinsic to a doubly-terminated transmission
line is acceptable, implement a matched impedance trans-
mission line using microstrip or stripline techniques (consult
an ECL design handbook for microstrip and stripline layout
techniques). A 50Ω environment is normally not necessary
on board, and in fact, a higher impedance environment will
improve distortion as shown in the “Distortion vs Load” plots.
With a characteristic board trace impedance defined based
on board material and trace dimensions, a matching series
resistor into the trace from the output of the OPA692 is used
as well as a terminating shunt resistor at the input of the
destination device. Remember also that the terminating im-
pedance will be the parallel combination of the shunt resistor
+VCC
External
Pin
Internal
Circuitry
–VCC
FIGURE 10. Internal ESD Protection.
These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
currents are possible (e.g., in systems with ±15V supply parts
driving into the OPA692), current-limiting series resistors
should be added into the two inputs. Keep these resistor
values as low as possible since high values degrade both
noise performance and frequency response.
OPA692
20
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PACKAGE DRAWINGS
D (R-PDSO-G**)
PLASTIC SMALL-OUTLINE PACKAGE
8 PINS SHOWN
0.020 (0,51)
0.014 (0,35)
0.050 (1,27)
8
0.010 (0,25)
5
0.244 (6,20)
0.228 (5,80)
0.008 (0,20) NOM
0.157 (4,00)
0.150 (3,81)
Gage Plane
1
4
0.010 (0,25)
0°– 8°
A
0.044 (1,12)
0.016 (0,40)
Seating Plane
0.010 (0,25)
0.069 (1,75) MAX
0.004 (0,10)
0.004 (0,10)
PINS **
8
14
16
DIM
A MAX
0.197
(5,00)
0.344
(8,75)
0.394
(10,00)
0.189
(4,80)
0.337
(8,55)
0.386
(9,80)
A MIN
4040047/E 09/01
NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15).
D. Falls within JEDEC MS-012
OPA692
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PACKAGE DRAWINGS (Cont.)
DBV (R-PDSO-G6)
PLASTIC SMALL-OUTLINE
0,50
0,25
M
0,20
0,95
6
6X
4
0,15 NOM
1,70
1,50
3,00
2,60
1
3
Gage Plane
3,00
2,80
0,25
0 –8
0,55
0,35
Seating Plane
0,10
1,45
0,95
0,05 MIN
4073253-5/G 01/02
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Body dimensions do not include mold flash or protrusion.
D. Leads 1, 2, 3 may be wider than leads 4, 5, 6 for package orientation.
OPA692
22
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PACKAGE OPTION ADDENDUM
www.ti.com
1-Feb-2005
PACKAGING INFORMATION
Orderable Device
Status (1)
Package Package
Pins Package Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
Qty
Type
Drawing
OPA692ID
OPA692IDBVR
OPA692IDBVT
OPA692IDR
ACTIVE
ACTIVE
ACTIVE
ACTIVE
SOIC
D
8
6
6
8
100
3000
250
None
None
None
None
CU NIPDAU Level-3-240C-168 HR
CU NIPDAU Level-3-235C-168 HR
CU NIPDAU Level-3-235C-168 HR
CU NIPDAU Level-3-240C-168 HR
SOT-23
SOT-23
SOIC
DBV
DBV
D
2500
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - May not be currently available - please check http://www.ti.com/productcontent for the latest availability information and additional
product content details.
None: Not yet available Lead (Pb-Free).
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens,
including bromine (Br) or antimony (Sb) above 0.1% of total product weight.
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications, and peak solder
temperature.
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Addendum-Page 1
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