A3921KLPTR-T [ALLEGRO]
Full Bridge Based MOSFET Driver,;
| 型号： | A3921KLPTR-T |
| 厂家： | 
ALLEGRO MICROSYSTEMS |
| 描述： | Full Bridge Based MOSFET Driver, 驱动 光电二极管 接口集成电路 驱动器 |
| 文件： | 总21页 (文件大小：406K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
A3921
Automotive Full Bridge MOSFET Driver
Features and Benefits
Description
▪ High current gate drive for N-channel MOSFET full bridge
▪ High-side or low-side PWM switching
▪ Charge pump for low supply voltage operation
▪ Top-off charge pump for 100% PWM
▪ Cross-conduction protection with adjustable dead time
▪ 5.5 to 50 V supply voltage range
The A3921 is a full-bridge controller for use with external
N-channel power MOSFETs and is specifically designed for
automotiveapplicationswithhigh-powerinductiveloads,such
as brush DC motors.
A unique charge pump regulator provides full (>10 V) gate
drive for battery voltages down to 7 V and allows the A3921
to operate with a reduced gate drive, down to 5.5 V.
▪ Integrated 5 V regulator
▪ Diagnostics output
▪ Low current sleep mode
A bootstrap capacitor is used to provide the above-battery
supply voltage required for N-channel MOSFETs.An internal
charge pump for the high-side drive allows DC (100% duty
cycle) operation.
Package: 28-pin TSSOP with exposed
thermal pad (suffix LP)
The full bridge can be driven in fast or slow decay modes using
diode or synchronous rectification. In the slow decay mode,
current recirculation can be through the high-side or the low-
side FETs. The power FETs are protected from shoot-through
by resistor adjustable dead time.
Integrated diagnostics provide indication of undervoltage,
overtemperature, and power bridge faults, and can be
configured to protect the power MOSFETs under most short
circuit conditions.
TheA3921 is supplied in a 28-pin TSSOPpower package with
an exposed thermal pad (suffix LP). This package is lead (Pb)
free, with 100% matte-tin leadframe plating.
Not to scale
Typical Application
VBAT
PWM
A3921
Direction
Fault Flags
3921-DS
A3921
Automotive Full Bridge MOSFET Driver
Selection Guide
Part Number
Packing
A3921KLPTR-T
4000 pieces per reel
Absolute Maximum Ratings*
Characteristic
Symbol
Notes
Rating
–0.3 to 50
–0.3 to 6.5
–0.3 to 7
–4 to 6.5
–0.3 to 6.5
–5 to 55
Units
V
Load Supply Voltage
Logic Inputs and Outputs
V5 Pin
VBB
V
V
LSS Pin
V
VDSTH Pin
V
SA and SB Pins
V
VDRAIN Pin
–5 to 55
V
GHA and GHB Pins
GLA and GLB Pins
CA and CB Pins
Sx to Sx+15
–5 to 16
V
V
–0.3 to Sx+15
–40 to 150
150
V
Operating Temperature Range
Junction Temperature
TA
Range K
ºC
ºC
TJ(max)
Overtemperature event not exceeding
1 s, lifetime duration not exceeding 10 hr;
guaranteed by design characterization
Transient Junction Temperature
TtJ
175
ºC
ºC
Storage Temperature Range
*With respect to GND.
Tstg
–55 to 150
THERMAL CHARACTERISTICS may require derating at maximum conditions
Characteristic
Symbol
Test Conditions*
Value
Units
4-layer PCB based on JEDEC standard
28
ºC/W
RθJA
Package Thermal Resistance
2-layer PCB with 3.8 in.2 of copper area each side
32
2
ºC/W
ºC/W
RθJP
*Additional thermal information available on Allegro website.
Allegro MicroSystems, LLC
115 Northeast Cutoff
2
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
Functional Block Diagram
Battery +
CP
VBB
CP2
CP1
VBAT
Charge
Pump
Regulator
VREG
V5
+5V Reg
CREG
FF1
FF2
VDRAIN
CA
Diagnostics
and Protection
Charge
Pump
Bootstrap
Monitor
VDSTH
CBOOTA
GHA
SA
High
Side
RGHA
RGHB
Low
Side
GLA
LSS
PWMH
PWML
RGLA
RGLB
Control
Logic
Charge
Pump
Bootstrap
Monitor
CB
PHASE
SR
CBOOTB
High
Side
GHB
SB
RESET
Low
Side
GLB
LSS
PAD
RDEAD
GND
Allegro MicroSystems, LLC
115 Northeast Cutoff
3
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
ELECTRICAL CHARACTERISTICS valid at TJ = –40°C to 150°C, VBB = 7 to 50 V, unless noted otherwise
Characteristics
Supply and Reference
Symbol
Test Conditions
Min.
Typ.
Max.
Units
Load Supply Voltage Functional
Operating Range1
VBB
5.5
–
50
V
IBBQ
IBBS
RESET = high, outputs = low, VBB = 12 V
RESET = low, Sleep mode, VBB = 12 V
VBB > 9 V, IREG = 0 to 10 mA
–
10
–
14
10
mA
μA
V
Load Supply Quiescent Current
VREG Output Voltage
–
12.5
12.5
13
13
13.75
13.75
7.5 V < VBB ≤ 9 V, IREG = 0 to 7 mA
V
VREG
2×VBB
– 2.5
6 V < VBB ≤ 7.5 V, IREG = 0 to 7 mA
–
–
V
5.5 V < VBB ≤ 6 V, IREG < 5.5 mA
No load
8.5
4.5
–
9.5
5
–
V
V
V5 Output Voltage
V5(out)
V5(line)
V5(load)
I5M
5.5
40
V5 Line Regulation
V5 Load Regulation
V5 Short-Circuit Current
I5 = –2 mA
15
50
28
0.7
2.2
mV
mV
mA
V
I5 = 0 to –2 mA
VBB = 40 V, V5 = 0 V
ID = 10 mA
–
100
35
–
0.4
1.5
1.0
2.8
Bootstrap Diode Forward Voltage
Bootstrap Diode Resistance
VfBOOT
ID = 100 mA
V
rD(100mA)
=
rD
6
10
20
Ω
(VfBOOT(150mA) – VfBOOT(50mA)) / 100 mA
Bootstrap Diode Current Limit
Top-off Charge Pump Current Limit
High-Side Gate Drive Static Load Resistance
Gate Output Drive
IDBOOT
ITOCPM
RGSH
250
–
500
400
–
750
–
mA
μA
kΩ
250
–
Turn-On Time
tr
tf
CLOAD = 1 nF, 20% to 80%
CLOAD = 1 nF, 80% to 20%
TJ = 25°C, IGHx = –150 mA
TJ = 150°C, IGHx = –150 mA
TJ = 25°C, IGLx = 150 mA
TJ = 150°C, IGLx = 150 mA
–
–
35
20
8
–
–
ns
ns
Ω
Turn-Off Time
6
12
16
4
Pullup On Resistance
RDS(on)UP
10
2
13
3
Ω
Ω
Pulldown On Resistance
RDS(on)DN
3
4.5
6
Ω
VCx
– 0.2
GHx Output Voltage
VGHX
VGLX
tP(off)
tP(on)
Bootstrap capacitor fully charged
–
–
–
V
V
VREG
– 0.2
GLx Output Voltage
–
Input change to unloaded gate output
change
Turn-Off Propagation Delay2
Turn-On Propagation Delay2
60
60
90
90
150
150
ns
ns
Input change to unloaded gate output
change
Measured between corresponding
transition points on both phases
Propagation Delay Matching, Phase-to-Phase
Propagation Delay Matching, On-to-Off
∆tPP
∆tOO
–
–
10
10
–
–
ns
ns
Measured across one phase
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
4
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
ELECTRICAL CHARACTERISTICS (continued) valid at TJ = –40°C to 150°C, VBB = 7 to 50 V, unless noted otherwise
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Units
Gate Output Drive (continued)
RDEAD = 3 kΩ
RDEAD = 30 kΩ
DEAD = 240 kΩ
–
815
–
180
960
3.5
6
–
1110
–
ns
ns
μs
μs
Dead Time2
tDEAD
R
RDEAD tied to V5
–
–
Logic Inputs and Outputs
FF1 and FF2 Fault Output (Open Drain)
FF1 and FF2 Fault Output Leakage Current3
RDEAD Current3
VFF(L)
IFF(H)
IDEAD
VIN(L)
VIN(H)
VINhys
VINRSThys
IIN
IFF = 1 mA, fault not present
VFF = 5 V, fault present
RDEAD = GND
–
–1
–
–
0.4
1
V
μA
μA
V
–200
–
–
–70
0.8
–
Input Low Voltage
–
Input High Voltage
2.0
100
200
–1
–
V
Input Hysteresis (Except RESET Pin)
Input Hysteresis (RESET Pin)
Input Current (Except RESET Pin)3
Input Pulldown Resistor (RESET Pin)
RESET Pulse Time
250
–
–
mV
mV
μA
kΩ
μs
–
0 V < VIN < V5
–
1
RPD
–
50
–
–
tRES
0.1
3.5
Protection
VREGUVon
VREGUVoff
VBOOTUV
VBOOTUVhys
V5UVoff
VREG rising
7.5
6.75
59
8
7.25
–
8.5
7.75
69
–
V
VREG Undervoltage Lockout Threshold
VREG falling
V
%VREG
%VREG
V
Bootstrap Undervoltage Threshold
Bootstrap Undervoltage Hysteresis
V5 Undervoltage Turn-Off Threshold
V5 Undervoltage Hysteresis
VDSTH Input Range
Cx with respect to Sx
–
13
3.6
400
–
V5 falling
3.4
300
0.1
–
3.8
500
2
V5UVhys
VDSTH
mV
V
VDSTH Input Current
IDSTH
0 V < VDSTH < 2 V
10
–
30
–
μA
VDSTH Disable Voltage
VDSDIS
When not connected directly to V5
VDSTH = 2 V, VBB = 12 V,
4.95
7
V
VDRAIN Input Voltage
VDRAIN
VBB
50
V
VDSTH = 2 V, VBB = 12 V,
0 V < VDRAIN < VBB
VDRAIN Input Current
IDRAIN
–
–
250
μA
High-side on, VDSTH ≥ 1 V
High-side on, VDSTH < 1 V
Low-side on, VDSTH ≥ 1 V
Low-side on, VDSTH < 1 V
Temperature increasing
Recovery = TJF – TJFhys
–
–150
–
±100
±50
±100
±50
170
15
–
150
–
mV
mV
mV
mV
ºC
Short-to-Ground Threshold Offset4
VSTGO
Short-to-Battery Threshold Offset5
VSTBO
–150
150
–
150
–
Overtemperature Fault Flag Threshold
Overtemperature Fault Hysteresis
TJF
TJFhys
–
ºC
1Functions correctly, but parameters are not guaranteed, below the general limits (7 V).
2See Gate Drive Timing diagrams.
3For input and output current specifications, negative current is defined as coming out of (sourcing) the specified device pin.
4As VSx decreases, fault occurs if VBAT –VSx > VSTG. STG threshold, VSTG = VDSTH + VSTGO
5As VSx increases, fault occurs if VSx – VLSS > VSTB. STB threshold, VSTB = VDSTH+VSTBO
.
.
Allegro MicroSystems, LLC
115 Northeast Cutoff
5
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
Timing Diagrams
Gate Drive Timing – PWM inputs, Slow Decay, Synchronous Rectification
PWMH
PWML
tP(off)
tP(off)
tP(off)
tP(off)
GHx
GLx
GHx
GLx
tDEAD
tDEAD
tDEAD
tDEAD
SR=1, PWML=1
SR=1, PWMH=1
Gate Drive Timing – PWM inputs, Slow Decay, Diode Rectification
PWMH
PWML
tP(off)
tP(on)
tP(off)
tP(on)
GHx
GLx
GHx
GLx
SR=0, PWML=1
SR=0, PWMH=1
Gate Drive Timing – Phase Input, Fast Decay,
Synchronous Rectification
Gate Drive Timing – PWM Input, Fast Decay,
Diode Rectification
PHASE
PWMH
PWML
tP(off)
tP(off)
tDEAD
tDEAD
tP(off)
tP(on)
GHA
GLA
GHB
GHA
GLA
GHB
GLB
GLB
PWML=1, PWMH=1
PHASE=1
Allegro MicroSystems, LLC
115 Northeast Cutoff
6
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
Functional Description
The A3921 is a full-bridge MOSFET driver (pre-driver) requiring used by the internal logic circuits and must always be decoupled
a single unregulated supply of 7 to 50 V. It includes an integrated
5 V logic supply regulator.
by at least a 100 nF capacitor between the V5 pin and GND. The
5 V regulator is disabled when RESET is held low.
The four high current gate drives are capable of driving a wide
range of N-channel power MOSFETs, and are configured as two
high-side drives and two low-side drives. The A3921 provides
all the necessary circuits to ensure that the gate-source voltage
of both high-side and low-side external FETs are above 10 V, at
supply voltages down to 7 V. For extreme battery voltage drop
conditions, correct functional operation is guaranteed at supply
voltages down to 5.5 V, but with a reduced gate drive voltage.
Gate Drives
The A3921 is designed to drive external, low on-resistance,
power N-channel MOSFETs. It supplies the large transient cur-
rents necessary to quickly charge and discharge the external FET
gate capacitance in order to reduce dissipation in the external
FET during switching. The charge and discharge rate can be
controlled using an external resistor in series with the connection
to the gate of the FET.
The A3921 can be driven with a single PWM input from a
microcontroller and can be configured for fast or slow decay.
Fast decay can provide four-quadrant motor control, while slow
decay is suitable for two-quadrant motor control or simple induc-
tive loads. In slow decay, current recirculation can be through
the high-side or the low-side MOSFETs. In either case, bridge
efficiency can be enhanced by synchronous rectification. Cross-
conduction (shoot through) in the external bridge is avoided by
an adjustable dead time.
Gate Drive Voltage Regulation The gate drives are powered by
an internal regulator which limits the supply to the drives and
therefore the maximum gate voltage. When the VBB supply is
greater than about 16 V, the regulator is a simple linear regulator.
Below 16 V, the regulated supply is maintained by a charge pump
boost converter, which requires a pump capacitor connected
between the CP1 and CP2 pins. This capacitor must have a mini-
mum value of 220 nF, and is typically 470 nF.
A low power sleep mode allows the A3921, the power bridge, and
the load to remain connected to a vehicle battery supply without
the need for an additional supply switch.
The regulated voltage, nominally 13 V, is available on the VREG
pin. A sufficiently large storage capacitor must be connected to
this pin to provide the transient charging current to the low-side
drives and the bootstrap capacitors.
The A3921 includes a number of protection features against
undervoltage, overtemperature, and power bridge faults. Fault
states enable responses by the device or by the external control-
ler, depending on the fault condition and logic settings. Two fault
flag outputs, FF1 and FF2, are provided to signal detected faults
to an external controller.
Top-off Charge Pump An additional top-off charge pump is
provided for each phase. The charge pumps allow the high-side
drives to maintain the gate voltage on the external FETs indefi-
nitely, ensuring so-called 100% PWM if required. This is a low
current trickle charge pump, and is operated only after a high-side
FET has been signaled to turn on. The floating high-side gate
drive requires a small bias current (<20 μA) to maintain the high-
level output. Without the top-off charge pump, this bias current
would be drawn from the bootstrap capacitor through the Cx pin.
The charge pump provides sufficient current to ensure that the
bootstrap voltage and thereby the gate-source voltage is main-
tained at the necessary level.
Power Supplies
A single power supply connection is required to the VBB pin
through a reverse voltage protection circuit. The supply should be
decoupled with a ceramic capacitor connected close to the VBB
and ground pins.
The A3921 operates within specified parameters with a VBB
supply from 7 to 50 V and functions correctly with a supply down
to 5.5 V. This provides a very rugged solution for use in the harsh
automotive environment.
Note that the charge required for initial turn-on of the high-side
gate is always supplied by the bootstrap capacitor. If the bootstrap
capacitor becomes discharged, the top-off charge pump will not
provide sufficient current to allow the FET to turn on.
V5 Pin A 5 V low current supply for external pullup resistors is
provided by an integrated 5 V regulator. This regulator is also
Allegro MicroSystems, LLC
115 Northeast Cutoff
7
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
In some applications a safety resistor is added between the gate
and source of each FET in the bridge. When a high-side FET is
held in the on-state, the current through the associated high-side
LSS Pin This is the low-side return path for discharge of the
capacitance on the FET gates. It should be tied directly to the
common sources of the low-side external FETs through an inde-
gate-source resistor (RGSH) is provided by the high-side drive and pendent low impedance connection.
therefore appears as a static resistive load on the top-off charge
RDEAD Pin This pin controls internal generation of dead time
during FET switching.
pump. The minimum value of RGSH for which the top-off charge
pump can provide current is shown in the Electrical Characteris-
tics table.
• When a resistor greater than 3 kΩ is connected between
RDEAD and AGND, cross-conduction is prevented by the gate
drive circuits, which introduce a dead time, tDEAD , between
switching one FET off and the complementary FET on. The
dead time is derived from the resistor value connected between
the RDEAD and AGND pins.
• When RDEAD is connected directly to V5, cross-conduction is
prevented by the gate drive circuits. In this case, tDEAD defaults
to a value of 6 μs typical.
GLA and GLB Pins These are the low-side gate drive outputs for
the external N-channel MOSFETs. External resistors between the
gate drive output and the gate connection to the FET (as close as
possible to the FET) can be used to control the slew rate seen at
the gate, thereby providing some control of the di/dt and dv/dt of
the SA and SB outputs. GLx going high turns on the upper half of
the drive, sourcing current to the gate of the low-side FET in the
external power bridge, turning it on. GLx going low turns on the
lower half of the drive, sinking current from the external FET gate
circuit to the LSS pin, turning off the FET.
Logic Control Inputs
SA and SB Pins Directly connected to the motor, these terminals
sense the voltages switched across the load. These terminals are
also connected to the negative side of the bootstrap capacitors
and are the negative supply connections for the floating high-side
drives. The discharge current from the high-side FET gate capaci-
tance flows through these connections, which should have low
impedance circuit connections to the FET bridge.
Four low-voltage level digital inputs provide control for the
gate drives. These logic inputs all have a nominal hysteresis of
500 mV to improve noise performance. They are used together
to provide fast decay or slow decay with high-side or low-side
recirculation. They also provide brake, coast, and sleep modes as
defined in tables 1 and 2.
PWMH and PWML Pins These inputs can be used to control
current in the power bridge. PWMH provides high-side chopping
and PWML provides low-side chopping. When used together
they control the power bridge in fast decay mode. The PWM
options are provided in table 2.
GHA and GHB Pins These terminals are the high-side gate
drive outputs for the external N-channel FETs. External resistors
between the gate drive output and the gate connection to the FET
(as close as possible to the FET) can be used to control the slew
rate seen at the gate, thereby controlling the di/dt and dv/dt of the
SA and SB outputs. GHx going high turns on the upper half of
the drive, sourcing current to the gate of the high-side FET in the
external motor-driving bridge, turning it on. GHx going low turns
on the lower half of the drive, sinking current from the external
FET gate circuit to the corresponding Sx pin, turning off the FET.
• Setting PWMH low turns off active high-side drives. This
provides high-side–chopped slow-decay PWM.
• Setting PWML low turns off active low-side drives. This
provides low-side–chopped slow-decay PWM.
• PWMH and PWML may also be connected together and driven
with a single PWM signal. This provides fast-decay PWM.
CA and CB Pins These are the high-side connections for the
bootstrap capacitors and are the positive supply for the high-side
gate drives. The bootstrap capacitors are charged to approxi-
mately VREG when the associated output Sx terminal is low.
When the Sx output swings high, the charge on the bootstrap
capacitor causes the voltage at the corresponding Cx terminal to
rise with the output to provide the boosted gate voltage needed
for the high-side FETs.
PHASE Pin The state of the PHASE pin determines the positive
direction of load current (see table 1). The PHASE pin can also
be used as a PWM input when full four-quadrant control (fast
decay synchronous rectification) is required (see table 2).
SR Pin This enables or disables synchronous rectification. When
SR is high, synchronous rectification is enabled during the PWM
off time. PWM off time is present when there is a low on either
Allegro MicroSystems, LLC
115 Northeast Cutoff
8
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
Table 1. Phase Control Truth Table
Inputs
Outputs
Bridge
Mode of Operation
PWMH PWML PHASE SR GHA GLA GHB GLB SA SB
1
1
0
1
0
0
1
1
0
1
1
1
0
1
1
0
0
0
1
0
X
X
1
0
1
0
X
X
X
1
1
0
0
0
0
X
H
L
L
H
L
L
H
L
L
L
H
H
L
L
H
L
H
L
H
L
H
L
L
L
L
HS
LS
LS
LS Bridge driven with A high and B low
HS Bridge driven with B high and A low
LS Slow decay, both low-side on or low-side brake
H
L
HS HS Slow decay, both high-side on or high-side brake
L
Z
LS
HS
Z
LS Slow decay, current flow A to B, low-side diode rectification
H
L
L
Z
Z
Slow decay, current flow B to A, low-side diode rectification
Slow decay, current flow A to B, high-side diode rectification
L
L
H
L
HS Slow decay, current flow B to A, high-side diode rectification
Fast decay, diode rectification/coast
L
Z
Z
X = don’t care (same for input 1 or input 0), HS = high-side FET active, LS = low-side FET active, Z = high impedance, both FETs off
Table 2. PWM Options
Inputsa
PWM Effectb
Decay
Fast
Mode of Operation
SR
PWMH
PWML
PHASE 100%
0%
X
1
1
PWM
A to B B to A
Full four-quadrant control, zero average load current at 50% PWM
Fast decay, diode recirculation or coast
1
0
1
0
1
0
1
0
1
0
X
A to B
Coast
B to A
0
1
1
0
PWM
PWM
1
Fast
A to B
Brake
B to A
PWM
1
Slow High-side PWM, low-side MOSFET recirculation
Slow Low-side PWM, high-side MOSFET recirculation
A to B
Brake
B to A
PWM
1
A to B
PWM
Brakec Slow High-side PWM, low-side diode recirculation
Brakec Slow Low-side PWM, high-side diode recirculation
B to A
A to B
B to A
0
1
0
PWM
0
X
Coast Coast
Fast
Coast, all MOSFETs off
aX indicates don’t care condition. The action is the same for input 1 or input 0.
bPWM Effect indicates the effect on the load current direction or the equivalent action.
cWith SR disabled, braking is only effective in one direction when sufficient forward voltage is available to allow the diode to conduct.
Allegro MicroSystems, LLC
115 Northeast Cutoff
9
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
but not both of the PWMH and PWML inputs. Synchronous
Several brake states are possible using combinations of inputs
rectification turns on the complementary MOSFET to the one that on PWMH, PWML, and SR. For example, holding PWML and
SR high, while PWMH is low, turns on both low-side FETs to
short the load. The shorting path is always present and provides
braking in both directions of motor rotation. Another example is
holding SR low, when PWML is high and PWMH is low, mak-
ing only one low-side FET active, and the braking current flow
through the body diode of the opposite low-side FET. This pro-
vides braking in only one direction, because the diode does not
permit the braking current to flow if the motor is reversed. Also,
the braking current can be made to circulate around the high-side
switches by swapping PWMH and PWML.
is turned off. This ensures that the recirculating current passes
through the lower resistance conduction path, rather than the
body diode of the MOSFET.
When SR is low, synchronous rectification is disabled. In this
case, fewer MOSFET switching cycles occur, reducing dissipa-
tion in the A3921. However, load current recirculates through the
higher resistance body diode of the power MOSFETs, causing
greater power dissipation in the power bridge.
RESET Pin This is an active-low input, and when active it allows
the A3921 to enter sleep mode. When RESET is held low, the
regulator and all internal circuitry are disabled and the A3921
enters sleep mode. Before fully entering sleep mode, there is a
short delay while the regulator decoupling and storage capacitors
discharge. This typically takes a few milliseconds, depending on
the application conditions and component values.
Diagnostics
Several diagnostic features are integrated into the A3921 to
provide indication of fault conditions and, if required, take action
to prevent permanent damage. In addition to system-wide faults
such as undervoltage and overtemperature, the A3921 integrates
individual drain-source monitors for each external FET, to pro-
vide short circuit detection.
During sleep mode, current consumption from the VBB supply
is reduced to a minimal level. In addition, latched faults and the
corresponding fault flags are cleared. When the A3921 is coming
out of sleep mode, the protection logic ensures that the gate drive
outputs are off until the charge pump reaches its correct operat-
ing condition. The charge pump stabilizes in approximately 3 ms
under nominal conditions.
Diagnostic Management Pins
VDSTH Pin Faults on the external FETs are determined by
measuring the drain-source voltage, VDS , of each active FET
and comparing it to the threshold voltage applied to the VDSTH
input, VDSTH. To avoid false fault detection during switching
transients, the comparison is delayed by an internal blanking
timer. If the voltage applied to the VDSTH pin is greater than the
disable threshold voltage, VDSDIS, then FET short circuit detec-
tion is disabled.
RESET can be used also to clear latched fault flags without
entering sleep mode. To do so, hold RESET low for less then the
reset pulse time, tRES. This clears any latched fault that disables
the outputs, such as short circuit detection or bootstrap capacitor
undervoltage.
Note that the A3921 can be configured to start without any exter-
nal logic input. To do so, pull up the RESET pin to VBB by means
of an external resistor. The resistor value should be between
20 and 33 kΩ.
VDRAIN Pin This is a low current sense input from the top of the
external FET bridge. This input allows accurate measurement of
the voltage at the drain of the high-side FETs. It should be con-
nected directly to the common connection point for the drains of
the power bridge FETs at the positive supply connection point.
The input current to the VDRAIN pin is proportional to the volt-
age on the VDSTH pin and can be approximated by:
Coast and Brake States
To put the power bridge into a coast state, that is all power bridge
MOSFETs switched off, the two PWM inputs, PWMH and
PWML, must be held low and at the same time SR must be held
low. This forces all gate drive outputs low.
IVDRAIN = 72 × VDSTH + 52 ,
where IVDRAIN is the current into the VDRAIN pin, in μA, and
VDSTH is the voltage on the VDSTH pin, in V.
Braking is achieved by forcing the power bridge to apply a short
across the load, allowing the back EMF of the load to generate a
braking torque.
FF1 and FF2 Pins These are open drain output fault flags, which
indicate fault conditions by their state, as shown in table 3. In
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Automotive Full Bridge MOSFET Driver
the event that two or more faults are detected simultaneously, the
state of the fault flags will be determined by a logical OR of the
flag states for all detected faults.
Bootstrap Capacitor Undervoltage The A3921 monitors the
voltage across the individual bootstrap capacitors to ensure they
have sufficient charge to supply the current pulse for the high-
side drive. Before a high-side drive can be turned on, the voltage
across the associated bootstrap capacitor must be higher than the
turn-on voltage limit. If this is not the case, then the A3921 will
start a bootstrap charge cycle by activating the complementary
low-side drive. Under normal circumstances, this will charge the
bootstrap capacitor above the turn-on voltage in a few microsec-
onds and the high-side drive will then be enabled.
Table 3. Fault Definitions
Flag State
Disable
Outputs*
Fault
Latched
Fault Description
FF1
FF2
Low No fault
Low
No
Yes
Yes
Yes
No
–
Low High Short-to-ground
Low High Short-to-supply
Low High Shorted load
Yes
Yes
Yes
No
The bootstrap voltage monitor remains active while the high-side
drive is active and, if the voltage drops below the turn-off volt-
age, a charge cycle is initiated.
High Low Overtemperature
High High V5 undervoltage
High High VREG undervoltage
High High Bootstrap undervoltage
Yes
Yes
Yes
No
No
In either case, if there is a fault that prevents the bootstrap capaci-
tor charging, then the charge cycle will timeout, the fault flags
(indicating an undervoltage) will be set, and the outputs will be
disabled. The bootstrap undervoltage fault state remains latched
until RESET is set low.
Yes
*Yes indicates all gate drives low, and all FETs off.
Fault States
V5 Undervoltage The output of the logic supply regulator volt-
age at V5 is monitored to ensure correct logical operation. If
the voltage at V5, V5 , drops below the falling V5 undervoltage
lockout threshold, V5UVoff , then the A3921 will enter the V5
undervoltage fault state. In this fault state, both FF1 and FF2 will
be high, and the outputs will be disabled. In addition, because
the state of other reported faults cannot be guaranteed, all fault
states and fault flags are reset and replaced by the fault flags cor-
responding to a V5 undervoltage fault state. For example, a V5
undervoltage will reset an existing short circuit fault condition
and replace it with a V5 undervoltage fault. The V5 undervoltage
fault state and the fault flags will be cleared when V5 rises above
the rising V5 undervoltage lockout threshold defined by V5UVoff
Overtemperature If the junction temperature exceeds the over-
temperature threshold, typically 165°C, the A3921 will enter the
overtemperature fault state and FF1 will go high. The overtem-
perature fault state, and FF1, will only be cleared when the tem-
perature drops below the recovery level defined by TJF – TJFhys
.
No circuitry will be disabled. External control circuits must take
action to limit the power dissipation in some way so as to prevent
overtemperature damage to the A3921 chip and unpredictable
device operation.
VREG Undervoltage VREG supplies the low-side gate driver
and the bootstrap charge current. It is critical to ensure that the
voltages are sufficiently high before enabling any of the outputs.
If the voltage at VREG, VREG, drops below the falling VREG
undervoltage lockout threshold, VREGUVoff, then the A3921 will
enter the VREG undervoltage fault state. In this fault state, both
FF1 and FF2 will be high, and the outputs will be disabled. The
VREG undervoltage fault state and the fault flags will be cleared
when VREG rises above the rising VREG undervoltage lockout
+ V5UVhys
.
The V5 undervoltage monitor circuit is active during power-up,
and the A3921 remains in the V5 undervoltage fault state until V5
is greater than the rising VREG undervoltage lockout threshold,
V
5UVoff +V5UVhys.
Short Fault Operation Shorts in the power bridge are determined
by monitoring the drain-souce voltage, VDS, of each active FET
and comparing it to the fault threshold voltage at the VDSTH pin.
Because power MOSFETs take a finite time to reach the rated
on-resistance, the measured drain-source voltages will show a
fault as the phase switches. To avoid such false short fault detec-
threshold, VREGUVon
.
The VREG undervoltage monitor circuit is active during
power-up, and the A3921 remains in the VREG undervoltage
fault state until VREG is greater than the rising VREG undervolt-
age lockout threshold, VREGUVon
.
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A3921
Automotive Full Bridge MOSFET Driver
tions, the output from the comparators are ignored under two
conditions:
The result of this comparison is ignored if the FET is not active.
It also is ignored for one fault blank time interval after the FET is
turned on. If, when the comparator is not being ignored, its output
indicates that VDS exceeds the voltage at the VDSTH pin, then
FF2 will be high.
▪ while the external FET is off, and
▪ until the end of the period, referred to as the fault blank time,
after the FET is turned on.
Short to Ground When VDSTH is less than the disable thresh-
old voltage, VDSDIS, a short from any of the motor phase con-
nections to ground is detected by monitoring the voltage across
the high-side FETs in each phase, using the appropriate Sx pin
and the voltage at VDRAIN. This drain-source voltage, VDS , is
continuously compared to the voltage on the VDSTH pin. The
result of this comparison is ignored if the FET is not active. It
also is ignored for one fault blank time interval after the FET is
turned on. If, when the comparator is not being ignored, its output
indicates that VDS exceeds the voltage at the VDSTH pin, FF2
will be high.
When the FET is turned on, if the drain-source voltage exceeds
the voltage at the VDSTH pin at any time after the fault blank
time, then a short fault will be detected. This fault will be latched
and the FET disabled until reset.
In applications where short detection is not required, this feature
may be disabled by connecting VDSTH to V5 or by applying a
voltage greater than the disable threshold voltage, VDSDIS. This
completely disables the VDS monitor circuits, preventing detec-
tion of short faults and any indication of short faults by the fault
flags. In this condition the external FETs will not be protected by
the A3921.
Shorted Load The short-to-ground and short-to-supply monitor
circuits will also detect a short across a motor phase winding. In
most cases, a shorted winding will be indicated by a high-side
and low-side fault being detected at the same time. In some cases
the relative impedances may permit only one of the shorts to be
Short to Supply When VDSTH is less than the disable threshold
voltage, VDSDIS, a short from any of the motor phase connec-
tions to the battery or VBB connection is detected by monitoring
the voltage across the low-side FETs in each phase, using the
appropriate Sx pin and the LSS pin. This drain-source voltage,
VDS, is continuously compared to the voltage on the VDSTH pin. detected.
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A3921
Automotive Full Bridge MOSFET Driver
Applications Information
Power Bridge Management Using PWM Control
The A3921 provides two PWM control signals, a phase control
for current direction, and the ability to enable or disable syn-
chronous rectification. This allows a wide variety of full bridge
control schemes to be implemented. The six basic schemes are
shown in table 2 and described further below.
GHA
GLA
GHB
GLB
GHA
GLA
GHB
GLB
LOAD
LOAD
Driving
Recirculating
Inputs
Outputs
Inputs
Outputs
Slow Decay Slow decay is the simplest and most common
control configuration. Figure 1A shows the path of the bridge
and load current when a PWM signal is applied to PWMH, with
PWML and PHASE tied high, and SR low.
Phase A
B
Phase A
B
PWMH
PWML
PHASE
SR
1
1
1
0
PWMH
PWML
PHASE
SR
0
1
1
0
GHx
H
GHx
H
L
L
GLx
GLx
L
H
L
H
In this case the high-side MOSFETs are switched off during the
current decay time (PWM off-time) and load current recirculates
through the low-side MOSFETs. This is commonly referred to as
high-side chopping or high-side PWM. The recirculating current
flows through the body diode of the low-side MOSFET, which
is complementary to the high-side MOSFET being switched off.
Improved efficiency can be achieved by turning on the comple-
mentary MOSFETs during the PWM off-time to short the reverse
diode and provide synchronous rectification. This can be easily
achieved by taking SR high as shown in figure 1B.
(A) Slow decay, diode recirculation, high-side PWM
GHA
GLA
GHB
GLB
GHA
GLA
GHB
GLB
LOAD
LOAD
Driving
Recirculating
By applying the PWM signal to the PWML pin instead of the
PWMH pin, the low-side MOSFET is turned off during the PWM
off-time and the load current recirculates through the high-side
MOSFETs as in figure 1C.
Inputs
Outputs
Phase A
Inputs
Outputs
Phase A
B
B
PWMH
PWML
PHASE
SR
1
1
1
1
PWMH
PWML
PHASE
SR
0
1
1
1
GHx
GLx
GHx
GLx
H
L
L
L
L
H
H
H
In the three slow decay configurations shown, the direction of the
average current in the load can be reversed by simply applying a
low level to the PHASE pin. Referring to the slow decay entries
in table 2, when PHASE is high the average current flows from
the phase A connection (SA) to the phase B connection (SB).
When PHASE is low the direction is from B to A.
(B) Slow decay, SR active, high-side PWM
Fast Decay While slow decay usually provides sufficient control
over the load current for most simple control systems, it is pos-
sible that current control stability can be affected by, for example,
the back EMF of the load. In these cases, typically actuator posi-
tioning or servo control systems, it may be necessary to use fast
decay to provide continuous control over the load current. The
A3921 can be configured to provide fast decay using either diode
recirculation or synchronous rectification.
GHA
GLA
GHB
GLB
GHA
GLA
GHB
GLB
LOAD
LOAD
Driving
Outputs
Phase A
Recirculating
Inputs
Inputs
Outputs
Phase A
B
B
PWMH
PWML
PHASE
SR
1
1
1
1
PWMH
PWML
PHASE
SR
1
0
1
1
GHx
GLx
GHx
GLx
H
L
L
H
L
H
L
H
Fast decay with diode recirculation is achieved by applying a
PWM signal at the same time to both PWM inputs, PWMH and
PWML, with SR disabled (figure 2A). Because current recircu-
lation is through the body diodes of the MOSFETs, the aver-
age load current cannot be negative so, as for the slow decay
(C) Slow decay, SR active, low-side PWM
Figure 1. Slow decay power bridge current paths
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A3921
Automotive Full Bridge MOSFET Driver
schemes, the PHASE input is still required to reverse the load
current.
GHA
GLA
GHB
GLB
GHA
GLA
GHB
GLB
Although fast decay with diode rectification provides a higher
degree of current control than slow decay schemes, it still may
not provide sufficient control for servo systems where full four-
quadrant control is required. This is only possible using fast
decay with synchronous rectification. By applying the PWM sig-
nal to the PHASE input, and holding PWMH and PWML and SR
high (figure 2B), the load current can be controlled in both direc-
tions with a single PWM signal. Because all four MOSFETs in
the bridge change state, the supply can be directly applied to the
load in either direction. The effect is: when the PWM duty cycle
is less than 50%, the average current flows from B to A; when
greater than 50%, the average current flows from A to B; and
when at 50%, the average current is zero. This allows the load
current to be independent of any back EMF voltage generated, for
example by a rotating motor, and effectively allowing the applied
torque to work with or against a motor in either direction.
LOAD
LOAD
Driving
Recirculating
Inputs
Outputs
Phase A
Inputs
Outputs
Phase A
B
B
PWMH
PWML
PHASE
SR
1
1
1
0
PWMH
PWML
PHASE
SR
0
0
1
0
GHx
GLx
GHx
GLx
H
L
L
L
L
L
L
H
(A) Fast decay, diode recirculation
GHA
GLA
GHB
GLB
GHA
GLA
GHB
GLB
LOAD
LOAD
Driving
Outputs
Phase A
Recirculating
Inputs Outputs
Phase A
Inputs
B
B
Synchronous Rectification Synchronous rectification is used to
reduce power dissipation in the external MOSFETs. As described
above, the A3921 can be instructed to turn on the appropriate
low-side and high-side driver during the load current recirculation
PWM off-cycle. During the decay time, synchronous rectifica-
tion allows current to flow through the selected MOSFET, rather
than through the source-drain body diode. The body diodes of the
recirculating power MOSFETs will conduct only during the dead
time that occurs at each PWM transition.
PWMH
PWML
PHASE
SR
1
1
1
1
PWMH
PWML
PHASE
SR
1
1
0
1
GHx
GLx
GHx
GLx
H
L
L
L
H
L
H
H
(B) Fast decay, SR active, full four-quadrant control
Figure 2. Fast decay power bridge current paths
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0
Dead Time
To prevent cross-conduction (shoot through) in any phase of
the power FET bridge, it is necessary to have a dead time delay,
tDEAD, between a high-side or low-side turn-off and the next
complementary turn-on event. The potential for cross-conduction
occurs when any complementary high-side and low-side pair of
FETs are switched at the same time; for example, when using
synchronous rectification or after a bootstrap capacitor charg-
ing cycle. In the A3921, the dead time for both phases is set by
a single dead-time resistor, RDEAD , between the RDEAD and
AGND pins.
0
50
100 150 200 250 300 350 400 450
RDEAD (kΩ)
Figure 3. Dead time versus RDEAD, (full range)
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For RDEAD values between 3 kΩ and 240 kΩ, at 25°C the nomi-
nal value of tDEAD in ns can be approximated by:
7200
PWML=0). This effectively short-circuits the back EMF of the
motor, creating a breaking torque.
During braking, the load current can be approximated by:
,
tDEAD(nom)
50 +
=
(1)
1.2 + (200 / RDEAD
)
VBEMF
IBRAKE
,
=
(3)
where RDEAD is in kΩ. Greatest accuracy is obtained for values
of RDEAD between 6 and 60 kΩ, which are shown in figure 3.
RL
where VBEMF is the voltage generated by the motor and RL is the
resistance of the phase winding.
The IDEAD current can be estimated by:
1.2
Care must be taken during braking to ensure that maximum rat-
ings of the power FETs are not exceeded. Dynamic braking is
equivalent to slow decay with synchronous rectification.
IDEAD
.
=
(2)
RDEAD
The maximum dead time, 6 μs typical, can be set by connecting
the RDEAD pin directly to the V5 pin.
The choice of power FET and external series gate resistance
determine the selection of the dead-time resistor, RDEAD. The
dead time should be long enough to ensure that one FET in a
phase has stopped conducting before the complementary FET
starts conducting. This should also take into account the tolerance
and variation of the FET gate capacitance, the series gate resis-
tance, and the on-resistance of the A3921 internal drives.
Bootstrap Capacitor Selection
The bootstrap capacitors, CBOOTx, must be correctly selected to
ensure proper operation of the A3921. If the capacitances are too
high, time will be wasted charging the capacitor, resulting in a
limit on the maximum duty cycle and the PWM frequency. If the
capacitances are too low, there can be a large voltage drop at the
time the charge is transferred from CBOOTx to the FET gate, due
to charge sharing.
Dead time will be present only if the on-command for one FET
occurs within tDEAD after the off-command for its complementary
FET. In the case where one side of a phase drive is permanently
off, for example when using diode rectification with slow decay,
then the dead time will not occur. In this case the gate drive will
turn on within the specified propagation delay after the corre-
sponding phase input goes high. (Refer to the Gate Drive Timing
diagrams.)
To keep this voltage drop small, the charge in the bootstrap
capacitor, QBOOT, should be much larger than the charge required
by the gate of the FET, QGATE. A factor of 20 is a reasonable
value, and the following formula can be used to calculate the
value for CBOOT
:
QBOOT CBOOT × VBOOT = QGATE × 20 ,
=
therefore:
Fault Blank Time
QGATE × 20
To avoid false short fault detection, the output from the VDS
monitor for any FET is ignored when that FET is off and for a
period of time after it is turned on. This period of time is the fault
blank time. Its length is the dead time, tDEAD , plus an additional
period of time that compensates for the delay in the VDS moni-
tors. This additional delay is typically 300 to 600 ns.
,
CBOOT
=
(4)
(5)
VBOOT
where VBOOT is the voltage across the bootstrap capacitor.
The voltage drop across the bootstrap capacitor as the FET is
being turned on, ∆V, can be approximated by:
QGATE
.
∆V
≈
CBOOT
Braking
So, for a factor of 20, ∆V would be approximately 5% of VBOOT
.
The A3921 can be used to perform dynamic braking either by
forcing all low-side FETs on and all high-side FETs off (SR=1,
PWMH=0, and PWML=1) or conversely by forcing all low-
side FETs off and all high-side FETs on (SR=1, PWMH=1, and
The maximum voltage across the bootstrap capacitor under
normal operating conditions is VREG(max). However, in some
circumstances the voltage may transiently reach 18 V, the clamp
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voltage of the Zener diodes between the Cx and Sx pins. In most
applications, with a good ceramic capacitor the working voltage
can be limited to 16 V.
capacitors. When a low-side FET is turned on, the gate-drive
circuit will provide the high transient current to the gate that is
necessary to turn on the FET quickly. This current, which can be
several hundred milliamperes, cannot be provided directly by the
limited output of the VREG regulator, and must be supplied by an
external capacitor connected to VREG.
Bootstrap Charging
It is good practice to ensure the high-side bootstrap capacitor is
completely charged before a high-side PWM cycle is requested.
The time required to charge the capacitor, tCHARGE (μs), is
approximated by:
The turn-on current for the high-side FET is similar in value to
that for the low-side FET, but is mainly supplied by the boot-
strap capacitor. However the bootstrap capacitor must then be
recharged from the VREG regulator output. Unfortunately the
bootstrap recharge can occur a very short time after the low-
side turn-on occurs. This requires that the value of the capacitor
connected between VREG and AGND should be high enough to
minimize the transient voltage drop on VREG for the combina-
tion of a low-side FET turn-on and a bootstrap capacitor recharge.
A value of 20 × CBOOT is a reasonable value. The maximum
working voltage will never exceed VREG , so the capacitor can be
rated as low as 15 V. This capacitor should be placed as close as
possible to the VREG pin.
CBOOT × ∆V
tCHARGE
,
=
(6)
100
where CBOOT is the value of the bootstrap capacitor, in nF, and
∆V is the required voltage of the bootstrap capacitor.
At power-up and when the drives have been disabled for a long
time, the bootstrap capacitor can be completely discharged. In
this case ∆V can be considered to be the full high-side drive
voltage, 12 V. Otherwise, ∆V is the amount of voltage dropped
during the charge transfer, which should be 400 mV or less.
The capacitor is charged whenever the Sx pin is pulled low and
current flows from VREG through the internal bootstrap diode
circuit to CBOOT
.
Supply Decoupling
Because this is a switching circuit, there are current spikes from all
supplies at the switching points. As with all such circuits, the power
supply connections should be decoupled with a ceramic capacitor,
typically 100 nF, between the supply pin and ground. These capaci-
tors should be connected as close as possible to the device supply
pins VBB and V5, and the ground pin, GND.
Bootstrap Charge Management
The A3921 provides automatic bootstrap capacitor charge man-
agement. The bootstrap capacitor voltage for each phase is con-
tinuously checked to ensure that it is above the bootstrap under-
voltage threshold, VBOOTUV. If the bootstrap capacitor voltage
drops below this threshold, the A3921 will turn on the necessary
low-side FET, and continue charging until the bootstrap capacitor
exceeds the undervoltage threshold plus the hysteresis, VBOO-
TUV + VBOOTUVhys. The minimum charge time is typically 7 μs,
but may be longer for very large values of bootstrap capacitor
(>1000 nF). If the bootstrap capacitor voltage does not reach the
threshold within approximately 200 μs, an undervoltage fault will
be flagged.
Power Dissipation
In applications where a high ambient temperature is expected, the
on-chip power dissipation may become a critical factor. Careful
attention should be paid to ensure the operating conditions allow
the A3921 to remain in a safe range of junction temperature.
The power consumed by the A3921, PD, can be estimated by:
PD PBIAS + PCPUMP + PSWITCHING
,
=
(7)
(8)
VREG Capacitor Selection
given:
The internal reference, VREG, supplies current for the low-side
gate drive circuits and the charging current for the bootstrap
PBIAS VBB × IBB
;
=
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A3921
Automotive Full Bridge MOSFET Driver
• Sensitive connections such as RDEAD and VDSTH, which
PCPUMP [( 2 VBB) – VREG
]
IAV
, for VBB < 15 V,
, for VBB ≥ 15 V,
=
have very little ground current, should be connected to the Quiet
ground (refer to figure 4), which is connected independently,
closest to the GND pin. These sensitive components should
never be connected directly to the supply common or to a
common ground plane. They must be referenced directly to the
GND pin.
or
[VBB – VREG
]
IAV
=
(9)
(10)
PSWITCHING QGATE × VREG × N × fPWM × Ratio ;
=
where:
IAV QGATE × N × fPWM
,
=
• The supply decoupling for VBB, VREG, and V5 should
be connected to the Controller Supply ground, which is
independently connected close to the GND pin. The decoupling
capacitors should also be connected as close as practicable to
the relevant supply pin.
• If layout space is limited, then the Quiet and Controller Supply
grounds may be combined. In this case, ensure that the ground
return of the dead time resistor is close to the GND pin.
• Check the peak voltage excursion of the transients on the LSS
pin with reference to the GND pin, using a close grounded (tip
and barrel) probe. If the voltage at LSS exceeds the absolute
maximum shown in this datasheet, add either or both of
additional clamping and capacitance between the LSS pin and
the GND pin, as shown in figure 4.
N is the number of FETs switching during a PWM cycle, and
10
.
Ratio
=
R
GATE + 10
N=1 for slow decay with diode recirculation, N=2 for slow decay
with synchronous rectification or for fast decay with diode recir-
culation, and N=4 for fast decay with synchronous rectification.
Layout Recommendations
Careful consideration must be given to PCB layout when design-
ing high frequency, fast switching, high current circuits. The
following are recommendations regarding some of these consid-
erations:
• The A3921 ground, GND, and the high-current return of the
external FETs should return separately to the negative side
of the motor supply filtering capacitor. This will minimize
the effect of switching noise on the device logic and analog
reference.
• Gate charge drive paths and gate discharge return paths may
carry a large transient current pulse. Therefore, the traces from
GHx, GLx, Sx, and LSS should be as short as possible to reduce
the circuit trace inductance.
• The exposed thermal pad should be connected to the GND
pin and may form part of the Controller Supply ground (see
figure 4).
• Provide an independent connection from LSS to the common
point of the power bridge. It is not recommended to connect
LSS directly to the GND pin, as this may inject noise into
sensitive functions such as the timer for dead time.
• A low-cost diode can be placed in the connection to VBB to
provide reverse battery protection. In reverse battery conditions,
it is possible to use the body diodes of the power FETs to clamp
the reverse voltage to approximately 4 V. In this case, the
additional diode in the VBB connection will prevent damage
to the A3921 and the VDRAIN input will survive the
reverse voltage.
• Minimize stray inductance by using short, wide copper traces at
the drain and source terminals of all power FETs. This includes
motor lead connections, the input power bus, and the common
source of the low-side power FETs. This will minimize voltages
induced by fast switching of large load currents.
• Consider the use of small (100 nF) ceramic decoupling
capacitors across the sources and drains of the power FETs to
limit fast transient voltage spikes caused by the inductance of
the circuit trace.
• Keep the gate discharge return connections Sx and LSS as short
as possible. Any inductance on these traces will cause negative
transitions on the corresponding A3921 pins, which may exceed
the absolute maximum ratings. If this is likely, consider the use
of clamping diodes to limit the negative excursion on these pins
with respect to GND.
Note that the above are only recommendations. Each application
is different and may encounter different sensitivities. A driver
running a few amps will be less susceptible than one running with
150 A, and each design should be tested at the maximum current
to ensure any parasitic effects are eliminated.
Allegro MicroSystems, LLC
115 Northeast Cutoff
17
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
Optional reverse
battery protection
VBB
VDRAIN
+ Supply
GHB
VREG
GHA
A3921
VS
SA
SB
Motor
GLA
VDSTH
RDEAD
GLB
LSS
GND
RS
Optional components
to limit LSS transients
Supply
Common
Quiet Ground
Power Ground
Controller Supply Ground
Figure 4. Supply routing suggestions
Input and Output Structures
CP1
CP2
VDRAIN
VBB
VS
Cx
VBB
20 V
18 V
19 V
ESD
18 V
18 V
18 V
GHx
19 V
20 V
18 V
6 V
Sx
(B) Supply protection structures
VREG
ESD
18 V
18 V
GLx
LSS
3 kΩ
10 Ω
RESET
FFx
50 kΩ
6 V
6 V
(A) Gate drive outputs
(C) Fault output
(D) RESET input
ESD
ESD
ESD
1.2 V
100 Ω
1 kΩ
3 kΩ
PWMx
SR
VDSTH
RDEAD
PHASE
8.5 V
8.5 V
8.5 V
(E) Logic inputs, no pulldown
(F) VDS monitor threshold input
(G) RDEAD
Allegro MicroSystems, LLC
115 Northeast Cutoff
18
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
Pin-out Diagram
1
2
28
27
26
25
24
23
22
21
20
19
18
17
16
15
VDRAIN
LSS
GLB
SB
VDSTH
RDEAD
FF2
tDEAD
3
4
FF1
5
GHB
CB
RESET
PWMH
PWML
SR
6
7
NC
Reg
8
VREG
CA
9
Reg
V5
10
11
12
13
14
GHA
SA
PHASE
NC
GLA
VBB
NC
GND
CP1
Charge
Pump
CP2
Terminal List
Number
Name
VDRAIN
LSS
GLB
SB
Description
Number
16
Name
Description
Pump capacitor
1
2
High-side common drain
CP1
GND
Low-side common source
Low-side gate drive B
Load connection B
High-side gate drive B
Bootstrap capacitor B
No connection
17
Ground
3
18
NC
No connection
4
19
PHASE
V5
Phase control input
5 V regulator
5
GHB
CB
20
6
21
SR
SR control input
7
NC
22
PWML
PWMH
RESET
FF1
Low-side PWM control input
High-side PWM control input
Reset input
8
VREG
CA
Regulated 13 V
23
9
Bootstrap capacitor A
High-side gate drive A
Load connection A
Low-side gate drive A
Main supply
24
10
11
12
13
14
15
GHA
SA
25
Fault Flag 1 output
Fault Flag 2 output
Dead time setting input
26
FF2
GLA
VBB
NC
27
RDEAD
VDSTH
28
VDS threshold level Input
No connection
Exposed pad for enhanced thermal
dissipation (underside)
–
PAD
CP2
Pump capacitor
Allegro MicroSystems, LLC
115 Northeast Cutoff
19
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
Package LP 28-Pin TSSOP with Exposed Thermal Pad
0.45
9.70 ±0.10
0.65
28
4° ±4
0.15
28
+0.05
–0.06
1.65
B
4.40 ±0.10 6.40 ±0.20
3.00
6.10
3.00
0.60 ±0.15
(1.00)
A
1
2
5.00
0.25
1
2
5.00
C
28X
SEATING PLANE
GAUGE PLANE
SEATING
PLANE
0.10
C
C
PCB Layout Reference View
+0.05
–0.06
1.20 MAX
0.10 MAX
0.25
0.65
For reference only
(reference JEDEC MO-153 AET)
Dimensions in millimeters
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
Terminal #1 mark area
A
B
C
Exposed thermal pad (bottom surface)
Reference land pattern layout (reference IPC7351 SOP65P640X120-29CM);
All pads a minimum of 0.20 mm from all adjacent pads; adjust as necessary
to meet application process requirements and PCB layout tolerances; when
mounting on a multilayer PCB, thermal vias at the exposed thermal pad land
can improve thermal dissipation (reference EIA/JEDEC Standard JESD51-5)
Allegro MicroSystems, LLC
115 Northeast Cutoff
20
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A3921
Automotive Full Bridge MOSFET Driver
Copyright ©2011-2013, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in life support devices or systems, if a failure of an Allegro product can reasonably be expected to cause the
failure of that life support device or system, or to affect the safety or effectiveness of that device or system.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
www.allegromicro.com
Allegro MicroSystems, LLC
115 Northeast Cutoff
21
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
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