ADF7012BRU-REEL [ADI]
Multichannel ISM Band FSK/GFSK/OOK/GOOK/ASK Transmitter; 多通道ISM频段FSK / GFSK / OOK / GOOK / ASK发射器
| 型号： | ADF7012BRU-REEL |
| 厂家： | 
ADI |
| 描述： | Multichannel ISM Band FSK/GFSK/OOK/GOOK/ASK Transmitter |
| 文件： | 总28页 (文件大小：893K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Multichannel ISM Band
FSK/GFSK/OOK/GOOK/ASK Transmitter
ADF7012
FEATURES
GENERAL DESCRIPTION
Single-chip, low power UHF transmitter
75 MHz to 1 GHz frequency operation
Multichannel operation using Frac-N PLL
2.3 V to 3.6 V operation
On-board regulator—stable performance
Programmable output power:
−16 dBm to +14 dBm, 0.4 dB steps
Data rates: dc to 179.2 kbps
The ADF7012 is a low power FSK/GFSK/OOK/GOOK/ASK
UHF transmitter designed for short range devices (SRDs). The
output power, output channels, deviation frequency, and mod-
ulation type are programmable by using four, 32-bit registers.
The fractional-N PLL and VCO with external inductor enable
the user to select any frequency in the 75 MHz to 1 GHz band.
The fast lock times of the fractional-N PLL make the ADF7012
suitable in fast frequency hopping systems. The fine frequency
deviations available and PLL phase noise performance facilitates
narrow-band operation.
Low current consumption:
868 MHz, 10 dBm, 21 mA
433 MHz, 10 dBm, 17 mA
315 MHz, 0 dBm, 10 mA
There are five selectable modulation schemes: binary frequency
shift keying (FSK), Gaussian frequency shift keying (GFSK),
binary on-off keying (OOK), Gaussian on-off keying (GOOK),
and amplitude shift keying (ASK). In the compensation register,
the output can be moved in <1 ppm steps so that indirect com-
pensation for frequency error in the crystal reference can be
made.
Programmable low battery voltage indicator
24-lead TSSOP
APPLICATIONS
Low cost wireless data transfer
Security systems
RF remote controls
Wireless metering
A simple 3-wire interface controls the registers. In power-down,
the part has a typical quiescent current of <0.1 μA.
Secure keyless entry
FUNCTIONAL BLOCK DIAGRAM
C
REG
PRINTED
INDUCTOR
CLK
CPV
CP
GND
OUT
DD
OSC1
OSC2
L1
L2
C
VCO
V
DD
OOK\ASK
PA
÷CLK
VCO
RF
OUT
÷R
PFD/
CHARGE
PUMP
RF GND
DV
DD
D
C
REG
GND
LDO
REGULATOR
+FRACTIONAL N
OOK\ASK
FSK\GFSK
TxCLK
Σ-∆
TxDATA
PLL LOCK
DETECT
MUXOUT
LE
DATA
CLK
FREQUENCY
COMPENSATION
MUXOUT
SERIAL
INTERFACE
R
SET
BATTERY
MONITOR
CENTER
FREQUENCY
CE
A
GND
Figure 1.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.326.8703
www.analog.com
© 2004 Analog Devices, Inc. All rights reserved.
ADF7012
TABLE OF CONTENTS
Specifications..................................................................................... 3
GOOK Modulation.................................................................... 15
Output Divider ........................................................................... 16
MUXOUT Modes....................................................................... 16
Theory of Operation ...................................................................... 17
Choosing the External Inductor Value.................................... 17
Choosing the Crystal/PFD Value............................................. 17
Tips on Designing the Loop Filter ........................................... 17
PA Matching................................................................................ 18
Transmit Protocol and Coding Considerations ..................... 18
Application Examples .................................................................... 19
315 MHz Operation................................................................... 20
433 MHz Operation................................................................... 21
868 MHz Operation................................................................... 22
915 MHz Operation................................................................... 23
Register Descriptions..................................................................... 24
R Register..................................................................................... 24
N-Counter Latch ........................................................................ 25
Modulation Register .................................................................. 26
Function Register ....................................................................... 27
Outline Dimensions....................................................................... 28
Ordering Guide .......................................................................... 28
Timing Characteristics..................................................................... 5
Absolute Maximum Ratings............................................................ 6
Transistor Count ........................................................................... 6
ESD Caution.................................................................................. 6
Pin Configuration and Function Descriptions............................. 7
Typical Performance Characteristics ............................................. 8
315 MHz ........................................................................................ 8
433 MHz ........................................................................................ 9
868 MHz ...................................................................................... 10
915 MHz ...................................................................................... 11
Circuit Description......................................................................... 12
PLL Operation ............................................................................ 12
Crystal Oscillator........................................................................ 12
Crystal Compensation Register................................................ 12
Clock Out Circuit....................................................................... 12
Loop Filter ................................................................................... 13
Voltage-Controlled Oscillator (VCO) ..................................... 13
Voltage Regulators ...................................................................... 13
FSK Modulation.......................................................................... 13
GFSK Modulation ...................................................................... 14
Power Amplifier.......................................................................... 14
REVISION HISTORY
10/04—Revision 0: Initial Version
Rev. 0 | Page 2 of 28
ADF7012
SPECIFICATIONS
DVDD = 2.3 V – 3.6 V; AGND = DGND = 0 V; TA = TMIN to TMAX, unless otherwise noted. Operating temperature range is −40°C to +85°C.
Table 1.
Parameter
B Version
Unit
Conditions/Comments
RF OUTPUT CHARACTERISTICS
Operating Frequency
75/1000
FRF/128
MHz min/max
Hz min
VCO range adjustable using external inductor;
divide-by-2, -4, -8 options may be required
Phase Frequency Detector
MODULATION PARAMETERS
Data Rate FSK/GFSK
179.2
64
kbps
Kbps
Using 1 MHz loop bandwidth
Based on US FCC 15.247 specfications for ACP; higher
data rates are achievable depending on local regulations
For example, 10 MHz PFD – deviation min = 610 Hz
For example, 10 MHz PFD – deviation max = 311.7 kHz
Data Rate ASK/OOK
Deviation FSK/GFSK
PFD/214
511 × PFD/214
0.5
Hz min
Hz max
typ
GFSK BT
ASK Modulation Depth
OOK Feedthrough (PA Off)
25
−40
−80
dB max
dBm typ
dBm typ
FRF = Fvco
FRF = Fvco/2
POWER AMPLIFIER PARAMETERS
Max Power Setting, DVDD = 3.6 V
Max Power Setting, DVDD = 3.0 V
Max Power Setting, DVDD = 2.3 V
Max Power Setting, DVDD = 3.6 V
Max Power Setting, DVDD = 3.0 V
Max Power Setting, DVDD = 2.3 V
PA Programmability
14
dBm
dBm
dBm
dBm
dBm
dBm
dB typ
FRF = 915 MHz, PA is matched into 50 Ω
FRF = 915 MHz, PA is matched into 50 Ω
FRF = 915 MHz, PA is matched into 50 Ω
FRF = 433 MHz, PA is matched into 50 Ω
FRF = 433 MHz, PA is matched into 50 Ω
FRF = 433 MHz, PA is matched into 50 Ω
PA output = −20 dBm to +13 dBm
13.5
12.5
14.5
14
13
0.4
POWER SUPPLIES
DVDD
2.3/3.6
V min/V max
Current Comsumption
315 MHz, 0 dBm/5 dBm
433 MHz, 0 dBm/10 dBm
8/14
mA typ
mA typ
mA typ
mA typ
mA min/max
µA typ
µA typ
DVDD = 3.0 V, PA is matched into 50 Ω, IVCO = min
VCO current consumption is programmable
10/18
14/21/32
16/24/35
1/8
190
280
868 MHz, 0 dBm/10 dBm/14 dBm
915 MHz, 0 dBm/10 dBm/14 dBm
VCO Current Consumption
Crystal Oscillator Current Consumption
Regulator Current Consumption
Power-Down Current
0.1/1
µA typ/max
REFERENCE INPUT
Crystal Reference Frequency
3.4/26
3.4/26
MHz min/max
MHz min/max
ms typ
Single-Ended Reference Frequency
Crystal Power-On Time 3.4 MHz/26 MHz 1.8/2.2
Single-Ended Input Level CMOS Levels
CE to Clock Enable Valid
Refer to the LOGIC INPUTS parameter. Applied to OSC 2 –
oscillator circuit disabled.
Rev. 0 | Page 3 of 28
ADF7012
Parameter
B Version
Unit
Conditions/Comments
PHASE-LOCKED LOOP PARAMETERS
VCO Gain
315MHz
433MHz
868MHz
22
MHz/V typ
MHz/V typ
MHz/V typ
MHz/V typ
V min/max
dBc
VCO divide-by-2 active
VCO divide-by-2 active
24
80
915MHz
88
VCO Tuning Range
Spurious (IVCO Min/Max)
Charge Pump Current
Setting [00]
0.3/2.0
−65/−70
IVCO is programmable
0.3
0.9
1.5
2.1
mA typ
mA typ
mA typ
mA typ
Refering to DB[7:6] in Function Register
Refering to DB[7:6] in Function Register
Refering to DB[7:6] in Function Register
Refering to DB[7:6] in Function Register
Setting [01]
Setting [10]
Setting [11]
Phase Noise (In band)1
315MHz
−85
−83
−80
−80
dBc/Hz typ
dBc/Hz typ
dBc/Hz typ
dBc/Hz typ
PFD = 10 MHz, 5 kHz offset, IVCO = 2 mA
PFD = 10 MHz, 5 kHz offset, IVCO = 2 mA
PFD = 10 MHz, 5 kHz offset, IVCO = 3 mA
PFD = 10 MHz, 5 kHz offset, IVCO = 3 mA
433MHz
868MHz
915MHz
Phase Noise (Out of Band)1
315MHz
−103
−104
−115
−114
−20
dBc/Hz typ
dBc/Hz typ
dBc/Hz typ
dBc/Hz typ
dBc typ
PFD = 10 MHz, 1 MHz offset, IVCO = 2 mA
PFD = 10 MHz, 1 MHz offset, IVCO = 2 mA
PFD = 10 MHz, 1 MHz offset, IVCO = 3 mA
PFD = 10 MHz, 1 MHz offset, IVCO = 3 mA
FRF = FVCO
433MHz
868MHz
915MHz
Harmonic Content (Second)2
Harmonic Content (Third)2
Harmonic Content (Others)2
Harmonic Content (Second)2
Harmonic Content (Third)2
Harmonic Content (Others)2
LOGIC INPUTS
−30
dBc typ
−27
dBc typ
−24
dBc typ
FRF = FVCO/N (where N = 2, 4, 8)
−14
dBc typ
−19
dBc typ
Input High Voltage,VINH
0.7 × DVDD
V min
Input Low Voltage, VINL
Input Current, IINH/IINL
Input Capacitance, CIN
LOGIC OUTPUTS
0.2 × DVDD
V max
µA max
pF max
1
4.0
Output High Voltage, VOH
DVDD − 0.4
500
V min
CMOS output chosen
IOL = 500 µA
Output High Current, IOH,
Output Low Voltage, VOL
µA max
V max
0.4
1 Measurements made with NFRAC = 2048.
2 Measurements made without harmonic filter.
Rev. 0 | Page 4 of 28
ADF7012
TIMING CHARACTERISTICS
DVDD = 3 V 10ꢀ; AGND = DGND = 0 V; TA = TMIN to TMAX, unless otherwise noted.
Table 2.
Parameter
Limit at TMIN to TMAX (B Version)
Unit
Test Conditions/Comments
LE setup time
Data-to-clock setup time
Data-to-clock hold time
Clock high duration
Clock low duration
t1
t2
t3
t4
t5
t6
t7
20
10
10
25
25
10
20
ns min
ns min
ns min
ns min
ns min
ns min
ns min
Clock –to-LE setup time
LE pulse width
t4
t5
CLOCK
t2
t3
DB1
DB0 (LSB)
(CONTROL BIT C1)
DB23 (MSB)
DB22
DB2
DATA
LE
(CONTROL BIT C2)
t7
t1
t6
LE
Figure 2. Timing Diagram
Rev. 0 | Page 5 of 28
ADF7012
ABSOLUTE MAXIMUM RATINGS
TA = 25°C, unless otherwise noted.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those listed in the operational sections
of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Table 3.
Parameter
Rating
DVDD to GND
(GND = AGND = DGND = 0 V)
−0.3 V to +3.9 V
Digital I/O Voltage to GND
Analog I/O Voltage to GND
Operating Temperature Range
Maximum Junction Temperature
TSSOP θJA Thermal Impedance
Lead Temperature, Soldering
Vapor Phase (60 sec)
−0.3 V to DVDD + 0.3 V
−0.3 V to DVDD + 0.3 V
150°C
150.4°C/W
This device is a high performance RF integrated circuit with an
ESD rating of 1 kV and it is ESD sensitive. Proper precautions
should be taken for handling and assembly.
215°C
220°C
Infrared (15 sec)
TRANSISTOR COUNT
35819 (CMOS)
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0 | Page 6 of 28
ADF7012
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
DV
DD
1
24
C
REG2
C
2
23
R
REG1
SET
CP
3
22 AGND
OUT
TSSOP
TxDATA
4
21 DV
20 RF
19 RF
DD
TxCLK
MUXOUT
DGND
5
OUT
GND
ADF7012
TOP VIEW
(Not to Scale)
6
7
18 VCO
IN
OSC1
8
17 C
VCO
OSC2
9
16 L2
15 L1
14 CE
13 LE
CLK
10
OUT
CLK 11
DATA 12
Figure 3.
Table 4. Pin Functional Descriptions
Pin No. Mnemonic Description
1
2
3
DVDD
CREG1
CPOUT
Positive Supply for the Digital Circuitry. This must be between 2.3 V and 3.6 V. Decoupling capacitors to the analog
ground plane should be placed as close as possible to this pin.
A 2.2 µF capacitor should be added at CREG to reduce regulator noise and improve stability. A reduced capacitor
improves regulator power-on time, but may cause higher spurious noise.
Charge Pump Output. This output generates current pulses that are integrated in the loop filter. The integrated
current changes the control voltage on the input to the VCO.
4
5
TxDATA
TxCLK
Digital Data to Be Transmitted is inputted on this pin.
GFSK and GOOK only. This clock output is used to synchronize microcontroller data to the TxDATA pin of the
ADF7012. The clock is provided at the same frequency as the data rate. The microcontroller updates TxDATA on
the falling edge of TxCLK. The rising edge of TxCLK is used to sample TxDATA at the midpoint of each bit.
6
MUXOUT
Provides the Lock_Detect Signal. This determines if the PLL is locked to the correct frequency and also monitors
battery voltage. Other signals include Regulator_Ready, which indicates the status of the serial interface regulator.
7
8
9
DGND
OSC1
OSC2
Ground for Digital Section.
The reference crystal should be connected between this pin and OSC2.
The reference crystal should be connected between this pin and OSC1. A TCXO reference may be used, by driving
this pin with CMOS levels, and powering down the crystal oscillator bit in software.
10
11
12
13
14
15
CLKOUT
CLK
DATA
LE
A divided-down version of the crystal reference with output driver. The digital clock output may be used to drive
several other CMOS inputs, such as a microcontroller clock. The output has a 50:50 mark-space ratio.
Serial Clock Input. This serial clock is used to clock in the serial data to the registers. The data is latched into the
32-bit shift register on the CLK rising edge. This input is a high impedance CMOS input.
Serial Data Input. The serial data is loaded MSB first with the two LSBs being the control bits. This is a high
impedance CMOS input.
Load Enable, CMOS Input. When LE goes high, the data stored in the shift registers is loaded into one of the four
latches, the latch being selected using the control bits.
Chip Enable. Bringing CE low puts the ADF7012 into complete power-down, drawing < 1uA. Register values are
lost when CE is low and the part must be reprogrammed once CE is brought high.
Connected to external printed or discrete inductor. See Choosing the External Inductor Value for advice on the
value of the inductor to be connected between L1 and L2.
CE
L1
16
17
L2
CVCO
Connected to external printed or discrete inductor.
A 220 nF capacitor should be tied between the CVCO and CREG2 pins. This line should run underneath the ADF7012.
This capacitor is necessary to ensure stable VCO operation.
18
VCOIN
The tuning voltage on this pin determines the output frequency of the voltage controlled oscillator (VCO). The
higher the tuning voltage, the higher the output frequency.
19
20
RFGND
RFOUT
Ground for Output Stage of Transmitter.
The modulated signal is available at this pin. Output power levels are from –16 dBm to +12 dBm. The output
should be impedance matched using suitable components to the desired load. See the PA Matching section.
21
DVDD
Voltage supply for VCO and PA section. This should have the same supply as DVDD Pin 1, and should be between
2.3 V and 3.6 V. Place decoupling capacitors to the analog ground plane as close as possible to this pin.
22
23
24
AGND
RSET
CREG2
Ground Pin for the RF Analog Circuitry.
External Resistor to set charge pump current and some internal bias currents. Use 3.6 kV as default.
Add a 470 nF capacitor at CREG to reduce regulator noise and improve stability. A reduced capacitor improves
regulator power-on time and phase noise, but may have stability issues over the supply and temperature.
Rev. 0 | Page 7 of 28
ADF7012
TYPICAL PERFORMANCE CHARACTERISTICS
315 MHZ
RBW
VBW
SWT
RF ATT 30dB
–60
1MHz
1MHz
17.5ms
REF LVL
5dBm
0.27dBm
308.61723447MHz
= NORMAL
FREQUENCY = 9.08 kHz
LEVEL = –84.47dBc/Hz
UNIT
dBm
5
0
–70
1
A
3
–80
–90
–10
–20
–30
1MA
4
2
–100
–110
–120
–40
–50
D1 –41.5dBm
–11.48dBm
–60
–70
–130
–140
1 [T1]
2 [T1]
0.27dBm
3 [T1]
4 [T1]
308.61723447MHz
–35.43dBm
939.87975952MHz
–34.11dBm
–80
631.26252505MHz
1.26252505GHz
–90
–95
1.0k
10.0k
100.0k
PHASE NOISE (Hz)
1.0M
10.0M
CENTER 3.5MHz
700MHz/
SPAN 7GHz
Figure 4. Phase Noise Response—DVDD = 3.0 V, ICP = 0.86 mA
IVCO = 2.0 mA, FOUT = 315 MHz, PFD = 3.6864 MHZ, PA Bias = 5.5 mA
Figure 7. Harmonic Response, RFOUT Matched to 50 Ω, No Filter
RBW
VBW
SWT
RF ATT 30dB
5kHz
5kHz
500ms
RBW
VBW
SWT
RF ATT 30dB
1MHz
1MHz
17.5ms
REF LVL
5dBm
0.45dBm
REF LVL
5dBm
0.18dBm
308.61723447MHz
315.05060120MHz
UNIT
dBm
UNIT
dBm
5
0
5
0
1
2
1
A
A
–10
–20
–30
SGL
–10
–20
–30
1MA
1MA
–40
–50
3
–40
–50
D1 –41.5dBm
–42.93dBm
2
4
–60
–70
–60
–70
1 [T1]
2 [T1]
0.18dBm
3 [T1]
4 [T1]
–80
308.61723447MHz
–50.53dBm
631.26252505MHz
939.87975952MHz
–55.48dBm
–80
1.26252505GHz
–90
–95
–90
–95
CENTER 315MHz
50kHz/
SPAN 500kHz
CENTER 3.5MHz
700MHz/
SPAN 7GHz
Figure 5. FSK Modulation, Power = 0 dBm, Data Rate = 1 kbps,
FDEVIATION 50 kHz
Figure 8. Harmonic Response, Fifth-Order Butterworth Filter
=
RBW
RF ATT 30dB
500kHz
500kHz
5ms
RBW
VBW
SWT
RF ATT 30dB
5kHz
5kHz
500ms
REF LVL
5dBm
0.31dBm
315.40080160MHz
VBW
SWT
REF LVL
5dBm
20.33dBm
UNIT
dBm
26.55310621kHz
UNIT
dBm
5
0
5
0
1
A
A
1
–10
–20
–30
–10
–20
–30
2
3
1MA
1MA
–40
–50
–40
–50
D1 –41.5dBm
D2 –49dBm
–60
–70
–60
–70
1 [T1]
3 [T1]
2 [T1]
–3.49dBm
315.00012525MHz
–20.33dB
–80
–80
26.55310621kHz
–20.85dB
1 [T1]
0.31dBm
315.40080160MHz
–90
–95
–90
–95
–27.55511022kHz
CENTER 315MHz
40MHz/
SPAN 400MHz
CENTER 315MHz
50kHz/
SPAN 500kHz
Figure 6. Spurious Components—Meets FCC Specs
Figure 9. OOK Modulation, Power = 0 dBm, Data Rate = 10 kbps
Rev. 0 | Page 8 of 28
ADF7012
433 MHZ
RBW
VBW
SWT
RF ATT 40dB
1 2.00V/ 2 1.00V/
1.50ms 500µs TRIG'D 1 720mv
30kHz
30kHz
90ms
REF LVL
15dBm
10.01dBm
433.91158317MHz
UNIT
dBm
15
10
1
A
0
–10
–20
1MA
2
1
CLKOUT
–30
–40
D1 –36dBm
–50
–60
CE
–70
–80
–85
CENTER 433.9500601MHz 3.2MHz/
SPAN 32MHz
Figure 10. Crystal Power-On Time, 4 MHz, Time = 1.6 ms
Figure 13. Spurious Components—Meets ETSI Specs
–40
RBW
VBW
SWT
RF ATT 40dB
1MHz
1MHz
17.5ms
REF LVL
15dBm
10.10dBm
= NORMAL
FREQUENCY = 393.38 kHz
LEVEL = –102.34dBc/Hz
434.86973948MHz
UNIT
dBm
15
10
–60
–80
1
A
0
–10
–20
3
2
–100
–120
–140
–160
4
1MA
–30
–40
D1 –30dBm
D1 –36dBm
–50
–60
–180
–200
1 [T1]
2 [T1]
10.10dBm
3 [T1]
4 [T1]
–5.12dBm
1.30460922GHz
–17.57dBm
434.86973948MHz
–15.25dBm
–70
869.73947896MHz
1.73947896GHz
–80
–85
1.0k
10.0k
100.0k
PHASE NOISE (Hz)
1.0M
10.0M
CENTER 3.5GHz
700MHz/
SPAN 7GHz
Figure 11. Phase Noise Response—ICP = 2.0 mA, IVCO = 2.0 mA,
RFOUT = 433.92 MHz, PFD = 4 MHz, PA Bias = 5.5 mA
Figure 14. Harmonic Response, RFOUT Matched to 50 Ω, No Filter
RBW
VBW
SWT
RF ATT 40dB
RBW
VBW
SWT
RF ATT 40dB
10kHz
300kHz
44ms
1MHz
1MHz
17.5ms
REF LVL
15dBm
5.60dBm
REF LVL
15dBm
9.51dBm
434.86973948MHz
433.91158317MHz
UNIT
dBm
UNIT
dBm
15
10
15
10
1
A
A
1
0
–10
–20
0
–10
–20
SGL
1MA
1MA
–30
–40
–30
–40
2
D1 –30dBm
D1 –36dBm
D1 –36dBm
3
4
–50
–60
–50
–60
1 [T1]
2 [T1]
9.51dBm
3 [T1]
4 [T1]
–43.60dBm
1.30460922GHz
–43.44dBm
434.86973948MHz
–33.75dBm
–70
–70
869.73947896MHz
1.73947896GHz
–80
–85
–80
–85
START 433.05MHz
174kHz/
STOP 434.79kHz
CENTER 3.5GHz
700MHz/
SPAN 7GHz
Figure 15. Harmonic Response, Fifth-Order Butterworth Filter
Figure 12. FSK Modulation, Power = 10 dBm, Data Rate = 38.4 kbps, FDEVIATION
19.28 kHz
=
Rev. 0 | Page 9 of 28
ADF7012
868 MHZ
RBW
VBW
SWT
RF ATT 40dB
0
1MHz
1MHz
16ms
REF LVL
15dBm
12.27dBm
= NORMAL
FREQUENCY = 251.3 kHz
LEVEL = –99.39dBc/Hz
869.33867735MHz
UNIT
dBm
15
10
–20
–40
1
A
2
0
–10
–20
4
3
–60
1MAX
1MA
–80
–30
–40
D1 –30dBm
–16.88dBm
–100
–120
–140
–50
–60
1 [T1]
2 [T1]
12.27dBm
869.33867735MHz
–4.00dBm
3 [T1]
4 [T1]
2.59699399GHz
–15.06dBm
–70
–160
1.0k
1.72865731GHz
3.46913828GHz
–80
–85
10.0k
100.0k
PHASE NOISE (Hz)
1.0M
10.0M
CENTER 3.8GHz
640MHz/
SPAN 6.4GHz
Figure 16. Phase Noise Response–ICP = 2.5 mA, IVCO = 1.44 mA, RFOUT = 868.95
MHz, PFD = 4.9152 MHz, Power = 12.5 dBm, PA Bias = Max
Figure 19. Harmonic Response, RFOUT Matched to 50 Ω, No Filter
RBW
VBW
SWT
RF ATT
30dB
MIXER –20dBm
UNIT dBm
RBW
VBW
SWT
RF ATT
30dB
MIXER –20dBm
UNIT dBm
10kHz
10kHz
15ms
1kHz
1kHz
10ms
REF LVL
15dBm
–40.44dBm
869.20000000MHz
REF LVL
15dBm
10.39dBm
869.33867735MHz
15
10
15
10
2
1
A
A
1 [T1]
10.39dBm
869.33867735MHz
–50.92dBm
1.72000000GHz
–50.40dBm
2.59600000GHz
LN
LN
3 [T1]
2 [T1]
0
–10
–20
0
–10
–20
1MAX
1MAX
1MA
1MA
–30
–40
–30
–40
D2 –30dBm
D2 –36dBm
1
3
2
–50
–60
–50
–60
1 [T1]
2 [T1]
40.44dBm
869.20000000MHz
8.02dBm
–70
–70
868.96673347MHz
–80
–85
–80
–85
CENTER 868.944489MHz
60kHz/
SPAN 600kHz
START 3.8GHz
640MHz/
SPAN 6.4GHz
Figure 17. FSK Modulation, Power = 12.5 dBm, Data Rate = 38.4 kbps,
FDEVIATION 19.2 kHz
Figure 20. Harmonic Response, Fifth-Order Chebyshev Filter
=
RBW
VBW
SWT
RF ATT
30dB
MIXER –20dBm
UNIT dBm
2kHz
2kHz
16s
REF LVL
15dBm
12.55dBm
869.025050100MHz
15
10
1
A
1 [T1]
12.55dBm
LN
869.02505010MHz
–57.89dBm
2 [T1]
3 [T1]
0
–10
–20
859.16695500MHz
–81.97dBm
862.00000000MHz
1MAX
1MA
–30
–40
D2 –36dBm
D1 –54dBm
–50
–60
2
–70
3
START 856.5MHz
–80
–85
2.5MHz/
STOP 881.5MHz
Figure 18. Spurious Components—Meets ETSI Specs
Rev. 0 | Page 10 of 28
ADF7012
915 MHZ
–40
RBW
VBW
SWT
RF ATT 40dB
50MHz
50MHz
6.4s
REF LVL
15dBm
10.25dBm
= NORMAL
FREQUENCY = 992.38 kHz
LEVEL = –102.34dBc/Hz
907.81563126MHz
UNIT
dBm
15
10
–60
–80
A
1
0
–10
–20
2
–100
–120
–140
–160
–180
4
3
1MAX
1MA
–30
–40
D1 –41.5dBm
–20.29dBm
2.74188377GHz
–17.50dBm
3.65250501GHz
–50
–60
1 [T1]
2 [T1]
10.25dBm
3 [T1]
4 [T1]
907.81563126MHz
–10.06dBm
–70
–200
1.0k
10.0k
100.0k
1.0M
10.0M
1.83126253GHz
–80
–85
PHASE NOISE (Hz)
CENTER 3.8GHz
640MHz/
SPAN 6.4GHz
Figure 21. Phase Noise Response–ICP = 1.44 mA, IVCO = 3.0 mA, RFOUT = 915.2
MHz, PFD =10 MHz, Power = 10 dBm, PA Bias = 5.5 mA
Figure 24. Harmonic Response, RFOUT Matched to 50 Ω, No Filter
RBW
VBW
SWT
RF ATT 40dB
RBW
VBW
SWT
RF ATT 40dB
10kHz
300kHz
15ms
50MHz
50MHz
6.4s
REF LVL
15dBm
3.88dBm
REF LVL
15dBm
9.06dBm
907.81563126MHz
915.19098196MHz
UNIT
dBm
UNIT
dBm
15
10
15
10
A
A
1
1
0
–10
–20
0
–10
–20
1MAX
1MAX
1MA
1MA
–30
–40
–30
–40
D1 –41.5dBm
3
4
2
–50
–60
–50
–60
1 [T1]
2 [T1]
9.06dBm
3 [T1]
4 [T1]
–46.22dBm
2.74188377GHz
–46.96dBm
907.81563126MHz
–48.40dBm
–70
–70
1.83126253GHz
3.65250501GHz
–80
–85
–80
–85
CENTER 915.190982MHz
50kHz/
SPAN 500kHz
CENTER 3.8GHz
640MHz/
SPAN 6.4GHz
Figure 22. FSK Modulation, Power = 10 dBm, Data Rate = 38.4 kbps,
Fdeviation = 19.2 kHz
Figure 25. Harmonic Response, Fifth-Order Chebyshev Filter
RBW
VBW
SWT
RF ATT 40dB
10kHz
300kHz
100ms
REF LVL
15dBm
9.94dBm
915.23167977MHz
UNIT
dBm
15
10
*
A
1
SGL
0
–10
–20
1MA
–30
–40
D1 –41.5dBm
D1 –49.5dBm
–50
–60
–70
–80
–85
CENTER 915.2MHz
40MHz/
SPAN 400MHz
Figure 23. Spurious Components—Meets FCC Specs
Rev. 0 | Page 11 of 28
ADF7012
CIRCUIT DESCRIPTION
PLL OPERATION
A fractional-N PLL allows multiple output frequencies to be
generated from a single-reference oscillator (usually a crystal)
simply by changing the programmable N value found in the N
register. At the phase frequency detector (PFD), the reference is
compared to a divided-down version of the output frequency
(VCO/N). If VCO/N is too low a frequency, typically the output
frequency is lower than desired, and the PFD and charge-pump
combination sends additional current pulses to the loop filter.
This increases the voltage applied to the input of the VCO.
Because the VCO of the ADF7012 has a positive frequency vs.
voltage characteristic, any increase in the Vtune voltage applied
to the VCO input increases the output frequency at a rate of kV,
the tuning sensitivity of the VCO (MHz/V). At each interval of
1/PFD seconds, a comparison is made at the PFD until the PFD
and charge pump eventually force a state of equilibrium in the
PLL where PFD frequency = VCO/N. At this point, the PLL can
be described as locked.
OSC1
OSC2
CP1
CP2
Figure 27.
Two parallel resonant capacitors are required for oscillation at
the correct frequency—the value of these depend on the crystal
specification. They should be chosen so that the series value of
capacitance added to the PCB track capacitance adds to give the
load capacitance of the crystal, usually 20 pF. Track capacitance
values vary between 2 pF to 5 pF, depending on board layout.
Where possible, to ensure stable frequency operation over all
conditions, capacitors should be chosen so that they have a very
low temperature coefficient and/or opposite temperature
coefficients
CRYSTAL/R
LOOP FILTER
R
CRYSTAL COMPENSATION REGISTER
PFD CP
FVCO
VCO
The ADF7012 features a 15-bit fixed modulus, which allows the
output frequency to be adjusted in steps of FPFD/15. This fine
resolution can be used to easily compensate for initial error and
temperature drift in the reference crystal.
VCO/N
N
FADJUST = FSTEP × FEC
(3)
Figure 26.
where FSTEP = FPFD/215 and FEC = Bits F1 to F11 in the R
Register. Note that the notation is twos compliment, so F11
represents the sign of the FEC number.
FCRYSTAL × N
FOUT
=
= FPFD × N
(1)
(2)
R
Example
For a Fractional N PLL
F
F
F
PFD = 10 MHz
ADJUST = −11 kHz
NFRAC
⎛
⎞
⎟
STEP = 10 MHz/215 = 305.176 Hz
FOUT = FPFD × N
+
⎜
INT
212
⎝
⎠
FEC = −11 kHz/305.17 Hz = −36 = −(00000100100) =
11111011100 = 0x7DC
where NFRAC can be bits M1 to M12 in the fractional N register.
CLOCK OUT CIRCUIT
CRYSTAL OSCILLATOR
The clock out circuit takes the reference clock signal from the
oscillator section above and supplies a divided-down 50:50
mark-space signal to the CLKOUT pin. An even divide from
2 to 30 is available. This divide is set by the DB[19:22] in the R
register. On power-up, the CLKOUT defaults to divide by 16.
The on-board crystal oscillator circuitry (Figure 27) allows an
inexpensive quartz crystal to be used as the PLL reference. The
oscillator circuit is enabled by setting XOEB low. It is enabled by
default on power-up and is disabled by bringing CE low. Errors
in the crystal can be corrected using the error correction
register within the R register.
DV
DD
CLK
ENABLE BIT
OUT
A single-ended reference may be used instead of a crystal, by
applying a square wave to the OSC2 pin, with XOEB set high.
DIVIDER
1 TO 15
OSC1
÷2
CLK
OUT
Figure 28.
Rev. 0 | Page 12 of 28
ADF7012
The output buffer to CLKOUT is enabled by setting Bit DB4 in
the function register high. On power-up, this bit is set high.
The output buffer can drive up to a 20 pF load with a 10ꢀ rise
time at 4.8 MHz. Faster edges can result in some spurious
feedthrough to the output. A small series resistor (50 Ω) can be
Figure 32 shows the VCO gain over temperature and frequency.
VCO gain is important in determining the loop filter design—
predictable changes in VCO gain resulting in a change in the
loop filter bandwidth can be offset by changing the charge-
pump current in software.
used to slow the clock edges to reduce these spurs at FCLK
.
VCO Bias Current
VCO bias current may be adjusted using bits VB1 to VB4 in the
function register. Additional bias current will reduce spurious
levels, but increase overall current consumption in the part. A
bias value of 0x5 should ensure oscillation at most frequencies
and supplies. Settings 0x0, 0xE ,and 0xF are not recommended.
Setting 0x3 and Setting 0x4 are recommended under most
conditions. Improved phase noise can be achieved for lower
bias currents.
LOOP FILTER
The loop filter integrates the current pulses from the charge
pump to form a voltage that tunes the output of the VCO to the
desired frequency. It also attenuates spurious levels generated by
the PLL. A typical loop filter design is shown in Figure 29.
CHARGE
VCO
PUMP OUT
VOLTAGE REGULATORS
There are two band gap voltage regulators on the ADF7012
providing a stable 2.25 V internal supply: a 2.2 µF capacitor
(X5R, NP0) to ground at CREG1 and a 470 nF capacitor at CREG2
should be used to ensure stability. The internal reference
ensures consistent performance over all supplies and reduces
the current consumption of each of the blocks.
Figure 29.
In FSK, the loop should be designed so that the loop bandwidth
(LBW) is a minimum of two to three times the data rate.
Widening the LBW excessively reduces the time spent jumping
between frequencies, but results in reduced spurious
The combination of regulators, band gap reference, and biasing
typically consume 1.045 mA at 3.0 V and can be powered down
by bringing the CE line low. The serial interface is supplied by
Regulator 1, so powering down the CE line causes the contents
of the registers to be lost. The CE line must be high and the
regulators must be fully powered on to write to the serial
interface. Regulator power-on time is typically 100 µs and
should be taken into account when writing to the ADF7012
after power-up. Alternatively, regulator status may be monitored
at the MUXOUT pin once CE has been asserted, because
MUXOUT defaults to the regulator ready signal. Once
Regulator_ready is high, the regulator is powered up and the
serial interface is active.
attenuation. See the section Tips on Designing the Loop Filter.
For OOK/ASK systems, a wider loop bandwidth than for FSK
systems is desirable. The sudden large transition between two
power levels results in VCO pulling (VCO temporarily goes to
incorrect frequency) and can cause a wider output spectrum.
By widening the loop bandwidth a minimum of 10 × data rate,
VCO pulling is minimized because the loop settles quickly back
to the correct frequency. The free design tool ADIsimPLL™ can
be used to design loop filters for the ADI family of transmitters.
VOLTAGE-CONTROLLED OSCILLATOR (VCO)
The ADF7012 features an on-chip VCO with an external tank
inductor, which is used to set the frequency range. The center
frequency of oscillation is governed by the internal varactor
capacitance and that of the external inductor combined with the
bond-wire inductance. An approximation for this is given in the
Equation 4. For a more accurate selection of the inductor, see
the section Choosing the External Inductor Value.
FSK MODULATION
FSK modulation is performed internally in the PLL loop by
switching the value of the N register based on the status of the
TxDATA line. The TxDATA line is sampled at each cycle of the
PFD block (every 1/FPFD seconds). When TxDATA makes a low-
to-high transition, an N value representing the deviation
frequency is added to the N value representing the center
frequency. Immediately the loop begins to lock to the new
frequency of FCENTER + FDEVIATION. Conversely, when TxDATA
makes a high-to-low transition, the N value representing the
deviation is subtracted from the PLL N value representing the
1
F
=
(4)
VCO
2π (LINT + LEXT )×
(CVAR + CFIXED
)
The varactor capacitance can be adjusted in software to increase
the effective VCO range by writing to bits VA1 and VA2 in the
R register. Under typical conditions, setting VA1 and VA2 high
increases the center frequency by reducing the varactor
capacitance by approximately 1.3 pF.
center frequency and the loop transitions to FCENTER − FDEVIATION
.
Rev. 0 | Page 13 of 28
ADF7012
For GFSK and GOOK, the incoming bit stream to be trans-
mitted needs to be synchronized with an on-chip sampling
clock which provides one sample per bit to the Gaussian FIR
filter. To facilitate this, the sampling clock is routed to the
TxCLK pin where data is fetched from the host microcontroller
or microprocessor on the falling edge of TxCLK, and the data is
sampled at the midpoint of each bit on TxCLK’s rising edge.
Inserting external RC LPFs on TxDATA and TxCLK lines
creates smoother edge transitions and improves spurious
performance. As an example, suitable components would be a
1 kV resistor and 10 nF capacitor for a data rate of 5 kbps.
PFD/
CHARGE
PUMP
PA STAGE
4R
VCO
FSK DEVIATION
FREQUENCY
÷N
–F
+F
DEV
THIRD-ORDER
Σ-∆ MODULATOR
DEV
TxDATA
FRACTIONAL-N
INTEGER-N
Figure 30.
I/O
TxDATA
ADF7012
TxCLK
µC
The deviation from the center frequency is set using bits D1 to
D9 in the modulation register. The frequency deviation may be
set in steps of
INT
FETCH
FETCH
FETCH
FETCH
SAMPLE SAMPLE SAMPLE
FPFD
Figure 31. TxCLK/TxDATA Synchronization.
FSTEP (Hz) =
(5)
214
The number of steps between symbol ‘0’ and symbol ‘1’ is
determined by the setting for the index counter.
The deviation frequency is therefore
FPFD × ModulationNumber
The GFSK deviation is set up as
FDEVIATION (Hz) =
(6)
214
where ModulationNumber is set by bits D1 to D9.
FPFD × 2m
GFSKDEVIATION (Hz) =
(7)
(8)
212
The maximum data rate is a function of the PLL lock time (and
the requirement on FSK spectrum). Because the PLL lock time
is reduced by increasing the loop-filter bandwidth, highest data
rates can be achieved for the wider loop filter bandwidths. The
absolute maximum limit on loop filter bandwidth to ensure
stability for a fractional-N PLL is FPFD/7. For a 20 MHz PFD
frequency, the loop bandwidth could be as high as 2.85 MHz.
FSK modulation is selected by setting bits S1 and S2 in the
modulation register low.
where m is the mod control (Bits MC1 to MC3 in the
modulation register).
The GFSK sampling clock samples data at the data rate:
FPFD
DataRate (bps) =
DividerFactor × IndexCounter
where DividerFactor can be bits D1 to D7, and IndexCounter
can be bits IC1 and IC2 in the modulation register.
POWER AMPLIFIER
GFSK MODULATION
The output stage is based on a Class E amplifier design, with an
open drain output switched by the VCO signal. The output
control consists of six current mirrors operating as a
programmable current source.
Gaussian Frequency Shift Keying, or GFSK, represents a filtered
form of frequency shift keying. The data to be modulated to RF
is prefiltered digitally using an finite impulse response filter
(FIR). The filtered data is then used to modulate the sigma-
delta fractional-N to generate spectrally-efficient FSK.
To achieve maximum voltage swing, the RFOUT pin needs to be
biased at DVDD. A single pull-up inductor to DVDD ensures a
current supply to the output stage, PA biased to DVDD volts, and
with the correct choice of value transforms the impedance.
FSK consists of a series of sharp transitions in frequency as the
data is switched from one level to an other. The sharp switching
generates higher frequency components at the output, resulting
in a wider output spectrum.
The output power can be adjusted by changing the value of
bits P1 to P6. Typically, this is P1 to P6 output −20dBm at 0x0,
and 13 dBm at 0x7E at 868MHz, with the optimum matching
network.
With GFSK, the sharp transitions are replaced with up to 128
smaller steps. The result is a gradual change in frequency. As a
result, the higher frequency components are reduced and the
spectrum occupied is reduced significantly. GFSK does require
some additional design work as the data is only sampled once
per bit, and so the choice of crystal is important to ensure the
correct sampling clock is generated.
Rev. 0 | Page 14 of 28
ADF7012
The nonlinear characteristic of the output stage results in an
output spectrum containing harmonics of the fundamental,
especially the third and fifth. To meet local regulations, a low-
pass filter usually is required to filter these harmonics.
As is the case with GFSK, GOOK requires the bit stream
applied at TxDATA to be synchronized with the sampling clock,
TxCLK (see the GFSK Modulation section).
10
The output stage can be powered down by setting Bit PD2 in
the function register low.
0
–10
OOK
GOOK MODULATION
–20
Gaussian on-off keying (GOOK) represents a prefiltered form
of OOK modulation. The usually sharp symbol transitions are
replaced with smooth Gaussian-filtered transitions with the
result being a reduction in frequency pulling of the VCO.
Frequency pulling of the VCO in OOK mode can lead to a
wider than desired bandwidth, especially if it is not possible to
increase the loop filter bandwidth to > 300kHz.
–30
–40
–50
–60
GOOK
–70
–80
909.43
910.43
910.93
The GOOK sampling clock samples data at the data rate:
FREQUENCY (MHz)
FPFD
Figure 33. GOOK vs. OOK Frequency Spectra
(Narrow-Band Measurement)
(9)
DataRate(bps) =
DividerFactor× IndexCounter
20
10
0
Bits D1 to D6 represent the output power for the system for a
positive data bit. Divider Factor = 0x3F represents the max-
imum possible deviation from PA at minimum to PA at
maximum output. An index counter setting of 128 is
recommended.
–10
–20
–30
–40
–50
–60
–70
OOK
Figure 32 shows the step response of the Gaussian FIR filter.
An index counter of 16 is demonstrated for simplicity. While
the pre-filter data would switch the PA directly from off to on
with a low-to-high data transition, the filtered data gradually
increases the PA output in discrete steps. This has the effect of
making the output spectrum more compact.
GOOK
–80
–90
885.43
910.43
FREQUENCY (MHz)
935.93
PRE-FILTER DATA
(0 TO 1 TRANSITION)
PA SETTING
16 (MAX)
15
14
Figure 34. GOOK vs. OOK Frequency Spectra
(Wideband Measurement)
13
12
11
10
9
8
7
6
5
4
DISCRETIZED
FILTER OUTPUT
3
2
1 (PA OFF)
Figure 32. Varying PA Output for GOOK (Index Counter = 16).
Rev. 0 | Page 15 of 28
ADF7012
OUTPUT DIVIDER
Battery Voltage Read back
By setting MUXOUT to 1010 to 1101, the battery voltage can be
estimated. The battery measuring circuit features a voltage
divider and a comparator where the divided-down supply
voltage is compared to the regulator voltage.
An output divider is a programmable divider following the
VCO in the PLL loop. It is useful when using the ADF7012 to
generate frequencies of < 500 MHz.
REFERENCE
DIVIDER
LOOP
FILTER
OUTPUT
DIVIDER
PFD
CP
PA
VCO
Table 6.
÷1/2/4/8
MUXOUT
MUXOUT High
DVDD > 3.25 V
DVDD > 3.0 V
MUXOUT Low
DVDD < 3.25 V
DVDD < 3.0 V
÷N
1010
1011
Figure 35. Output Divider Location in PLL.
1100
1101
DVDD > 2.75 V
DVDD > 2.35 V
DVDD < 2.75 V
DVDD < 2.35 V
The output divider may be used to reduce feedthrough of the
VCO by amplifying only the VCO/2 component, restricting the
VCO feedthrough to leakage.
The accuracy of the measurement is limited by the accuracy of
the regulator voltage and also the internal resistor tolerances.
Regulator Ready
Because the divider is in loop, the N register values should be
set up according to the usual formula. However, the VCO gain
(KV) should be scaled according to the divider setting, as shown
in the following example.
The regulator has a power-up time, dependant on process and
the external capacitor. The regulator ready signal indicates that
the regulator is fully powered, and that the serial interface is
active. This is the default setting on power-up at MUXOUT.
Fout = 433 MHz, Fvco = 866 MHz, KV @ 868 MHz =
60 MHz/V
Digital Lock Detect
Digital lock detect indicates that the status of the PLL loop.
The PLL loop takes time to settle on power-up and when the
frequency of the loop is changed by changing the N value.
When lock detect is high, the PFD has counted a number of
consecutive cycles where the phase error is < 15 ns. The lock
detect precision bit in the function register determines whether
this is 3 cycles (LDP = 0), or 5 cycles (LDP=1). It is recom-
mended that LDP be set to 1. The lock detect is not completely
accurate and goes high before the output has settled to exactly
the correct frequency. In general, add 50ꢀ to the indicated
lock time to obtain lock time to within 1 kHz. The lock detect
signal can be used to decide when the power amplifier (PA)
should be enabled.
Therefore, KV for loop filter design = 30 MHz/V.
The divider value is set in the R register.
Table 5.
OD1
OD2
Divider Status
Divider off
Divide by 2
Divide by 4
Divide by 8
0
0
1
1
0
1
0
1
MUXOUT MODES
The MUXOUT pin allows the user access to various internal
signals in the transmitter, and provides information on the
PLL lock status, the regulator, and the battery voltage. The
MUXOUT is accessed by programming Bits M1 to M4 in the
function register and observing the signal at the MUXOUT pin.
R Divider
MUXOUT provides the output of the R divider. This is a
narrow pulsed digital signal at frequency FPFD. This signal may
be used to check the operation of the crystal circuit and the R
divider. R divider/2 is a buffered version of this signal at FPFD/2.
Rev. 0 | Page 16 of 28
ADF7012
THEORY OF OPERATION
CHOOSING THE EXTERNAL INDUCTOR VALUE
Standard Crystal Values
Standard crystal values are 3.6864 MHz, 4 MHz, 4.096 MHz,
4.9152 MHz, 7.3728 MHz, 9.8304 MHz, 10 MHz, 11.0592 MHz,
12 MHz, and 14.4792 MHz. Crystals with these values are
usually available in stock and cost less than crystals with
nonstandard values.
The ADF7012 allows operation at many different frequencies by
choosing the external VCO inductor to give the correct output
frequency. Figure 36 shows both the minimum and maximum
frequency vs. the inductor value. These are measurements based
on 0603 CS type inductors from Coilcraft, and are intended as
guidelines in choosing the inductor because board layout and
inductor type varies between applications.
Reference Spurious Levels
Reference spurious levels (spurs) occur at multiples of the PFD
frequency. The reference spur closest to the carrier is usually
highest with the spur further out being attenuated by the loop
filter. The level of reference spur is lower for lower PFD
frequencies. In designs with high output power where spurious
levels are the main concern, a lower PFD frequency (<5 MHz)
may be desirable.
The inductor value should be chosen so it is between the
minimum and maximum value.
1200
MIN (meas)
1100
MAX (meas)
MIN (eqn)
1000
MAX (eqn)
Beat Note Spurs
900
These are spurs occurring for very small or very large values in
the fractional register. These are quickly attenuated by the loop
filter. Selection of the PFD therefore determines their location,
and ensures that they have negligible effect on the transmitter
spectrum.
800
700
600
500
Phase Noise
400
300
The phase noise of a frequency synthesizer improves by 3dB for
every doubling of the PFD frequency. Because ACP is related to
the phase noise, the PFD may be increased to reduce the ACP
in the system. PFD frequencies of < 5MHz typically deliver
sufficient phase noise performance for most systems.
0
5
10
15
20
25
30
35
INDUCTANCE (nH)
Figure 36. Output Frequency vs. External Inductor Value
Ibias = 2.0 mA.
For frequencies between 270 MHz and 550 MHz, it is
Deviation Frequency
recommended to operate the VCO at twice the desired output
frequency and use the divide-by-2 option. This ensures reliable
operation over temperature and supply.
The deviation frequency is adjustable in steps of
FPFD
214
FSTEP (Hz) =
(10)
For frequencies between 130 MHz and 270 MHz, it is
recommended to operate the VCO at four times the desired
output frequency and use the divide-by-4 option.
To get the exact deviation frequency required, ensure FSTEP is a
factor of the desired deviation.
For frequencies below 130 MHz, it is best to use the divide-by-8
option. It is not necessary to use the VCO divider for
frequencies above 550 MHz.
TIPS ON DESIGNING THE LOOP FILTER
The loop filter design is crucial in ensuring stable operation of
the transmitter, meeting Adjacent Channel Power (ACP)
specifications, and meeting spurious requirements for the
relevant regulations. ADIsimPLL is a free tool available to aid
the design of loop filters. The user enters the desired frequency
range, the reference crystal and PFD values, and the desired
loop bandwidth. ADIsimPLL gives a good starting point for the
filter, and the filter can be further optimized based on the
criteria below.
ADIsimPLL is a PLL design tool which can perform the
frequency calculations for the ADF7012, and is available at
www.analog.com/pll.
CHOOSING THE CRYSTAL/PFD VALUE
The choice of crystal value is an important one. The PFD
frequency must be the same as the crystal value or an integer
division of it. The PFD determines the phase noise, spurious
levels and location, deviation frequency, and the data rate in the
case of GFSK. The following sections describe some factors that
should be considered when choosing the crystal value.
Rev. 0 | Page 17 of 28
ADF7012
Setting Tuning Sensitivity Value
PA MATCHING
The ADF7012 exhibits optimum performance in terms of
transmit power and current consumption only if the RF output
port is properly matched to the antenna impedance.
The tuning sensitivity or kV is usually denoted in MHz/V and is
required for the loop filter design. It refers to the amount that a
change of a volt in the voltage applied to VCOIN pin, changes
the output frequency. Typical data for the ADF7012 over a
frequency range is shown.
ZOPT_PA depends primarily on the required output power,
and the frequency range. Selecting the optimum ZOPT_PA
helps to minimize the current consumption. This data sheet
contains a number of matching networks for common
frequency bands. Under certain conditions it is recommended
to obtain a suitable ZOPT_PA value by means of a load-pull
measurement.
120
100
80
60
DV
DD
40
RF
OUT
20
0
PA
ANTENNA
LPF
ZOPT_PA
200
300
400
500
600
700
800
900 1000 1100
FREQUENCY (MHz)
Figure 37. kV vs. VCO Frequency
Figure 38. ADF7012 with Harmonic Filter
Charge-Pump Current
The impedance matching values provided in the next section
are for 50 Ω environments. An additional matching network
may be required after the harmonic filter to match to the
antenna impedance. This can be incorporated into the filter
design itself in order to reduce external components.
The charge-pump current allows the loop filter bandwidth to be
changed using the registers. The loop bandwidth reduces as the
charge pump current is reduced and vice versa.
Selecting Loop Filter Bandwidth
Data Rate
TRANSMIT PROTOCOL AND CODING
CONSIDERATIONS
The loop filter bandwidth should usually be at two to three
times the data rate. This ensures that the PLL has ample time
to jump between the mark and space frequencies.
SYNC
ID
PREAMBLE
WORD
FIELD
DATA FIELD
CRC
ACP
Figure 39. Typical Format of a Transmit Protocol
In the case where the ACP specifications are difficult to meet,
the loop filter bandwidth can be reduced further to reduce the
phase noise at the adjacent channel. The filter rolls off at 20 dB
per decade.
A dc-free preamble pattern such as 10101010… is recom-
mended for FSK/ASK/OOK demodulation. Preamble patterns
with longer run-length constraints such as 11001100…. can also
be used. However, this can result in a longer synchronization
time of the received bit stream in the chosen receiver.
Spurious Levels
In the case where the output power is quite high, a reduced loop
filter bandwidth reduces the spurious levels even further, and
provides additional margin on the specification.
The following sections provide examples of loop filter designs
for typical applications in specific frequencies.
Rev. 0 | Page 18 of 28
ADF7012
APPLICATION EXAMPLES
R2
C1
C2
R1
C3
V
DD
U1
ADF7012
C5
2.2µF
C5
100pF
C10
470nF
C5+
10µF
C11
0.22µF
C
REG2
V
DD
1
2
24
DV
DD
C
REG2
C
C6
2.2µF
REG1
R9
3.6kΩ
23
C
R
SET
REG1
3
4
22
21
CP
OUT
AGND
C12
100pF
C13
2.2µF
V
J1
DD
TxDATA
1kΩ
TxDATA
TxCLK
TxDATA
TxCLK
DV
DD
L1
J4
1kΩ
5
6
7
8
20
19
18
17
C14
L2
LF1
CF1
LF2
RF
TxCLK
OUT
C15
1kΩ
MUXOUT
CF2
CF2
MUXOUT
RF
MUXOUT
Y1
GND
DGND
OSC1
OSC2
VCO
IN
C7
27pF
C8
27pF
OSC1
OSC2
C
VCO
L2
J2
C9
9
16
15
L3
1kΩ
CLK
OUT
10
CLK
OUT
CLK
OUT
L1
CE
LE
R3
CLK
CLK
1kΩ
11
12
14
13
V
DD
J3–3
J3–5
CLK
R4
DATA
DATA
LE
1kΩ
DATA
CE
R5
LE
1kΩ
J3–7
J3–6
J3–8
TxDATA
J5–1
J5–2
J5–4
J5–5
J5–8
TxCLK
CLK
MUXOUT
CLK
J5–3
J5–5
J5–7
OUT
R6
1kΩ
R7
1kΩ
R8
1kΩ
DATA
CE
LE
V
J5–9 J5–10
DD
10 PIN HEADER (5X2)
Figure 40. Applications Diagrams with Harmonic Filter
Rev. 0 | Page 19 of 28
ADF7012
315 MHZ OPERATION
Bias Current
The recommendations here are guidelines only. The design
should be subject to internal testing prior to FCC site testing.
Matching components need to be adjusted for board layout.
Because low current is desired, a 2.0 mA VCO bias can be used.
Additional bias current reduces any spur, but increases current
consumption.
The FCC standard 15.231 regulates operation in the band
from 260MHz to 470MHz in the US. This is used generally in
the transmission of RF control signals, such as in a satellite-
decoder remote control, or remote keyless entry system. The
band cannot be used to send any continuous signal. The
maximum output power allowed is governed by the duty cycle
of the system. A typical design example for a remote control is
shown next.
The PA bias can be set to 5.5 mA and achieve 0 dBm.
Loop Filter Bandwidth
The loop filter is designed with ADIsimPLL Version 2.5. The
loop bandwidth design is straightforward because the 20 dB
bandwidth is generally of the order of >400 kHz (0.25ꢀ of
center frequency). A loop bandwidth of close to 100 kHz strikes
a good balance between lock time and spurious suppression. If
it is found that pulling of the VCO is more than desired in OOK
mode, the bandwidth could be increased.
Design Criteria
315 MHz center frequency
FSK/OOK modulation
1 mW output power
House range
Design of Harmonic Filter
The main requirement of the harmonic filter should ensure that
the third harmonic level is < −41.5 dBm. A fifth-order
Chebyshev filter is recommended to achieve this, and a
suggested starting point is given next. The Pi format is chosen
to minimize the more expensive inductors.
Meets FCC 15.231
The main requirements in the design of this remote are a long
battery life and sufficient range. It is possible to adjust the
output power of the ADF7012 to increase the range depending
on the antenna performance.
Component Values—Crystal: 3.6864MHz
Loop Filter
ICP
0.866 mA
100 kHz
680 pF
12 nF
220 pF
1.1 kV
3 kV
The center frequency is 315 MHz. Because the ADF7012
VCO is not recommended for operation in fundamental mode
for frequencies below 400 MHz, the VCO needs to operate at
630 MHz. Figure 36 (Output Frequency vs. External Inductor
Value) implies an inductor value of 7.6 nH or close to this. The
chip inductor chosen = 7.5 nH (0402CS-7N5 from Coilcraft).
Coil inductors are recommended to provide sufficient Q for
oscillation.
LBW
C1
C2
C3
R1
R2
Matching
L1
L2
C14
C15
56 nH
1 nF
Short
Crystal and PFD
Phase noise requirements are not excessive as the adjacent
channel power requirement is −20 dB. The PFD is chosen so as
to minimize spurious levels (beat note and reference), and to
ensure a quick crystal power-up time.
Open
Harmonic Filter
L4
L5
CF1
CF2
CF3
22 nH
22 nH
3.3 pF
8.2 pF
3.3 pF
PFD = 3.6864 MHz − Power-Up Time 1.6ms. Figure 10 shows a
typical power-on time for a 4 MHz crystal.
N-Divider
The N Divider is determined as being:
Nint = 85
Nfrac = (1850)/4096
VCO divide-by-2 is enabled
Deviation
The deviation is set to 50 kHz so as to accommodate a simple
receiver architecture.
The modulation steps available are in 3.6864 MHz/214 :
Modulation steps = 225 Hz
Modulation number = 50 kHz/225 Hz = 222
Rev. 0 | Page 20 of 28
ADF7012
433 MHZ OPERATION
Loop Filter Bandwidth
The recommendations here are guidelines only. The design
should be subject to internal testing prior to ETSI site testing.
Matching components need to be adjusted for board layout.
The loop filter is designed with ADIsimPLL Version 2.5. The
loop bandwidth design requires that the channel power be
< −36 dBm at 870 kHz from the center. A loop bandwidth of
close to 160 kHz strikes a good balance between lock time for
data rates, including 32 kbps and spurious suppression. If it is
found that pulling of the VCO is more than desired in OOK
mode, the bandwidth could be increased.
The ETSI standard EN 300-220 governs operation in the
433.050 MHz to 434.790 MHz band. For many systems, 10ꢀ
duty is sufficient for the transmitter to output 10 dBm.
Design Criteria
433.92 MHz center frequency
FSK modulation
10 mW output power
200 m range
Design of Harmonic Filter
The main requirement of the harmonic filter should ensure
that the third harmonic level is < −30 dBm. A fifth-order
Chebyshev filter is recommended to achieve this, and a
suggested starting point is given next. The Pi format is chosen
to minimize the more expensive inductors.
Meets ETSI 300-220
The main requirement in the design of this remote is a long
battery life and sufficient range. It is possible to adjust the
output power of the ADF7012 to increase the range depending
on the antenna performance.
Component Values—Crystal: 4.9152 MHz
Loop Filter
Icp
LBW
C1
2.0 mA
100 kHz
680 pF
12 nF
The center frequency is 433.92 MHz. It is possible to operate the
VCO at this frequency. Figure 36 shows the inductor value vs.
center frequency. The inductor chosen is 22 nH. Coilcraft
inductors such as 0603-CS-22NXJBU are recommended.
C2
C3
270 pF
R1
R2
910 V
3.3 kV
Crystal and PFD
The phase noise requirement is such to ensure the power at the
edge of the band is < −36 dBm. The PFD is chosen so as to
minimize spurious levels (beat note and reference), and to
ensure a quick crystal power-up time.
Matching
L1
22 nH
L2
C14
C15
10 pF
Short
Open
PFD = 4.9152 MHz − Power-Up Time 1.6 ms. Figure 10 shows a
typical power-up time for a 4 MHz crystal.
Harmonic Filter
N-Divider
L4
L5
CF1
CF2
CF3
22 nH
22 nH
3.3 pF
8.2 pF
3.3 pF
The N Divider is determined as being:
Nint = 88
Nfrac = (1152)/4096
VCO divide-by-2 is not enabled
Deviation
The deviation is set to 50 kHz so as to accommodate a simple
receiver architecture.
The modulation steps available are in 4.9152 MHz/214 :
Modulation steps = 300 Hz
Modulation number = 50 kHz/300Hz = 167
Bias Current
Because low current is desired, a 2.0 mA VCO bias can be used.
Additional bias current reduces any spurious, but increases
current consumption.
The PA bias can be set to 5.5 mA and achieve 10 dBm.
Rev. 0 | Page 21 of 28
ADF7012
868 MHZ OPERATION
The recommendations here are guidelines only. The design
should be subject to internal testing prior to ETSI site testing.
Matching components need to be adjusted for board layout.
The modulation steps available are in 4.9152 MHz/214 :
Modulation steps = 300 Hz
Modulation number = 19.2 kHz/300 Hz = 64.
Bias Current
The ETSI standard EN 300-220 governs operation in the
868 MHz to 870MHz band. The band is broken down into
several subbands each having a different duty cycle and output
power requirement. Narrowband operation is possible in the
50kHz channels, but both the output power and data rate are
limited by the −36 dBm adjacent channel power specification.
There are many different applications in this band, including
remote controls for security, sensor interrogation, metering
and home control.
Because low current is desired, a 2.5 mA VCO bias can be used.
Additional bias current reduces any spurious, but increases
current consumption. A 2.5 mA bias current gives the best
spurious vs. phase noise trade-off.
The PA bias should be set to 7.5 mA to achieve 12 dBm.
Loop Filter Bandwidth
The loop filter is designed with ADIsimPLL Version 2.5. The
loop bandwidth design requires that the channel power be
< −36 dBm at 250 kHz from the center. A loop bandwidth of
close to <60 kHz is required to bring the phase noise at the edge
of the band sufficiently low to meet the ACP specification. This
represents a compromise between the data rate requirement and
the phase noise requirement.
Design Criteria
868.95 MHz center frequency (band 868.7MHz − 869.2 MHz)
FSK modulation
12 dBm output power
300 m range
Meets ETSI 300-220
38.4 kbps data rate
Design of Harmonic Filter
The main requirement of the harmonic filter should ensure that
the second and third harmonic levels are < −30 dBm. A fifth-
order Chebyshev filter is recommended to achieve this, and a
suggested starting point is given next. The Pi format is chosen
to minimize the more expensive inductors.
The design challenge is to enable the part to operate in this
particular subband and meet the ACP requirement 250 kHz
away from the center.
The center frequency is 868.95 MHz. It is possible to operate the
VCO at this frequency. Figure 31 shows the inductor value vs.
center frequency. The inductor chosen is 1.9 nH. Coilcraft
inductors such as 0402-CS-1N9XJBU are recommended.
Component Values—Crystal: 4.9152 MHz
Loop Filter
Icp
LBW
C1
1.44 mA
60 kHz
1.5 nF
22 nF
Crystal and PFD
The phase noise requirement is such to ensure the power at
the edge of the band is < −36 dBm. This requires close to
−100 dBc/Hz phase noise at the edge of the band.
C2
C3
560 pF
R1
390 V
The PFD is chosen so as to minimize spurious levels (beat note
and reference), and to ensure a quick crystal power-up time. A
PFD of < 6 MHz places the largest PFD spur at a frequency of
greater than 862 MHz, and so reduces the requirement on the
spur level to −36 dBm instead of −54 dBm.
R2
910 V
Matching
L1
L2
C14
27 nH
6.2 nH
470 pF
Open
PFD = 4.9152 MHz − Power Up-Time 1.6 ms. Figure 10 shows a
typical power-on time for a 4MHz crystal.
C15
Harmonic Filter
L4
L5
CF1
CF2
CF3
N-Divider
8.2 nH
8.2 nH
4.7 pF
6.8 pF
4.7 pF
The N divider is determined as being:
Nint = 176
Nfrac = (3229)/4096
VCO divide-by-2 is not enabled.
Deviation
The deviation is set to 19.2 kHz so as to accommodate a
simple receiver architecture and also ensure that the
modulation spectrum is narrow enough to meet the adjacent
channel power (ACP) requirements.
Rev. 0 | Page 22 of 28
ADF7012
915 MHZ OPERATION
Deviation
The recommendations here are guidelines only. The design
should be subject to internal testing prior to FCC site testing.
Matching components need to be adjusted for board layout.
The deviation is set to 19.2 kHz so as to accommodate a
simple receiver architecture, and also to ensure the available
spectrum is used efficiently.
FCC 15.247 and FCC 15.249 are the main regulations governing
operation in the 902 MHz to 928 MHz Band. FCC 15.247
requires some form of spectral spreading. Typically, the
ADF7012 would be used in conjunction with the frequency
hopping spread spectrum (FHSS) or it may be used in
conjunction with the digital modulation standard which
requires large deviation frequencies. Output power of < 1 W
is tolerated on certain spreading conditions.
The modulation steps available are in 10 MHz/214 :
Modulation steps = 610 Hz
Modulation number = 19.2 kHz/610 Hz = 31.
Bias Current
Because low current is desired, a 3 mA VCO bias can be used
and still ensure oscillation at 928 MHz. Additional bias current
reduces any spurious noise, but increases current consumption.
A 3 mA bias current gives the best spurious vs. phase noise
trade-off.
Compliance with FCC 15.249 limits the output power to
−1.5 dBm, but does not require spreading. There are many
different applications in this band, including remote controls
for security, sensor interrogation, metering, and home control.
The PA bias should be set to 5.5 mA to achieve 10 dBm power.
Loop Filter Bandwidth
Design Criteria
915.2MHz center frequency
FSK modulation
10 dBm output power
200 m range
The loop filter is designed with ADIsimPLL Version 2.5. A
data rate of 170 kHz is chosen, which allows for data rates of
> 38.4 kbps. It also attenuates the beat note spurs quickly to
ensure they have no effect on system performance.
Meets FCC 15.247
38.4 kbps data rate
Design of Harmonic Filter
The main requirement of the harmonic filter should ensure
that the third harmonic level is < −41.5 dBm. A fifth-order
Chebyshev filter is recommended to achieve this, and a
suggested starting point is given next. The Pi format is chosen
to minimize the number of inductors in the system.
The center frequency is 915.2 MHz. It is possible to operate
the VCO at this frequency. Figure 36 shows the inductor value
vs. center frequency. The inductor chosen is 1.6 nH. Coilcraft
inductors such as 0603-CS-1N6XJBU are recommended.
Additional hopping frequencies can easily be generated by
changing the N value.
Component Values—Crystal: 10 MHz
Loop Filter
Crystal and PFD
Icp
LBW
C1
1.44 mA
170 kHz
470 pF
12 nF
The phase noise requirement is such to ensure that the 20 dB
bandwidth requirements are met. These are dependant on the
channel spacing chosen. A typical channel spacing would be
400 kHz, which would allow 50 channels in 20 MHz and enable
the design to avoid the edges of the band.
C2
C3
R1
120 pF
470 V
R2
1.8 kV
The PFD is chosen so as to minimize spurious levels. There are
beat note spurious levels at 910 MHz and 920 MHz, but the
level is usually significantly less than the modulation power.
They are also attenuated quickly by the loop filter to ensure a
quick crystal power-up time.
Matching
L1
L2
C14
C15
27 nH
6.2 nH
470 pF
Open
PFD = 10 MHz − Power-Up Time 1.8 ms (approximately).
Figure 10 shows a typical power-on time for a 4 MHz crystal.
Harmonic Filter
L4
L5
CF1
CF2
CF3
8.2 nH
8.2 nH
4.7 pF
6.8 pF
4.7 pF
N-Divider
The N divider is determined as being:
Nint = 91
Nfrac = (2130)/4096
VCO divide-by-2 is not enabled
Rev. 0 | Page 23 of 28
ADF7012
REGISTER DESCRIPTIONS
R REGISTER
OUTPUT
DIVIDER ADJUST
VCO
CLOCK OUT
DIVIDER
4-BIT R DIVIDER
11-BIT FREQUENCY ERROR CORRECTION
F-COUNTER
OFFSET
D1 CRYSTAL DOUBLER
F11 ....... F3
F2
F1
0
1
CRYSTAL DOUBLER OFF
CRYSTAL DOUBLER ON
+1023
+1022
.
+1
+0
0
0
0
0
0
.......
.......
.......
.......
.......
1
1
.
0
0
0
0
.
0
0
0
1
.
1
0
X1 XOEB
OD2 OD1
OUTPUT DIVIDER
0
1
XTAL OSCILLATOR ON (DEFAULT)
XTAL OSCILLATOR OFF
–1
–0
–1
–1023
–1024
1
1
.
1
1
.......
.......
.......
.......
.......
1
1
.
0
0
1
1
.
0
0
1
0
.
1
0
0
0
1
1
0
1
0
1
DISABLED
DIVIDE BY 2
DIVIDE BY 4
DIVIDE BY 8
e.g., F-COUNTER OFFSET = 1, FRACTIONAL
15
OFFSET = 1/2
VA2 VA1
VCO ADJUST
NO VCO ADJUSTMENT
0
0
1
1
0
1
0
1
(F
(F
(F
– (1 × f))
– (2 × f))
– (3 × f))
CENTER
CENTER
CENTER
RF R COUNTER
RL4
RL3
RL2
RL1
DIVIDE RATIO
1
0
0
0
0
.
0
0
0
1
.
0
1
1
0
.
1
0
1
0
.
2
3
4
.
CLK
OUT
CL4
CL3
CL2
CL1
DIVIDE RATIO
2
0
0
0
0
.
0
0
0
1
.
0
1
1
0
.
1
0
1
0
.
4
.
.
.
.
.
.
6
.
.
.
.
8
12
13
14
15
1
1
1
1
1
1
1
1
0
0
1
1
0
1
0
1
.
.
.
.
.
16 (DEFAULT)
.
.
.
.
.
28
1
1
1
1
1
1
0
1
30
Figure 41.
Rev. 0 | Page 24 of 28
ADF7012
N-COUNTER LATCH
8-BIT INTEGER-N
12-BIT FRACTIONAL-N
MODULUS
DIVIDE RATIO
M12 M11
M10 ....... M3
M2
M1
0
1
2
.
.
.
4092
4093
4094
4095
0
0
0
.
.
.
1
1
1
1
0
0
0
.
.
.
1
1
1
1
0
0
0
.
.
.
1
1
1
1
.......
.......
.......
.......
.......
.......
.......
.......
.......
.......
0
0
0
.
.
.
1
1
1
1
0
0
1
.
.
.
0
0
1
1
0
1
0
.
.
.
0
1
0
1
e.g., SETTING F = 0 IN FSK MODE TURNS ON THE Σ-∆ WHILE
THE PLL IS AN INTEGER VALUE
N-COUNTER
N8
0
0
0
.
N7
0
0
0
.
N6
0
0
0
.
N5
0
0
0
.
N4
0
0
0
.
N3
0
0
0
.
N2
0
0
1
.
N1
0
1
0
.
DIVIDE RATIO
1
2
3
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
254
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
1
255
e.g., MODULUS DIVIDE RATIO = 2048 - > 1/2
P1 PRESCALER
THE N-VALUE CHOSEN IS A MINIMUM OF
2
P
+ 3P + 3. FOR PRESCALER 8/9 THIS
0
1
4/5
8/9
MEANS A MINIMUM N-DIVIDE OF 91.
Figure 42.
Rev. 0 | Page 25 of 28
ADF7012
MODULATION REGISTER
GFSK MOD
CONTROL
TEST BITS
MODULATION DEVIATION
POWER AMPLIFIER
MUST BE LOW
G1 GAUSSIAN OOK
ON
OFF
0
1
MODULATION
SCHEME
S2
S1
0
0
1
1
0
1
0
1
FSK
GFSK
ASK
OOK
IF AMPLITUDE SHIFT KEYING SELECTED, TxDATA = 0
POWER AMPLIFIER OUTPUT LEVEL
D6
0
.
.
.
D2
X
0
0
1
.
D1
X
0
1
0
.
P6
0
0
0
0
.
.
.
P2
X
0
1
1
.
P1
X
1
0
1
.
PA OFF
PA OFF
.
.
.
.
.
.
–16.0dBm
1/31 * 14dBm
2/31 * 14dBm
.
.
.
–16.0dBm
1/31 * 14dBm
2/31 * 14dBm
.
0
0
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
14dBm
13dBm
1
1
1
1
1
1
1
1
1
1
IF FREQUENCY SHIFT KEYING SELECTED
14
/2
STEP = PFD
F
F
D9
0
....... D3
D2
0
D1
0
1
0
1
.
F DEVIATION
PLL MODE
.......
.......
.......
.......
.......
.......
0
0
0
0
.
0
0
1 × –F
2 × –F
3 × –F
.......
STEP
STEP
STEP
0
1
0
1
.
.
1
1
1
.
511 × –F
STEP
IF GAUSSIAN FREQUENCY SHIFT KEYING SELECTED
IC2
0
IC1
0
INDEX COUNTER
D7
0
.......
.......
.......
.......
.......
.......
.......
D3
0
D2
0
D1
0
DIVIDER FACTOR
16
0
0
1
32
0
0
0
1
1
1
0
64
0
0
1
0
2
1
1
128
0
0
1
1
3
.
.
.
.
.......
127
MC3 MC2 MC1 GFSK MOD CONTROL
1...
1
1
1
0
0
.
0
0
.
0
1
.
0
1
.
1
1
1
7
Figure 43.
Rev. 0 | Page 26 of 28
ADF7012
FUNCTION REGISTER
SD TEST
MODES
PLL TEST
MODES
PA BIAS
VCO BIAS
MUXOUT
BLEED CHARGE
CURRENT PUMP
I1 DATA INVERT
0
1
DATA
DATA
CP4 BLEED DOWN
PD1 PLL ENABLE
PA3 PA2 PA1 PA BIAS
5µA
0
0
0
.
0
0
1
.
0
1
0
.
0
1
BLEED OFF
BLEED ON
0
1
PLL OFF
PLL ON
6µA
7µA
.
CP3 BLEED UP
PD2 PA ENABLE
.
.
.
.
0
1
BLEED OFF
BLEED ON
0
1
PAOFF
PAON
1
1
1
12µA
CHARGE PUMP
CURRENT
CP2 CP1
0
0
1
1
0
1
0
1
0.3mA
0.9mA
1.5mA
2.1mA
PD3 CLK
OUT
VD1 VCO DISABLE
VCO BIAS
VB4 VB3 VB2 VB1 CURRENT
0
1
VCO ON
VCO OFF
0
CLK
CLK
OFF
ON
OUT
OUT
0.5mA
0
0
.
0
0
.
0
1
.
1
0
.
1
1mA
.
1
1
1
1
8mA
M4
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
M3
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
M2
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
M1
MUXOUT
LOGIC LOW
LOGIC HIGH
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
INVALID MODE – DO NOT USE
REGULATOR READY (DEFAULT)
DIGITAL LOCK DETECT
ANALOG LOCK DETECT
R DIVIDER/2 OUTPUT
N DIVIDER/2 OUTPUT
RF R DIVIDER OUTPUT
DATA RATE
BATTERY MEASURE IS > 3.25V
BATTERY MEASURE IS > 3V
BATTERY MEASURE IS > 2.75V
BATTERY MEASURE IS > 2.35V
NORMAL TEST MODES
Σ-∆ TEST MODES
Figure 44.
Rev. 0 | Page 27 of 28
ADF7012
OUTLINE DIMENSIONS
7.90
7.80
7.70
24
13
12
4.50
4.40
4.30
6.40 BSC
1
PIN 1
0.65
BSC
1.20
MAX
0.15
0.05
0.75
0.60
0.45
8°
0°
0.30
0.19
0.20
0.09
SEATING
PLANE
0.10 COPLANARITY
COMPLIANT TO JEDEC STANDARDS MO-153AD
Figure 45. 24-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-24)
ORDERING GUIDE
Model
Temperature Range
Package Description
Package Option
Frequency Range
ADF7012BRU
TSSOP
RU-24
RU-24
RU-24
50 MHz to 1 GHz
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
ADF7012BRU-REEL
ADF7012BRU-REEL7
EVAL-ADF7012EB1
EVAL-ADF7012EB2
EVAL-ADF7012EB3
EVAL-ADF7012EB4
EVAL-ADF7012EB5
TSSOP, 13”REEL
TSSOP, 7” REEL
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
50 MHz to 1 GHz
50 MHz to 1 GHz
902 MHz to 928 MHz
860 MHz to 880 MHz
418 MHz to 435 MHz
310 MHz to 330 MHz
50 MHz to 1 GHz
©
2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D04617–0–10/04(0)
Rev. 0 | Page 28 of 28
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