AD7730LBR-REEL [ADI]
IC 2-CH DELTA-SIGMA ADC, SERIAL ACCESS, PDSO24, SOIC-24, Analog to Digital Converter;型号: | AD7730LBR-REEL |
厂家: | ADI |
描述: | IC 2-CH DELTA-SIGMA ADC, SERIAL ACCESS, PDSO24, SOIC-24, Analog to Digital Converter 传感器 |
文件: | 总52页 (文件大小:499K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
a
Bridge Transducer ADC
AD7730/AD7730L
T he modulator output is processed by a low pass programmable
digital filter, allowing adjustment of filter cutoff, output rate and
settling time.
KEY FEATURES
Resolution of 230,000 Counts (Peak-to-Peak)
Offset Drift: 5 nV/ ؇C
Gain Drift: 2 ppm / ؇C
T he part features two buffered differential programmable gain
analog inputs as well as a differential reference input. T he part
operates from a single +5 V supply. It accepts four unipolar
analog input ranges: 0 mV to +10 mV, +20 mV, +40 mV and
+80 mV and four bipolar ranges: ±10 mV, ±20 mV, ±40 mV
and ±80 mV. T he peak-to-peak resolution achievable directly
from the part is 1 in 230,000 counts. An on-chip 6-bit DAC
allows the removal of T ARE voltages. Clock signals for synchro-
nizing ac excitation of the bridge are also provided.
Line Frequency Rejection: >150 dB
Buffered Differential Inputs
Program m able Filter Cutoffs
Specified for Drift Over Tim e
Operates w ith Reference Voltages of 1 V to 5 V
ADDITIONAL FEATURES
Tw o-Channel Program m able Gain Front End
On-Chip DAC for Offset/ TARE Rem oval
FASTStep™ Mode
AC or DC Excitation
Single Supply Operation
T he serial interface on the part can be configured for three-wire
operation and is compatible with microcontrollers and digital
signal processors. T he AD7730 contains self-calibration and
system calibration options, and features an offset drift of less
than 5 nV/°C and a gain drift of less than 2 ppm/°C.
APPLICATIONS
Weigh Scales
Pressure Measurem ent
T he AD7730 is available in a 24-pin plastic DIP, a 24-lead
SOIC and 24-lead T SSOP package. T he AD7730L is available
in a 24-lead SOIC and 24-lead T SSOP package.
GENERAL D ESCRIP TIO N
T he AD7730 is a complete analog front end for weigh-scale and
pressure measurement applications. T he device accepts low-
level signals directly from a transducer and outputs a serial
digital word. T he input signal is applied to a proprietary pro-
grammable gain front end based around an analog modulator.
NO TE
T he description of the functions and operation given in this data
sheet apply to both the AD7730 and AD7730L. Specifications
and performance parameters differ for the parts. Specifications
for the AD7730L are outlined in Appendix A.
FUNCTIO NAL BLO CK D IAGRAM
REF IN(–)
DV
REF IN(+)
AV
DD
DD
AD7730
REFERENCE DETECT
VBIAS
AV
100nA
DD
STANDBY
SYNC
AIN1(+)
AIN1(–)
SIGMA-DELTA A/D CONVERTER
BUFFER
SIGMA-
DELTA
PROGRAMMABLE
DIGITAL
+
MODULATOR
FILTER
MUX
PGA
+/–
100nA
AGND
AIN2(+)/D1
AIN2(–)/D0
MCLK IN
CLOCK
GENERATION
6-BIT
DAC
SERIAL INTERFACE
AND CONTROL LOGIC
MCLK OUT
REGISTER BANK
SCLK
CS
CALIBRATION
MICROCONTROLLER
DIN
ACX
AC
DOUT
EXCITATION
CLOCK
ACX
AGND
DGND
POL
RDY
RESET
FASTStep is a trademark of Analog Devices, Inc.
REV. A
Inform ation furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assum ed by Analog Devices for its
use, nor for any infringem ents of patents or other rights of third parties
which m ay result from its use. No license is granted by im plication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norw ood, MA 02062-9106, U.S.A.
Tel: 781/ 329-4700
Fax: 781/ 326-8703
World Wide Web Site: http:/ / w w w .analog.com
© Analog Devices, Inc., 1998
AD7730–SPECIFICATIONS (0AVV; fC=LK I+N5=V4,.D9V152=M+Hz3.VAlolrsp+e5cVif;icRaEtFioInNs(+TM)IN=toAVTMA;XRuEnFleIsNs(–o)th=erAwGiNseDn=otDeGdN.)D =
DD
DD
DD
P aram eter
B Version1
Units
Conditions/Com m ents
ST AT IC PERFORMANCE (CHP = 1)
No Missing Codes2
24
Bits min
Output Noise and Update Rates2
Integral Nonlinearity
See Tables I & II
18
See Note 3
5
ppm of FSR max
Offset Error2
Offset Error and Offset Drift Refer to Both
Unipolar Offset and Bipolar Zero Errors
Offset Drift vs. T emperature2
Offset Drift vs. T ime4
nV/°C typ
nV/1000 Hours typ
25
Positive Full-Scale Error2, 5
Positive Full-Scale Drift vs T emp2, 6, 7
Positive Full-Scale Drift vs T ime4
Gain Error2, 8
See Note 3
2
10
See Note 3
2
10
ppm of FS/°C max
ppm of FS/1000 Hours typ
Gain Drift vs. T emperature2, 6, 9
Gain Drift vs. T ime4
ppm/°C max
ppm/1000 Hours typ
Bipolar Negative Full-Scale Error2
Negative Full-Scale Drift vs. T emp2, 6
Power Supply Rejection
See Note 3
2
ppm of FS/°C max
dB typ
dB min
nA max
pA/°C typ
nA max
120
120
50
100
10
Measured with Zero Differential Voltage
At DC. Measured with Zero Differential Voltage
Common-Mode Rejection (CMR)
Analog Input DC Bias Current2
Analog Input DC Bias Current Drift2
Analog Input DC Offset Current2
Analog Input DC Offset Current Drift2
50
pA/°C typ
ST AT IC PERFORMANCE (CHP = 0)2
No Missing Codes
24
Bits min
SKIP = 010
Output Noise and Update Rates
Integral Nonlinearity
See T ables III & IV
18
ppm of FSR max
Offset Error
See Note 3
0.5
2.5
See Note 3
0.6
3
See Note 3
2
Offset Error and Offset Drift Refer to Both
Unipolar Offset and Bipolar Zero Errors
Offset Drift vs. T emperature6
Offset Drift vs. T ime4
µV/°C typ
µV/1000 Hours typ
Positive Full-Scale Error5
Positive Full-Scale Drift vs. T emp6, 7
Positive Full-Scale Drift vs. T ime4
Gain Error8
µV/°C typ
µV/1000 Hours typ
Gain Drift vs. T emperature6, 9
Gain Drift vs. T ime4
ppm/°C typ
ppm/1000 Hours typ
10
Bipolar Negative Full-Scale Error
Negative Full-Scale Drift vs. T emp
Power Supply Rejection
See Note 3
0.6
90
µV/°C typ
dB typ
Measured with Zero Differential Voltage
Common-Mode Rejection (CMR) on AIN 100
dB typ
dB typ
nA max
pA/°C typ
nA max
pA/°C typ
At DC. Measured with Zero Differential Voltage
At DC. Measured with Zero Differential Voltage
CMR on REF IN
120
60
Analog Input DC Bias Current
Analog Input DC Bias Current Drift
Analog Input DC Offset Current
Analog Input DC Offset Current Drift
150
30
100
ANALOG INPUTS/REFERENCE INPUTS
Normal-Mode 50 Hz Rejection2
Normal-Mode 60 Hz Rejection2
Common-Mode 50 Hz Rejection2
Common-Mode 60 Hz Rejection2
Analog Inputs
88
88
120
120
dB min
dB min
dB min
dB min
From 49 Hz to 51 Hz
From 59 Hz to 61 Hz
From 49 Hz to 51 Hz
From 59 Hz to 61 Hz
Differential Input Voltage Ranges11
Assuming 2.5 V or 5 V Reference with
HIREF Bit Set Appropriately
Gain = 250
Gain = 125
Gain = 62.5
0 to +10 or ±10
0 to +20 or ±20
0 to +40 or ±40
0 to +80 or ±80
AGND + 1.2 V
AVDD – 0.95 V
mV nom
mV nom
mV nom
mV nom
V min
Gain = 31.25
Absolute/Common-Mode Voltage12
V max
Reference Input
REF IN(+) – REF IN(–) Voltage
REF IN(+) – REF IN(–) Voltage
Absolute/Common-Mode Voltage13
+2.5
+5
AGND – 30 mV
AVDD + 30 mV
0.3
V nom
V nom
V min
V max
V min
V max
HIREF Bit of Mode Register = 0
HIREF Bit of Mode Register = 1
NO REF T rigger Voltage
NO REF Bit Active If VREF Below T his Voltage
NO REF Bit Inactive If VREF Above T his Voltage
0.65
REV. A
–2–
AD7730/AD7730L
P aram eter
B Version1
Units
Conditions/Com m ents
LOGIC INPUT S
Input Current
±10
µA max
All Inputs Except SCLK and MCLK IN
VINL, Input Low Voltage
VINL, Input Low Voltage
VINH, Input High Voltage
SCLK Only (Schmitt T riggered Input)
VT +
VT +
VT –
VT –
VT + – VT –
0.8
0.4
2.0
V max
V max
V min
DVDD = +5 V
DVDD = +3 V
1.4/3
1/2.5
0.8/1.4
0.4/1.1
0.4/0.8
0.4/0.8
V min to V max
V min to V max
V min to V max
V min to V max
V min to V max
V min to V max
DVDD = +5 V
DVDD = +3 V
DVDD = +5 V
DVDD = +3 V
DVDD = +5 V
DVDD = +3 V
VT + – VT –
MCLK IN Only
VINL, Input Low Voltage
VINL, Input Low Voltage
VINH, Input High Voltage
VINH, Input High Voltage
0.8
0.4
3.5
2.5
V max
V max
V min
V min
DVDD = +5 V
DVDD = +3 V
DVDD = +5 V
DVDD = +3 V
LOGIC OUT PUT S (Including MCLK OUT )
VOL, Output Low Voltage
ISINK = 800 µA Except for MCLK OUT 14
;
;
15
0.4
0.4
4.0
V max
V max
V min
VDD = +5 V
VOL, Output Low Voltage
VOH , Output High Voltage
VOH , Output High Voltage
ISINK = 100 µA Except for MCLK OUT 14
15
VDD = +3 V
ISOURCE = 200 µA Except for MCLK OUT 14
;
;
15
VDD = +5 V
ISOURCE = 100 µA Except for MCLK OUT 14
15
VDD – 0.6 V
V min
VDD = +3 V
Floating State Leakage Current
±10
6
µA max
pF typ
Floating State Output Capacitance2
T RANSDUCER BURNOUT
AIN1(+) Current
AIN1(–) Current
–100
100
±10
0.1
nA nom
nA nom
% typ
Initial T olerance @ 25°C
Drift2
%/°C typ
OFFSET (T ARE) DAC
Resolution
6
Bit
LSB Size
2.3/2.6
2.5
25
mV min/mV max
ppm/°C max
ppm/1000 Hours typ
LSB max
2.5 mV Nominal with 5 V Reference (REF IN/2000)
Guaranteed Monotonic
DAC Drift16
DAC Drift vs. T ime4, 16
Differential Linearity
–0.25/+0.75
SYST EM CALIBRAT ION
Positive Full-Scale Calibration Limit17
1.05 × FS
V max
FS Is the Nominal Full-Scale Voltage
(10 mV, 20 mV, 40 mV or 80 mV)
Negative Full-Scale Calibration Limit17
Offset Calibration Limit18
Input Span17
–1.05 × FS
–1.05 × FS
0.8 × FS
V max
V max
V min
V max
2.1 × FS
POWER REQUIREMENT S
Power Supply Voltages
AVDD – AGND Voltage
DVDD Voltage
+4.75 to +5.25 V min to V max
+2.7 to +5.25
V min to V max
With AGND = 0 V
Power Supply Currents
External MCLK. Digital I/Ps = 0 V or DVDD
All Input Ranges Except 0 mV to +10 mV and ±10 mV
Input Ranges of 0 mV to +10 mV and ±10 mV Only
DVDD of 2.7 V to 3.3 V
AVDD Current (Normal Mode)
AVDD Current (Normal Mode)
DVDD Current (Normal Mode)
DVDD Current (Normal Mode)
AVDD + DVDD Current (Standby Mode)
Power Dissipation
10.3
22.3
1.3
2.7
25
mA max
mA max
mA max
mA max
µA max
DVDD of 4.75 V to 5.25 V
T ypically 10 µA. External MCLK IN = 0 V or DVDD
AVDD = DVDD = +5 V. Digital I/Ps = 0 V or DVDD
All Input Ranges Except 0 mV to +10 mV and ±10 mV
Input Ranges of 0 mV to +10 mV and ±10 mV Only
T ypically 50 µW. External MCLK IN = 0 V or DVDD
Normal Mode
65
125
125
mW max
mW max
µW max
Standby Mode
REV. A
–3–
AD7730/AD7730L
NOT ES
11T emperature range: –40°C to +85°C.
12Sample tested during initial release.
13The offset (or zero) numbers with CHP = 1 are typically 3 µV precalibration. Internal zero-scale calibration reduces this by about 1 µV. Offset numbers with CHP = 0 can be up to
1 mV precalibration. Internal zero-scale calibration reduces this to 2 µV typical. System zero-scale calibration reduces offset numbers with CHP = 1 and CHP = 0 to the order of the
noise. Gain errors can be up to 3000 ppm precalibration with CHP = 0 and CHP = 1. Performing internal full-scale calibrations on the 80 mV range reduces the gain error to less than
100 ppm for the 80 mV and 40 mV ranges, to about 250 ppm for the 20 mV range and to about 500 ppm on the 10 mV range. System full-scale calibration reduces this to the order of
the noise. Positive and negative full-scale errors can be calculated from the offset and gain errors.
14T hese numbers are generated during life testing of the part.
15Positive Full-Scale Error includes Offset Errors (Unipolar Offset Error or Bipolar Zero Error) and applies to both unipolar and bipolar input ranges. See T erminology.
16Recalibration at any temperature will remove these errors.
17Full-Scale Drift includes Offset Drift (Unipolar Offset Drift or Bipolar Zero Drift) and applies to both unipolar and bipolar input ranges.
18Gain Error is a measure of the difference between the measured and the ideal span between any two points in the transfer function. T he two points used to calculate the gain
error are positive full scale and negative full scale. See T erminology.
19Gain Error Drift is a span drift and is effectively the drift of the part if zero-scale calibrations only were performed.
10No Missing Codes performance with CHP = 0 and SKIP = 1 is reduced below 24 bits for SF words lower than 180 decimal.
11T he analog input voltage range on the AIN1(+) and AIN2(+) inputs is given here with respect to the voltage on the AIN1(–) and AIN2(–) inputs respectively.
12T he common-mode voltage range on the input pairs applies provided the absolute input voltage specification is obeyed.
13T he common-mode voltage range on the reference input pair (REF IN(+) and REF IN(–)) applies provided the absolute input voltage specification is obeyed.
14T hese logic output levels apply to the MCLK OUT output only when it is loaded with a single CMOS load.
15
V
DD
refers to DVDD for all logic outputs expect D0, D1, ACX and ACX where it refers to AVDD. In other words, the output logic high for these four outputs is determined by AVDD.
16T his number represents the total drift of the channel with a zero input and the DAC output near full scale.
17After calibration, if the input voltage exceeds positive full scale, the converter will output all 1s. If the input is less than negative full scale, the device outputs all 0s.
18T hese calibration and span limits apply provided the absolute input voltage specification is obeyed. T he offset calibration limit applies to both the unipolar zero point and the
bipolar zero point.
Specifications subject to change without notice.
(AV = +4.75 V to +5.25 V; DV = +2.7 V to +5.25 V; AGND = DGND = 0 V; fCLK IN = 4.9152 MHz;
DD
DD
1, 2
TIMING CHARACTERISTICS Input Logic 0 = 0 V, Logic 1 = DV unless otherwise noted).
DD
Lim it at TMIN to TMAX
P aram eter
(B Version)
Units
Conditions/Com m ents
Master Clock Range
1
5
50
MHz min
MHz max
ns min
For Specified Performance
t1
t2
SYNC Pulsewidth
RESET Pulsewidth
50
ns min
Read O per ation
t3
0
0
0
ns min
ns min
ns min
ns max
ns max
ns min
ns max
ns max
ns min
ns min
ns min
ns min
ns max
ns max
RDY to CS Setup T ime
t44
CS Falling Edge to SCLK Active Edge Setup T ime3
SCLK Active Edge to Data Valid Delay3
DVDD = +4.75 V to +5.25 V
t5
60
80
0
60
80
100
100
0
DVDD = +2.75 V to +3.3 V
CS Falling Edge to Data Valid Delay
DVDD = +4.75 V to +5.25 V
DVDD = +2.7 V to +3.3 V
SCLK High Pulsewidth
4, 5
t5A
t6
t7
SCLK Low Pulsewidth
t86
CS Rising Edge to SCLK Inactive Edge Hold T ime3
Bus Relinquish T ime after SCLK Inactive Edge3
t9
10
80
100
t10
SCLK Active Edge to RDY High3, 7
Wr ite O per ation
t11
t12
t13
t14
t15
t16
0
ns min
ns min
ns min
ns min
ns min
ns min
CS Falling Edge to SCLK Active Edge Setup T ime3
Data Valid to SCLK Edge Setup T ime
Data Valid to SCLK Edge Hold T ime
SCLK High Pulsewidth
SCLK Low Pulsewidth
CS Rising Edge to SCLK Edge Hold T ime
30
25
100
100
0
NOT ES
1Sample tested during initial release to ensure compliance. All input signals are specified with tr = tf = 5 ns (10% to 90% of DV DD) and timed from a voltage level of 1.6 V.
2See Figures 18 and 19.
3SCLK active edge is falling edge of SCLK with POL = 1; SCLK active edge is rising edge of SCLK with POL = 0.
4T hese numbers are measured with the load circuit of Figure 1 and defined as the time required for the output to cross the VOL or VOH limits.
5T his specification only comes into play if CS goes low while SCLK is low (POL = 1) or if CS goes low while SCLK is high (POL = 0). It is primarily required for
interfacing to DSP machines.
6T hese numbers are derived from the measured time taken by the data output to change 0.5 V when loaded with the circuit of Figure 1. T he measured number is then
extrapolated back to remove effects of charging or discharging the 50 pF capacitor. T his means that the times quoted in the timing characteristics are the true bus
relinquish times of the part and as such are independent of external bus loading capacitances.
7RDY returns high after the first read from the device after an output update. T he same data can be read again, if required, while RDY is high, although care should
be taken that subsequent reads do not occur close to the next output update.
REV. A
–4–
AD7730/AD7730L
Plastic DIP Package, Power Dissipation . . . . . . . 450 mW
ABSO LUTE MAXIMUM RATINGS*
(T A = +25°C unless otherwise noted)
θ
JA T hermal Impedance . . . . . . . . . . . . . . . . . 105°C/W
Lead T emperature (Soldering, 10 sec) . . . . . . . +260°C
T SSOP Package, Power Dissipation . . . . . . . . . . 450 mW
AVDD to AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
AVDD to DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
DVDD to AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
DVDD to DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +7 V
AGND to DGND . . . . . . . . . . . . . . . . . . . . . . –5 V to +0.3 V
AVDD to DVDD . . . . . . . . . . . . . . . . . . . . . . . . . –2 V to +5 V
Analog Input Voltage to AGND . . . . –0.3 V to AVDD + 0.3 V
Reference Input Voltage to AGND . . –0.3 V to AVDD + 0.3 V
AIN/REF IN Current (Indefinite) . . . . . . . . . . . . . . . . 30 mA
Digital Input Voltage to DGND . . . . –0.3 V to DVDD + 0.3 V
Digital Output Voltage to DGND . . . –0.3 V to DVDD + 0.3 V
Output Voltage (ACX, ACX, D0, D1) to DGND
. . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to AVDD + 0.3 V
Operating T emperature Range
Industrial (B Version) . . . . . . . . . . . . . . . –40°C to +85°C
Storage T emperature Range . . . . . . . . . . . –65°C to +150°C
Junction T emperature . . . . . . . . . . . . . . . . . . . . . . . . +150°C
θ
JA T hermal Impedance . . . . . . . . . . . . . . . . . 128°C/W
Lead T emperature, Soldering
Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . +215°C
Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . +220°C
SOIC Package, Power Dissipation . . . . . . . . . . . . 450 mW
θ
JA T hermal Impedance . . . . . . . . . . . . . . . . . . 75°C/W
Lead T emperature, Soldering
Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . +215°C
Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . +220°C
*Stresses above those listed under Absolute Maximum Ratings may cause
permanent damage to the device. T his is a stress rating only; functional
operation of the device at these or any other conditions above those listed in
the operational sections of this specification is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device
reliability.
O RD ERING GUID E
Tem perature
Range
P ackage
D escription
P ackage
O ptions
Model
AD7730BN
AD7730BR
AD7730BRU
EVAL-AD7730EB
AD7730LBR
AD7730LBRU
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
Evaluation Board
–40°C to +85°C
–40°C to +85°C
Plastic DIP
Small Outline
T hin Shrink Small Outline
N-24
R-24
RU-24
Small Outline
T hin Shrink Small Outline
R-24
RU-24
EVAL-AD7730LEB Evaluation Board
I
(800A AT DV = +5V
DD
SINK
100A AT DV = +3V)
DD
TO OUTPUT
PIN
+1.6V
50pF
I
(200A AT DV = +5V
DD
SOURCE
100A AT DV = +3V)
DD
Figure 1. Load Circuit for Access Tim e and Bus Relinquish Tim e
CAUTIO N
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD7730 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. T herefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
–5–
REV. A
AD7730/AD7730L
PROGRAMMABLE
DIGITAL FILTER
DIFFERENTIAL
REFERENCE
PROGRAMMABLE GAIN
AMPLIFIER
SIGMA-DELTA ADC
BUFFER AMPLIFIER
THE SIGMA-DELTA
TWO STAGE FILTER THAT
ALLOWS PROGRAMMING OF
OUTPUT UPDATE RATE AND
SETTLING TIME AND WHICH HAS
A FAST STEP MODE
THE BUFFER AMPLIFIER
PRESENTS A HIGH IMPEDANCE
INPUT STAGE FOR THE ANALOG
INPUTS ALLOWING SIGNIFICANT
EXTERNAL SOURCE
THE REFERENCE INPUT TO THE
PART IS DIFFERENTIAL AND
FACILITATES RATIOMETRIC
OPERATION. THE REFERENCE
VOLTAGE CAN BE SELECTED TO
BE NOMINALLY +2.5V OR +5V
ARCHITECTURE ENSURES 24 BITS
NO MISSING CODES. THE
ENTIRE SIGMA-DELTA ADC CAN
BE CHOPPED TO REMOVE DRIFT
ERRORS
THE PROGRAMMABLE GAIN
AMPLIFIER ALLOWS FOUR
UNIPOLAR AND FOUR BIPOLAR
INPUT RANGES FROM
(SEE FIGURE 3)
+10mV TO +80mV
IMPEDANCES
SEE PAGE 26
SEE PAGE 26
SEE PAGE 24
SEE PAGE 24
SEE PAGE 25
BURNOUT CURRENTS
TWO 100nA BURNOUT
CURRENTS ALLOW THE USER
TO EASILY DETECT IF A
TRANSDUCER HAS BURNT
OUT OR GONE OPEN-CIRCUIT
REF IN(–)
STANDBY MODE
DV
REF IN(+)
AV
DD
DD
THE STANDBY MODE REDUCES
POWER CONSUMPTION TO 5A
AD7730
SEE PAGE 25
REFERENCE DETECT
VBIAS
SEE PAGE 33
AV
DD
STANDBY
AIN1(+)
AIN1(–)
SIGMA-DELTA A/D CONVERTER
SIGMA-
DELTA
MODULATOR
PROGRAMMABLE
DIGITAL
CLOCK OSCILLATOR
CIRCUIT
SYNC
+
FILTER
MUX
PGA
+/–
BUFFER
THE CLOCK SOURCE FOR THE
PART CAN BE PROVIDED BY AN
EXTERNALLY-APPLIED CLOCK OR
BY CONNECTING A CRYSTAL OR
CERAMIC RESONATOR ACROSS
THE CLOCK PINS
AIN2(+)/D1
AIN2(–)/D0
MCLK IN
CLOCK
GENERATION
6-BIT
DAC
SERIAL INTERFACE
AND CONTROL LOGIC
AGND
MCLK OUT
REGISTER BANK
SEE PAGE 32
SCLK
CS
CALIBRATION
MICROCONTROLLER
ANALOG MULTIPLEXER
DIN
ACX
AC
A TWO-CHANNEL DIFFERENTIAL
MULTIPLEXER SWITCHES ONE OF
THE TWO DIFFERENTIAL INPUT
CHANNELS TO THE BUFFER
AMPLIFIER. THE MULTIPLEXER IS
CONTROLLED VIA THE SERIAL
INTERFACE
DOUT
EXCITATION
CLOCK
ACX
SERIAL INTERFACE
AGND
DGND
POL
RDY
SPI*-COMPATIBLE OR DSP-
COMPATIBLE SERIAL INTERFACE
WHICH CAN BE OPERATED FROM
JUST THREE WIRES. ALL
RESET
SEE PAGE 24
FUNCTIONS ON THE PART
CAN BE ACCESSED VIA
THE SERIAL INTERFACE
SEE PAGE 35
AC EXCITATION
OUTPUT DRIVERS
OFFSET/TARE DAC
REGISTER BANK
FOR AC-EXCITED BRIDGE
APPLICATIONS, THE ACX
OUTPUTS PROVIDE SIGNALS
THAT CAN BE USED TO SWITCH
THE POLARITY OF THE BRIDGE
EXCITATION VOLTAGE
THE SECOND ANALOG INPUT
CHANNEL CAN BE
RECONFIGURED TO BECOME TWO
OUTPUT DIGITAL PORT LINES
WHICH CAN BE PROGRAMMED
OVER THE SERIAL INTERFACE
ALLOWS A PROGRAMMED
VOLTAGE TO BE EITHER ADDED
OR SUBTRACTED FROM THE
ANALOG INPUT SIGNAL BEFORE
IT IS APPLIED TO THE PGA
THIRTEEN REGISTERS CONTROL
ALL FUNCTIONS ON THE PART AND
PROVIDE STATUS INFORMATION
AND CONVERSION RESULTS
SEE PAGE 11
SEE PAGE 24
SEE PAGE 41
SEE PAGE 33
*SPI IS A TRADEMARK OF MOTOROLA, INC.
Figure 2. Detailed Functional Block Diagram
REV. A
–6–
AD7730/AD7730L
INPUT CHOPPING
3
22-TAP FIR FILTER
SINC FILTER
SKIP MODE
THE ANALOG INPUT TO THE PART CAN BE
CHOPPED. IN CHOPPING MODE, WITH
AC EXCITATION DISABLED, THE INPUT
CHOPPING IS INTERNALTO THE DEVICE. IN
CHOPPING MODE, WITH AC EXCITATION
ENABLED, THE CHOPPING IS ASSUMED
TO BE PERFORMED EXTERNAL TO THE PART
AND NO INTERNAL INPUT CHOPPING IS
PERFORMED. THE INPUT CHOPPING CAN
BE DISABLED, IF DESIRED.
IN NORMAL OPERATING MODE, THE
SECOND STAGE OF THE DIGITAL FILTERING
ON THE PART IS A FIXED 22-TAP FIR
FILTER. IN SKIP MODE, THIS FIR FILTER IS
BYPASSED. WHEN FASTSTEP™MODE IS
ENABLED AND A STEP INPUT IS
THE FIRST STAGE OF THE DIGITAL FILTERING
ON THE PART IS THE SINC FILTER. THE
IN SKIP MODE, THERE IS NO SECOND
3
STAGE OF FILTERING ON THE PART. THE
3
OUTPUT UPDATE RATE AND BANDWIDTH
OF THIS FILTER CAN BE PROGRAMMED. IN
SINC FILTER IS THE ONLY FILTERING
PERFORMED ON THE PART.
3
SKIP MODE, THE SINC FILTER IS THE
ONLY FILTERING PERFORMED ON THE PART.
SEE PAGE 29
DETECTED, THE SECOND STAGE FILTERING
IS PERFORMED BY THE FILTER
SEE PAGE 26
UNTIL THE OUTPUT OF THIS FILTER
HAS FULLY SETTLED.
SEE PAGE 26
SEE PAGE 27
SKIP
PGA +
SIGMA-DELTA
MODULATOR
22-TAP
CHOP
OUTPUT
SCALING
ANALOG
INPUT
DIGITAL
OUTPUT
3
SINC FILTER
CHOP
BUFFER
FIR FILTER
FASTSTEP
FILTER
OUTPUT SCALING
THE OUTPUT WORD FROM THE DIGITAL
FILTER IS SCALED BY THE CALIBRATION
COEFFICIENTS BEFORE BEING PROVIDED
AS THE CONVERSION RESULT.
OUTPUT CHOPPING
BUFFER
PGA + SIGMA-DELTA MODULATOR
THE OUTPUT OF THE FIRST STAGE
OF FILTERING ON THE PART CAN
BE CHOPPED. IN CHOPPING MODE,
REGARDLESS OF WHETHER AC
EXCITATION IS ENABLED OR DISABLED,
THE OUTPUT CHOPPING IS
THE INPUT SIGNAL IS BUFFERED
ON-CHIP BEFORE BEING APPLIED TO
THE SAMPLING CAPACITOR OF THE
SIGMA-DELTA MODULATOR. THIS
ISOLATES THE SAMPLING CAPACITOR
CHARGING CURRENTS FROM THE
ANALOG INPUT PINS.
THE PROGRAMMABLE GAIN CAPABILITY
OF THE PART IS INCORPORATED
AROUND THE SIGMA-DELTA MODULATOR.
THE MODULATOR PROVIDES A HIGH-
FREQUENCY 1-BIT DATA STREAM
TO THE DIGITAL FILTER.
SEE PAGE 29
FASTSTEP FILTER
PERFORMED. THE CHOPPING CAN
BE DISABLED, IF DESIRED.
SEE PAGE 26
WHEN FASTSTEP MODE IS ENABLED
AND A STEP CHANGE ON THE INPUT
HAS BEEN DETECTED, THE SECOND
STAGE FILTERING IS PERFORMED BY THE
FASTSTEP FILTER UNTIL THE FIR
FILTER HAS FULLY SETTLED.
SEE PAGE 24
SEE PAGE 26
SEE PAGE 29
Figure 3. Signal Processing Chain
P IN CO NFIGURATIO N
SCLK
1
2
24 DGND
DV
23
MCLK IN
DD
22 DIN
MCLK OUT
POL
3
4
5
6
7
8
9
21 DOUT
SYNC
20
19
18
17
16
15
14
RDY
CS
AD7730
TOP VIEW
(Not to Scale)
RESET
V
STANDBY
ACX
BIAS
AGND
AV
DD
ACX
AIN1(+) 10
REF IN(–)
AIN1(–)
11
12
REF IN(+)
13 AIN2(–)/D0
AIN2(+)/D1
P IN FUNCTIO N D ESCRIP TIO NS
P in
No. Mnem onic
Function
1
SCLK
Serial Clock. Schmitt-T riggered Logic Input. An external serial clock is applied to this input to transfer serial
data to or from the AD7730. T his serial clock can be a continuous clock with all data transmitted in a con-
tinuous train of pulses. Alternatively, it can be a noncontinuous clock with the information being transmitted
to or from the AD7730 in smaller batches of data.
2
MCLK IN
Master Clock signal for the device. T his can be provided in the form of a crystal/resonator or external clock. A
crystal/resonator can be tied across the MCLK IN and MCLK OUT pins. Alternatively, the MCLK IN pin
can be driven with a CMOS-compatible clock and MCLK OUT left unconnected. T he AD7730 is specified
with a clock input frequency of 4.9152 MHz while the AD7730L is specified with a clock input frequency of
2.4576 MHz.
–7–
REV. A
AD7730/AD7730L
P in
No. Mnem onic
Function
3
MCLK OUT When the master clock for the device is a crystal/resonator, the crystal/resonator is connected between MCLK IN
and MCLK OUT. If an external clock is applied to the MCLK IN, MCLK OUT provides an inverted clock sig-
nal. This clock can be used to provide a clock source for external circuits and MCLK OUT is capable of driving
one CMOS load. If the user does not require it, MCLK OUT can be turned off with the CLKDIS bit of the Mode
Register. This ensures that the part is not burning unnecessary power driving capacitance on the MCLK OUT pin.
4
POL
Clock Polarity. Logic Input. T his determines the polarity of the serial clock. If the active edge for the proces-
sor is a high-to-low SCLK transition, this input should be low. In this mode, the AD7730 puts out data on the
DAT A OUT line in a read operation on a low-to-high transition of SCLK and clocks in data from the DAT A
IN line in a write operation on a high-to-low transition of SCLK. In applications with a noncontinuous serial
clock (such as most microcontroller applications), this means that the serial clock should idle low between
data transfers. If the active edge for the processor is a low-to-high SCLK transition, this input should be high.
In this mode, the AD7730 puts out data on the DAT A OUT line in a read operation on a high-to-low transi-
tion of SCLK and clocks in data from the DAT A IN line in a write operation on a low-to-high transition of
SCLK. In applications with a noncontinuous serial clock (such as most microcontroller applications), this
means that the serial clock should idle high between data transfers.
5
SYNC
Logic Input that allows for synchronization of the digital filters and analog modulators when using a number
of AD7730s. While SYNC is low, the nodes of the digital filter, the filter control logic and the calibration
control logic are reset and the analog modulator is also held in its reset state. SYNC does not affect the digital
interface but does reset RDY to a high state if it is low. While SYNC is asserted, the Mode Bits may be set up
for a subsequent operation which will commence when the SYNC pin is deasserted.
6
7
8
RESET
VBIAS
Logic Input. Active low input that resets the control logic, interface logic, digital filter, analog modulator and
all on-chip registers of the part to power-on status. Effectively, everything on the part except for the clock
oscillator is reset when the RESET pin is exercised.
Analog Output. T his analog output is an internally-generated voltage used as an internal operating bias point.
T his output is not for use external to the AD7730 and it is recommended that the user does not connect any-
thing to this pin.
AGND
Ground reference point for analog circuitry.
9
AVDD
Analog Positive Supply Voltage. T he AVDD to AGND differential is 5 V nominal.
10
AIN1(+)
Analog Input Channel 1. Positive input of the differential, programmable-gain primary analog input pair. T he
differential analog input ranges are 0 mV to +10 mV, 0 mV to +20 mV, 0 mV to +40 mV and 0 mV to +80 mV
in unipolar mode, and ±10 mV, ±20 mV, ±40 mV and ±80 mV in bipolar mode.
11
12
AIN1(–)
Analog Input Channel 1. Negative input of the differential, programmable gain primary analog input pair.
AIN2(+)/D1 Analog Input Channel 2 or Digital Output 1. T his pin can be used either as part of a second analog input
channel or as a digital output bit as determined by the DEN bit of the Mode Register. When selected as an
analog input, it is the positive input of the differential, programmable-gain secondary analog input pair. T he
analog input ranges are 0 mV to +10 mV, 0 mV to +20 mV, 0 mV to +40 mV and 0 mV to +80 mV in unipo-
lar mode and ±10 mV, ±20 mV, ±40 mV and ±80 mV in bipolar mode. When selected as a digital output,
this output can programmed over the serial interface using bit D1 of the Mode Register.
13
14
AIN2(–)/D0
Analog Input Channel 2 or Digital Output 0. This pin can be used either as part of a second analog input channel
or as a digital output bit as determined by the DEN bit of the Mode Register. When selected as an analog input, it
is the negative input of the differential, programmable-gain secondary analog input pair. When selected as a digital
output, this output can programmed over the serial interface using bit D0 of the Mode Register.
REF IN(+)
Reference Input. Positive terminal of the differential reference input to the AD7730. REF IN(+) can lie
anywhere between AVDD and AGND. T he nominal reference voltage (the differential voltage between REF
IN(+) and REF IN(–)) should be +5 V when the HIREF bit of the Mode Register is 1 and +2.5 V when the
HIREF bit of the Mode Register is 0.
15
16
REF IN(–)
ACX
Reference Input. Negative terminal of the differential reference input to the AD7730. T he REF IN(–) poten-
tial can lie anywhere between AVDD and AGND.
Digital Output. Provides a signal that can be used to control the reversing of the bridge excitation in ac-
excited bridge applications. When ACX is high, the bridge excitation is taken as normal and when ACX is
low, the bridge excitation is reversed (chopped). If AC = 0 (ac mode turned off) or CHP = 0 (chop mode
turned off), the ACX output remains high.
17
ACX
Digital Output. Provides a signal that can be used to control the reversing of the bridge excitation in ac-
excited bridge applications. T his output is the complement of ACX. In ac mode, this means that it toggles in
anti-phase with ACX . If AC = 0 (ac mode turned off) or CHP = 0 (chop mode turned off), the ACX output
remains low. When toggling, it is guaranteed to be nonoverlapping with ACX. T he non-overlap interval, when
both ACX and ACX are low, is one master clock cycle.
REV. A
–8–
AD7730/AD7730L
P in
No. Mnem onic
Function
18
STANDBY
Logic Input. T aking this pin low shuts down the analog and digital circuitry, reducing current consumption to
the 5 µA range. T he on-chip registers retain all their values when the part is in standby mode.
19
CS
Chip Select. Active low Logic Input used to select the AD7730. With this input hardwired low, the AD7730
can operate in its three-wire interface mode with SCLK, DIN and DOUT used to interface to the device. CS
can be used to select the device in systems with more than one device on the serial bus or as a frame synchro-
nization signal in communicating with the AD7730.
20
RDY
Logic Output. Used as a status output in both conversion mode and calibration mode. In conversion mode, a
logic low on this output indicates that a new output word is available from the AD7730 data register. T he
RDY pin will return high upon completion of a read operation of a full output word. If no data read has taken
place after an output update, the RDY line will return high prior to the next output update, remain high while
the update is taking place and return low again. T his gives an indication of when a read operation should not
be initiated to avoid initiating a read from the data register as it is being updated. In calibration mode, RDY
goes high when calibration is initiated and it returns low to indicate that calibration is complete. A number of
different events on the AD7730 set the RDY high and these are outlined in T able XVIII.
21
22
DOUT
DIN
Serial Data Output with serial data being read from the output shift register on the part. T his output shift
register can contain information from the calibration registers, mode register, status register, filter register,
DAC register or data register, depending on the register selection bits of the Communications Register.
Serial Data Input with serial data being written to the input shift register on the part. Data from this input
shift register is transferred to the calibration registers, mode register, communications register, DAC register
or filter registers depending on the register selection bits of the Communications Register.
23
24
DVDD
Digital Supply Voltage, +3 V or +5 V nominal.
Ground reference point for digital circuitry.
DGND
TERMINO LO GY
BIP O LAR NEGATIVE FULL-SCALE ERRO R
INTEGRAL NO NLINEARITY
T his is the deviation of the first code transition from the ideal
AIN(+) voltage (AIN(–) – VREF/GAIN + 0.5 LSB) when operat-
ing in the bipolar mode. Negative full-scale error is a summation
of zero error and gain error.
T his is the maximum deviation of any code from a straight line
passing through the endpoints of the transfer function. T he end-
points of the transfer function are zero scale (not to be confused
with bipolar zero), a point 0.5 LSB below the first code transi-
tion (000 . . . 000 to 000 . . . 001) and full scale, a point 0.5 LSB
above the last code transition (111 . . . 110 to 111 . . . 111). T he
error is expressed as a percentage of full scale.
P O SITIVE FULL-SCALE O VERRANGE
Positive Full-Scale Overrange is the amount of overhead avail-
able to handle input voltages on AIN(+) input greater than
AIN(–) + VREF/GAIN (for example, noise peaks or excess volt-
ages due to system gain errors in system calibration routines) with-
out introducing errors due to overloading the analog modulator
or overflowing the digital filter.
P O SITIVE FULL-SCALE ERRO R
Positive Full-Scale Error is the deviation of the last code transition
(111 . . . 110 to 111 . . . 111) from the ideal AIN(+) voltage
(AIN(–) + VREF/GAIN – 3/2 LSBs). It applies to both unipolar
and bipolar analog input ranges. Positive full-scale error is a
summation of offset error and gain error.
NEGATIVE FULL-SCALE O VERRANGE
T his is the amount of overhead available to handle voltages on
AIN(+) below AIN(–) – VREF/GAIN without overloading the
analog modulator or overflowing the digital filter.
UNIP O LAR O FFSET ERRO R
Unipolar Offset Error is the deviation of the first code transition
from the ideal AIN(+) voltage (AIN(–) + 0.5 LSB) when oper-
ating in the unipolar mode.
O FFSET CALIBRATIO N RANGE
In the system calibration modes, the AD7730 calibrates its
offset with respect to the analog input. T he Offset Calibration
Range specification defines the range of voltages the AD7730
can accept and still accurately calibrate offset.
BIP O LAR ZERO ERRO R
T his is the deviation of the midscale transition (0111 . . . 111 to
1000 . . . 000) from the ideal AIN(+) voltage (AIN(–) – 0.5 LSB)
when operating in the bipolar mode.
FULL-SCALE CALIBRATIO N RANGE
T his is the range of voltages that the AD7730 can accept in the
system calibration mode and still calibrate full scale correctly.
GAIN ERRO R
T his is a measure of the span error of the ADC. It is a measure
of the difference between the measured and the ideal span be-
tween any two points in the transfer function. T he two points
used to calculate the gain error are full scale and zero scale.
INP UT SP AN
In system calibration schemes, two voltages applied in sequence
to the AD7730’s analog input define the analog input range.
T he input span specification defines the minimum and maxi-
mum input voltages, from zero to full scale, the AD7730 can
accept and still accurately calibrate gain.
–9–
REV. A
AD7730/AD7730L
O UTP UT NO ISE AND RESO LUTIO N SP ECIFICATIO N
T he AD7730 can be programmed to operate in either chop mode or nonchop mode. T he chop mode can be enabled in ac-excited or
dc-excited applications; it is optional in dc-excited applications, but chop mode must be enabled in ac-excited applications. T hese
options are discussed in more detail in later sections. T he chop mode has the advantage of lower drift numbers and better noise im-
munity, but the noise is approximately 20% higher for a given –3 dB frequency and output data rate. It is envisaged that the majority
of weigh-scale users of the AD7730 will operate the part in chop mode to avail themselves of the excellent drift performance and
noise immunity when chopping is enabled. T he following tables outline the noise performance of the part in both chop and nonchop
modes over all input ranges for a selection of output rates. Settling time refers to the time taken to get an output that is 100% settled
to new value.
O utput Noise (CH P = 1)
T his mode is the primary mode of operation of the device. T able I shows the output rms noise for some typical output update rates
and –3 dB frequencies for the AD7730 when used in chopping mode (CHP of Filter Register = 1) with a master clock frequency of
4.9152 MHz. T hese numbers are typical and are generated at a differential analog input voltage of 0 V. T he output update rate is
selected via the SF0 to SF11 bits of the Filter Register. T able II, meanwhile, shows the output peak-to-peak resolution in counts for
the same output update rates. T he numbers in brackets are the effective peak-to-peak resolution in bits (rounded to the nearest 0.5
LSB). It is important to note that the numbers in T able II represent the resolution for which there will be no code flicker within a
six-sigma limit. T hey are not calculated based on rms noise, but on peak-to-peak noise.
T he numbers are generated for the bipolar input ranges. When the part is operated in unipolar mode, the output noise will be the
same as the equivalent bipolar input range. As a result, the numbers in T able I will remain the same for unipolar ranges while the
numbers in T able II will change. T o calculate the numbers for T able II for unipolar input ranges simply divide the peak-to-peak
resolution number in counts by two or subtract one from the peak-to-peak resolution number in bits.
Table I. O utput Noise vs. Input Range and Update Rate (CH P = 1)
Typical O utput RMS Noise in nV
O utput
D ata Rate Frequency
–3 dB
SF
Word
Settling Tim e
Norm al Mode
Settling Tim e
Fast Mode
Input Range
= ؎80 m V
Input Range Input Range Input Range
= ؎40 m V
= ؎20 m V
= ؎10 m V
50 Hz
1.97 Hz
3.95 Hz
5.92 Hz
7.9 Hz
2048
1024
683
512
256
460 ms
230 ms
153 ms
115 ms
57.5 ms
60 ms
30 ms
20 ms
15 ms
7.5 ms
115
155
200
225
335
75
55
75
95
100
40
60
70
80
110
100 Hz
150 Hz
200 Hz*
400 Hz
105
135
145
225
15.8 Hz
160
*Power-On Default
Table II. P eak-to-P eak Resolution vs. Input Range and Update Rate (CH P = 1)
P eak-to-P eak Resolution in Counts (Bits)
O utput
D ata Rate Frequency
–3 dB
SF
Word
Settling Tim e
Norm al Mode
Settling Tim e
Fast Mode
Input Range
= ؎80 m V
Input Range Input Range Input Range
= ؎40 m V
= ؎20 m V
= ؎10 m V
50 Hz
1.97 Hz
3.95 Hz
5.92 Hz
7.9 Hz
2048
1024
683
512
256
460 ms
230 ms
153 ms
115 ms
57.5 ms
60 ms
30 ms
20 ms
15 ms
7.5 ms
230k (18)
170k (17.5)
130k (17)
120k (17)
80k (16.5)
175k (17.5)
125k (17)
100k (16.5)
90k (16.5)
55k (16)
120k (17)
90k (16.5)
70k (16)
65k (16)
40k (15.5)
80k (16.5)
55k (16)
45k (15.5)
40k (15.5)
30k (15)
100 Hz
150 Hz
200 Hz*
400 Hz
15.8 Hz
*Power-On Default
O utput Noise (CH P = 0)
T able III shows the output rms noise for some typical output update rates and –3 dB frequencies for the AD7730 when used in non-
chopping mode (CHP of Filter Register = 0) with a master clock frequency of 4.9152 MHz. T hese numbers are typical and are gen-
erated at a differential analog input voltage of 0 V. T he output update rate is selected via the SF0 to SF11 bits of the Filter Register.
T able IV, meanwhile, shows the output peak-to-peak resolution in counts for the same output update rates. T he numbers in brackets
are the effective peak-to-peak resolution in bits (rounded to the nearest 0.5 LSB). It is important to note that the numbers in T able
IV represent the resolution for which there will be no code flicker within a six-sigma limit. T hey are not calculated based on rms
noise, but on peak-to-peak noise.
T he numbers are generated for the bipolar input ranges. When the part is operated in unipolar mode, the output noise will be the
same as the equivalent bipolar input range. As a result, the numbers in T able III will remain the same for unipolar ranges while the
numbers in T able IV will change. T o calculate the number for T able IV for unipolar input ranges simply divide the peak-to-peak
resolution number in counts by two or subtract one from the peak-to-peak resolution number in bits.
REV. A
–10–
AD7730/AD7730L
Table III. O utput Noise vs. Input Range and Update Rate (CH P = 0)
Typical O utput RMS Noise in nV
O utput
D ata Rate Frequency
–3 dB
SF
Word
Settling Tim e
Norm al Mode
Settling Tim e
Fast Mode
Input Range
= ؎80 m V
Input Range Input Range Input Range
= ؎40 m V
= ؎20 m V
= ؎10 m V
150 Hz
200 Hz
300 Hz
600 Hz
1200 Hz
5.85 Hz
7.8 Hz
11.7 Hz
23.4 Hz
46.8 Hz
2048
1536
1024
512
166 ms
125 ms
83.3 ms
41.6 ms
20.8 ms
26.6 ms
20 ms
13.3 ms
6.6 ms
3.3 ms
160
190
235
300
435
110
130
145
225
315
80
95
100
135
210
60
75
80
110
150
256
Table IV. P eak-to-P eak Resolution vs. Input Range and Update Rate (CH P = 0)
P eak-to-P eak Resolution in Counts (Bits)
O utput
D ata Rate Frequency
–3 dB
SF
Word
Settling Tim e
Norm al Mode
Settling Tim e
Fast Mode
Input Range
= ؎80 m V
Input Range Input Range Input Range
= ؎40 m V
= ؎20 m V
= ؎10 m V
150 Hz
200 Hz
300 Hz
600 Hz
1200 Hz
5.85 Hz
7.8 Hz
11.7 Hz
23.4 Hz
46.8 Hz
2048
1536
1024
512
166 ms
125 ms
83.3 ms
41.6 ms
20.8 ms
26.6 ms
20 ms
13.3 ms
6.6 ms
3.3 ms
165k (17.5)
140k (17)
115k (17)
90k (16.5)
60k (16)
120k (17)
100k (16.5)
90k (16.5)
60k (16)
80k (16.5)
70k (16)
65k (16)
50k (15.5)
32k (15)
55k (16)
45k (15.5)
40k (15.5)
30k (15)
256
43k (15.5)
20k (14.5)
O N-CH IP REGISTERS
T he AD7730 contains thirteen on-chip registers which can be accessed via the serial port of the part. T hese registers are summarized
in Figure 4 and in T able V and described in detail in the following sections.
COMMUNICATIONS REGISTER
DIN
DIN
RS2 RS1 RS0
DOUT
DOUT
DOUT
STATUS REGISTER
DATA REGISTER
MODE REGISTER
DIN
DOUT
DOUT
REGISTER
SELECT
DECODER
DIN
DIN
FILTER REGISTER
DAC REGISTER
DOUT
DIN
DIN
DIN
DOUT
DOUT
OFFSET REGISTER (x3)
GAIN REGISTER (x3)
TEST REGISTER
DOUT
Figure 4. Register Overview
–11–
REV. A
AD7730/AD7730L
Table V. Sum m ary of O n-Chip Registers
P ower-O n/Reset
Register Nam e
Type
Size
D efault Value
Function
Communications
Register
Write Only 8 Bits
Not Applicable
All operations to other registers are initiated through
the Communications Register. T his controls whether
subsequent operations are read or write operations
and also selects the register for that subsequent
operation. Most subsequent operations return con-
trol to the Communications Register except for the
continuous read mode of operation.
WEN ZERO RW1
RW0 ZERO
RS2
RS1
RS0
Status Register
Read Only
8 Bits
CX Hex
MS1 MS0
Provides status information on conversions, calibra-
tions, settling to step inputs, standby operation and
the validity of the reference voltage.
RDY
STDY STBY NOREF
MS3
MS2
Data Register
Read Only
16 Bits or 24 Bits 000000 Hex
Provides the most up-to-date conversion result from
the part. Register length can be programmed to be
16 bits or 24 bits.
Mode Register
Read/Write 16 Bits
01B0 Hex
Controls functions such as mode of operation, uni-
polar/bipolar operation, controlling the function of
AIN2(+)/D1 and AIN2(-)/D0, burnout current,
Data Register word length and disabling of MCLK
OUT . It also contains the reference selection bit, the
range selection bits and the channel selection bits.
MD2
MD1
MD0
RN1
B/U
DEN
D1
BO
D0
CH1
WL
HIREF ZERO
RN0 CLKDIS
CH0
Filter Register
Read/Write 24 Bits
200010 Hex
Controls the amount of averaging in the first stage
filter, selects the fast step and skip modes and con-
trols the ac excitation and chopping modes on the
part.
SF11 SF10 SF9
SF8
SF0
CHP
SF7
SF6
SF5
SF4
SF3 SF2
SF1
ZERO ZERO SKIP FAST
DL3 DL2 DL1 DL0
ZERO ZERO AC
DAC Register
Read/Write 8 Bits
DAC5 DAC4 DAC3 DAC2
20 Hex
DAC1 DAC0
Provides control of the amount of correction per-
formed by the Offset/T ARE DAC.
ZERO
ZERO
Offset Register
Gain Register
T est Register
Read/Write 24 Bits
Read/Write 24 Bits
Read/Write 24 Bits
800000 Hex
59AEE7 Hex
000000 Hex
Contains a 24-bit word which is the offset calibration
coefficient for the part. T he contents of this register
are used to provide offset correction on the output
from the digital filter. T here are three Offset Regis-
ters on the part and these are associated with the
input channels as outlined in T able XIII.
Contains a 24-bit word which is the gain calibration
coefficient for the part. T he contents of this register
are used to provide gain correction on the output
from the digital filter. T here are three Gain Registers
on the part and these are associated with the input
channels as outlined in T able XIII.
Controls the test modes of the part which are used
when testing the part. The user is advised not to
change the contents of this r egister .
REV. A
–12–
AD7730/AD7730L
Com m unications Register (RS2–RS0 = 0, 0, 0)
T he Communications Register is an 8-bit write-only register. All communications to the part must start with a write operation to the
Communications Register. T he data written to the Communications Register determines whether the next operation is a read or
write operation, the type of read operation, and to which register this operation takes place. For single-shot read or write operations,
once the subsequent read or write operation to the selected register is complete, the interface returns to where it expects a write op-
eration to the Communications Register. T his is the default state of the interface, and on power-up or after a RESET, the AD7730
is in this default state waiting for a write operation to the Communications Register. In situations where the interface sequence is
lost, a write operation of at least 32 serial clock cycles with DIN high, returns the AD7730 to this default state by resetting the
part. T able VI outlines the bit designations for the Communications Register. CR0 through CR7 indicate the bit location, CR denot-
ing the bits are in the Communications Register. CR7 denotes the first bit of the data stream.
Table VI. Com m unications Register
CR7
CR6
CR5
CR4
CR3
CR2
RS2
CR1
RS1
CR0
RS0
WEN
ZERO
RW1
RW0
ZERO
Bit
Bit
Location
Mnem onic
D escription
CR7
WEN
Write Enable Bit. A 0 must be written to this bit so the write operation to the Communications
Register actually takes place. If a 1 is written to this bit, the part will not clock on to subsequent
bits in the register. It will stay at this bit location until a 0 is written to this bit. Once a 0 is writ-
ten to the WEN bit, the next seven bits will be loaded to the Communications Register.
CR6
ZERO
A zero m ust be written to this bit to ensure correct operation of the AD7730.
CR5, CR4
RW1, RW0
Read/Write Mode Bits. T hese two bits determine the nature of the subsequent read/write opera-
tion. T able VII outlines the four options.
Table VII. Read/Write Mode
RW1
RW0
Read/Wr ite Mode
0
0
1
1
0
1
0
1
Single Write to Specified Register
Single Read of Specified Register
Start Continuous Read of Specified Register
Stop Continuous Read Mode
With 0, 0 written to these two bits, the next operation is a write operation to the register specified by
bits RS2, RS1, RS0. Once the subsequent write operation to the specified register has been com-
pleted, the part returns to where it is expecting a write operation to the Communications Register.
With 0,1 written to these two bits, the next operation is a read operation of the register specified
by bits RS2, RS1, RS0. Once the subsequent read operation to the specified register has been
completed, the part returns to where it is expecting a write operation to the Communications
Register.
Writing 1,0 to these bits, sets the part into a mode of continuous reads from the register speci-
fied by bits RS2, RS1, RS0. T he most likely registers with which the user will want to use this
function are the Data Register and the Status Register. Subsequent operations to the part will
consist of read operations to the specified register without any intermediate writes to the Com-
munications Register. T his means that once the next read operation to the specified register has
taken place, the part will be in a mode where it is expecting another read from that specified
register. T he part will remain in this continuous read mode until 30 Hex has been written to the
Communications Register.
When 1,1 is written to these bits (and 0 written to bits CR3 through CR0), the continuous read
mode is stopped and the part returns to where it is expecting a write operation to the Communi-
cations Register. Note, the part continues to look at the DIN line on each SCLK edge during
continuous read mode to determine when to stop the continuous read mode. T herefore, the user
must be careful not to inadvertently exit the continuous read mode or reset the AD7730 by
writing a series of 1s to the part. T he easiest way to avoid this is to place a logic 0 on the DIN
line while the part is in continuous read mode. Once the part is in continuous read mode, the
user should ensure that an integer multiple of 8 serial clocks should have taken place before
attempting to take the part out of continuous read mode.
–13–
REV. A
AD7730/AD7730L
Bit
Bit
Location
Mnem onic
D escription
CR3
ZERO
A zero m ust be written to this bit to ensure correct operation of the AD7730.
CR2–CR0
RS2–RS0
Register Selection Bits. RS2 is the MSB of the three selection bits. T he three bits select
which register type the next read or write operation operates upon as shown in T able VIII.
Table VIII. Register Selection
RS2
RS1
RS0
Register
0
0
0
0
0
1
1
1
1
0
0
0
1
1
0
0
1
1
0
0
1
0
1
0
1
0
1
Communications Register (Write Operation)
Status Register (Read Operation)
Data Register
Mode Register
Filter Register
DAC Register
Offset Register
Gain Register
T est Register
Status Register (RS2–RS0 = 0, 0, 0); P ower -O n/Reset Status: CX H ex
T he Status Register is an 8-bit read-only register. T o access the Status Register, the user must write to the Communications Register
selecting either a single-shot read or continuous read mode and load bits RS2, RS1, RS0 with 0, 0, 0. T able IX outlines the bit desig-
nations for the Status Register. SR0 through SR7 indicate the bit location, SR denoting the bits are in the Status Register. SR7
denotes the first bit of the data stream. Figure 5 shows a flowchart for reading from the registers on the AD7730. T he number
in brackets indicates the power-on/reset default status of that bit.
Table IX. Status Register
SR7
SR6
SR5
SR4
SR3
SR2
SR1
SR0
RDY (1)
STDY (1) ST BY (0) NOREF (0) MS3 (X)
MS2 (X)
MS1 (X)
MS0 (X)
Bit
Bit
Location
Mnem onic
D escription
SR7
SR6
RDY
Ready Bit. T his bit provides the status of the RDY flag from the part. T he status and function of
this bit is the same as the RDY output pin. A number of events set the RDY bit high as indi-
cated in T able XVIII.
STDY
Steady Bit. T his bit is updated when the filter writes a result to the Data Register. If the filter is
in FASTStep mode (see Filter Register section) and responding to a step input, the STDY bit
remains high as the initial conversion results become available. T he RDY output and bit are set
low on these initial conversions to indicate that a result is available. If the STDY is high, however,
it indicates that the result being provided is not from a fully settled second-stage FIR filter. When the
FIR filter has fully settled, the STDY bit will go low coincident with RDY. If the part is never placed
into its FASTStep mode, the STDY bit will go low at the first Data Register read and it is
not cleared by subsequent Data Register reads.
A number of events set the STDY bit high as indicated in T able XVIII. STDY is set high along
with RDY by all events in the table except a Data Register read.
SR5
ST BY
Standby Bit. T his bit indicates whether the AD7730 is in its Standby Mode or normal mode of
operation. T he part can be placed in its standby mode using the STANDBY input pin or by
writing 011 to the MD2 to MD0 bits of the Mode Register. T he power-on/reset status of this bit
is 0 assuming the STANDBY pin is high.
SR4
NOREF
MS3–MS0
No Reference Bit. If the voltage between the REF IN(+) and REF IN(–) pins is below 0.3 V, or
either of these inputs is open-circuit, the NOREF bit goes to 1. If NOREF is active on comple-
tion of a conversion, the Data Register is loaded with all 1s. If NOREF is active on completion
of a calibration, updating of the calibration registers is inhibited.
SR3–SR0
T hese bits are for factory use. T he power-on/reset status of these bits vary, depending on the
factory-assigned number.
REV. A
–14–
AD7730/AD7730L
D ata Register (RS2–RS0 = 0, 0, 1); P ower O n/Reset Status: 000000 H ex
T he Data Register on the part is a read-only register which contains the most up-to-date conversion result from the AD7730. Fig-
ure 5 shows a flowchart for reading from the registers on the AD7730. T he register can be programmed to be either 16 bits or 24 bits
wide, determined by the status of the WL bit of the Mode Register. T he RDY output and RDY bit of the Status Register are set low
when the Data Register is updated. T he RDY pin and RDY bit will return high once the full contents of the register (either 16 bits or
24 bits) have been read. If the Data Register has not been read by the time the next output update occurs, the RDY pin and RDY bit
will go high for at least 100 × tCLK IN, indicating when a read from the Data Register should not be initiated to avoid a transfer from
the Data Register as it is being updated. Once the updating of the Data Register has taken place, RDY returns low.
If the Communications Register data sets up the part for a write operation to this register, a write operation must actually take place
in order to return the part to where it is expecting a write operation to the Communications Register (the default state of the inter-
face). However, the 16 or 24 bits of data written to the part will be ignored by the AD7730.
Mode Register (RS2–RS0 = 0, 1, 0); P ower O n/Reset Status: 01B0 H ex
T he Mode Register is a 16-bit register from which data can be read or to which data can be written. T his register configures
the operating modes of the AD7730, the input range selection, the channel selection and the word length of the Data Register.
T able X outlines the bit designations for the Mode Register. MR0 through MR15 indicate the bit location, MR denoting the bits are
in the Mode Register. MR15 denotes the first bit of the data stream. T he number in brackets indicates the power-on/reset default
status of that bit. Figure 5 shows a flowchart for reading from the registers on the AD7730 and Figure 6 shows a flowchart for writ-
ing to the registers on the part.
Table X. Mode Register
MR15
MR14
MR13
MR12
MR11
MR10
MR9
MR8
MD2 (0)
MD1 (0)
MD0 (0)
B/U (0)
DEN (0)
D1 (0)
D0 (0)
WL (1)
MR7
MR6
MR5
MR4
MR3
MR2
MR1
MR0
HIREF (1) ZERO (0)
RN1 (1)
RN0 (1) CLKDIS (0)
BO (0)
CH1 (0)
CH0 (0)
Bit
Bit
Location
Mnem onic
D escription
MR15–MR13
MD2–MD0
Mode Bits. T hese three bits determine the mode of operation of the AD7730 as outlined in
T able XI. T he modes are independent, such that writing new mode bits to the Mode Register
will exit the part from the mode in which it is operating and place it in the new requested mode
immediately after the Mode Register write. T he function of the mode bits is described in more
detail below.
Table XI. O perating Modes
MD 2
MD 1
MD 0
Mode of O peration
0
0
0
0
1
1
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
Sync (Idle) Mode
Continuous Conversion Mode
Single Conversion Mode
Power-Down (Standby) Mode
Internal Zero-Scale Calibration
Internal Full-Scale Calibration
System Zero-Scale Calibration
System Full-Scale Calibration
Power-On/Reset Default
–15–
REV. A
AD7730/AD7730L
MD 2
MD 1 MD 0
O perating Mode
0
0
0
Sync (Idle) Mode. In this mode, the modulator and filter are held in reset mode and the AD7730 is not
processing any new samples or data. Placing the part in this mode is equivalent to exerting the SYNC
input pin. However, exerting the SYNC pin does not actually force these mode bits to 0, 0, 0. T he part
returns to this mode after a calibration or after a conversion in Single Conversion Mode. T his is the
default condition of these bits after Power-On/Reset.
0
0
0
1
0
1
1
0
1
0
1
0
Continuous Conversion Mode. In this mode, the AD7730 is continuously processing data and providing
conversion results to the Data Register at the programmed output update rate (as determined by the
Filter Register). For most applications, this would be the normal operating mode of the AD7730.
Single Conversion Mode. In this mode, the AD7730 performs a single conversion, updates the Data
Register, returns to the Sync Mode and resets the mode bits to 0, 0, 0. T he result of the single conversion
on the AD7730 in this mode will not be provided until the full settling time of the filter has elapsed.
Power-Down (Standby) Mode. In this mode, the AD7730 goes into its power-down or standby state.
Placing the part in this mode is equivalent to exerting the STANDBY input pin. However, exerting
STANDBY does not actually force these mode bits to 0, 1, 1.
Zero-Scale Self-Calibration Mode. T his activates zero-scale self-calibration on the channel selected by
CH1 and CH0 of the Mode Register. T his zero-scale self-calibration is performed at the selected gain on
internally shorted (zeroed) inputs. When this zero-scale self-calibration is complete, the part updates the
contents of the appropriate Offset Calibration Register and returns to Sync Mode with MD2, MD1 and
MD0 returning to 0, 0, 0. T he RDY output and bit go high when calibration is initiated and return low
when this zero-scale self-calibration is complete to indicate that the part is back in Sync Mode and ready
for further operations.
1
1
0
1
1
0
Full-Scale Self-Calibration Mode. T his activates full-scale self-calibration on the channel selected by
CH1 and CH0 of the Mode Register. T his full-scale self-calibration is performed at the selected gain on
an internally-generated full-scale signal. When this full-scale self-calibration is complete, the part updates
the contents of the appropriate Gain Calibration Register and Offset Calibration Register and returns to
Sync Mode with MD2, MD1 and MD0 returning to 0, 0, 0. T he RDY output and bit go high when
calibration is initiated and return low when this full-scale self-calibration is complete to indicate that the
part is back in Sync Mode and ready for further operations.
Zero-Scale System Calibration Mode. T his activates zero scale system calibration on the channel selected
by CH1 and CH0 of the Mode Register. Calibration is performed at the selected gain on the input volt-
age provided at the analog input during this calibration sequence. T his input voltage should remain
stable for the duration of the calibration. When this zero-scale system calibration is complete, the part
updates the contents of the appropriate Offset Calibration Register and returns to Sync Mode with MD2,
MD1 and MD0 returning to 0, 0, 0. T he RDY output and bit go high when calibration is initiated and
return low when this zero-scale calibration is complete to indicate that the part is back in Sync Mode and
ready for further operations.
1
1
1
Full-Scale System Calibration Mode. T his activates full-scale system calibration on the selected input
channel. Calibration is performed at the selected gain on the input voltage provided at the analog input
during this calibration sequence. T his input voltage should remain stable for the duration of the calibra-
tion. When this full-scale system calibration is complete, the part updates the contents of the appropriate
Gain Calibration Register and returns to Sync Mode with MD2, MD1 and MD0 returning to 0, 0, 0.
T he RDY output and bit go high when calibration is initiated and return low when this full-scale calibra-
tion is complete to indicate that the part is back in Sync Mode and ready for further operations.
REV. A
–16–
AD7730/AD7730L
Bit
Bit
Location
Mnem onic D escr iption
MR12
B/U
Bipolar/Unipolar Bit. A 0 in this bit selects bipolar operation and the output coding is 00 . . . 000 for
negative full-scale input, 10 . . . 000 for zero input, and 11 . . . 111 for positive full-scale input. A 1 in
this bit selects unipolar operation and the output coding is 00 . . . 000 for zero input and 11 . . . 111 for
positive full-scale input.
MR11
DEN
Digital Output Enable Bit. With this bit at 1, the AIN2(+)/D1 and AIN2(–)/D0 pins assume their
digital output functions and the output drivers connected to these pins are enabled. In this mode, the
user effectively has two port bits which can be programmed over the serial interface.
MR10–MR9 D1–D0
Digital Output Bits. T hese bits determine the digital outputs on the AIN2(+)/D1 and AIN2(–)/D0 pins,
respectively, when the DEN bit is a 1. For example, a 1 written to the D1 bit of the Mode Register
(with the DEN bit also a 1) will put a logic 1 on the AIN2(+)/D1 pin. T his logic 1 will remain on this
pin until a 0 is written to the D1 bit (in which case the AIN2(+)/D1 pin goes to a logic 0) or the digital
output function is disabled by writing a 0 to the DEN bit.
MR8
MR7
WL
Data Word Length Bit. T his bit determines the word length of the Data Register. A 0 in this bit selects
16-bit word length when reading from the data register (i.e., RDY returns high after 16 serial clock
cycles in the read operation). A 1 in this bit selects 24-bit word length for the Data Register.
HIREF
High Reference Bit. T his bit should be set in accordance with the reference voltage which is being used
on the part. If the reference voltage is 5 V, the HIREF bit should be set to 1. If the reference voltage is
2.5 V, the HIREF bit should be set to a 0. With the HIREF bit set correctly for the appropriate applied
reference voltage, the input ranges are 0 mV to +10 mV, +20 mV, +40 mV and +80 mV for unipolar
operation and ±10 mV, ±20 mV, ±40 mV and ±80 mV for bipolar operation.
It is possible for a user with a 2.5 V reference to set the HIREF bit to a 1. In this case, the part is oper-
ating with a 2.5 V reference but assumes it has a 5 V reference. As a result, the input ranges on the part
become 0 to +5 mV, +10 mV, +20 mV and +40 mV for unipolar operation and ±5 mV, ±10 mV,
±20 mV and ±40 mV for bipolar operation. However, the output noise from the part (in nV) will re-
main unchanged so the resolution of the part (in counts) will halve.
MR6
ZERO
A zero m ust be written to this bit to ensure correct operation of the AD7730.
MR5–MR4
RN1–RN0
Input Range Bits. T hese bits determine the analog input range for the selected analog input. T he dif-
ferent input ranges are outlined in T able XII. T he table is valid for a reference voltage of 5 V with the
HIREF bit at 1, or for a reference voltage of 2.5 V with the HIREF bit at a logic 0.
Table XII. Input Range Selection
Input Range
RN1
RN0
B/U Bit = 0
B/U Bit = 1
0
0
1
1
0
1
0
1
–10 mV to +10 mV
–20 mV to +20 mV
–40 mV to +40 mV
–80 mV to +80 mV
0 mV to +10 mV
0 mV to +20 mV
0 mV to +40 mV
0 mV to +80 mV
Power-On/Reset Default
Note that the input range given in the above table is the range that appears at the input of the PGA
after the DAC offset value has been applied. If the DAC adjusts out no offset (DAC Register is 0010
0000), then this is also the input voltage range at the analog input pins. If, for example, the DAC sub-
tracts out 50 mV of offset and the part is being operated in bipolar mode with RN1 and RN0 at 0, 0,
the actual input voltage range at the analog input is +40 mV to +60 mV.
MR3
CLKDIS
Master Clock Disable Bit. A 1 in the bit disables the master clock from appearing at the MCLK OUT
pin. When disabled, the MCLK OUT pin is forced low. It allows the user the flexibility of using the
MCLK OUT as a clock source for other devices in the system or of turning off the MCLK OUT as a
power saving feature. When using an external master clock at the MCLK IN pin, the AD7730 contin-
ues to have internal clocks and will convert normally with the CLKDIS bit active. When using a crystal
oscillator or ceramic resonator across the MCLK IN and MCLK OUT pins, the AD7730 clock is
stopped and no conversions take place when the CLKDIS bit is active.
–17–
REV. A
AD7730/AD7730L
Bit
Bit
Location
Mnem onic D escr iption
MR2
BO
Burnout Current Bit. A 1 in this bit activates the burnout currents. When active, the burnout currents
connect to the selected analog input pair, one source current to the AIN(+) input and one sink current to
the AIN(–) input. A 0 in this bit turns off the on-chip burnout currents.
MR1–MR0
CH1–CH0
Channel Selection Bits. T hese bits select the analog input channel to be converted or calibrated as
outlined in T able XIII. With CH1 at 1 and CH0 at 0, the part looks at the AIN1(–) input internally
shorted to itself. T his can be used as a test method to evaluate the noise performance of the part with
no external noise sources. In this mode, the AIN1(–) input should be connected to an external voltage
within the allowable common-mode range of the part. T he Offset and Gain Calibration Registers on
the part are paired. T here are three pairs of calibration registers labelled Register Pair 0 through Regis-
ter Pair 2. T hese are assigned to the input channel pairs as outlined in T able XIII.
Table XIII. Channel Selection
Input Channel P air
CH 1
CH 0
P ositive Input
AIN1(+)
Negative Input
AIN1(–)
Calibr ation Register P air
Register Pair 0
0
0
1
1
0
1
0
1
AIN2(+)
AIN2(–)
Register Pair 1
AIN1(–)
AIN1(–)
Register Pair 0
AIN1(–)
AIN2(–)
Register Pair 2
Filter Register (RS2-RS0 = 0, 1, 1); P ower -O n/Reset Status: 200010 H ex
T he Filter Register is a 24-bit register from which data can be read or to which data can be written. T his register determines the
amount of averaging performed by the filter and the mode of operation of the filter. It also sets the chopping mode and the delay
associated with chopping the inputs. T able XIV outlines the bit designations for the Filter Register. FR0 through FR23 indicate the
bit location, FR denoting the bits are in the Filter Register. FR23 denotes the first bit of the data stream. T he number in brackets
indicates the power-on/reset default status of that bit. Figure 5 shows a flowchart for reading from the registers on the AD7730 and
Figure 6 shows a flowchart for writing to the registers on the part.
Table XIV. Filter Register
FR23
FR22
FR21
FR20
FR19
FR18
FR17
FR16
SF11 (0)
SF10 (0)
SF9 (1)
SF8 (0)
SF7 (0)
SF6 (0)
SF5 (0)
SF4 (0)
FR15
FR14
FR13
FR12
FR11
FR10
FR9
FR8
SF3 (0)
SF2 (0)
SF1 (0)
SF0 (0)
ZERO (0) ZERO (0)
SKIP (0)
FAST (0)
FR7
FR6
FR5
FR4
FR3
FR2
FR1
FR0
ZERO (0) ZERO (0)
AC (0)
CHP (1)
DL3 (0)
DL2 (0)
DL1 (0)
DL0 (0)
Bit
Bit
Location
Mnem onic D escr iption
FR23–FR12 SF11–SF0
Sinc3 Filter Selection Bits. T he AD7730 contains two filters: a sinc3 filter and an FIR filter. T he 12 bits
programmed to SF11 through SF0 set the amount of averaging the sinc3 filter performs. As a result,
the number programmed to these 12 bits affects the –3 dB frequency and output update rate from the
part (see Filter Architecture section). T he allowable range for SF words depends on whether the part
is operated with CHOP on or off and SKIP on or off. T able XV outlines the SF ranges for different
setups. All output update rates will be one-half those quoted in T able XV for the AD7730L operating
with a 2.4576 MHz clock.
REV. A
–18–
AD7730/AD7730L
Table XV. SF Ranges
CH O P
SKIP
SF Range
O utput Update Rate Range (Assum ing 4.9152 MH z Clock)
0
1
0
1
0
0
1
1
2048 to 150
2048 to 75
2048 to 40
2048 to 20
150 Hz to 2.048 kHz
50 Hz to 1.365 kHz
150 Hz to 7.6 kHz
50 Hz to 5.12 kHz
Bit
Bit
Location
Mnem onic
D escription
FR11–FR10
FR9
ZERO
SKIP
A zero m ust be written to these bits to ensure correct operation of the AD7730.
FIR Filter Skip Bit. With a 0 in this bit, the AD7730 performs two stages of filtering before
shipping a result out of the filter. T he first is a sinc3 filter followed by a 22-tap FIR filter. With a
1 in this bit, the FIR filter on the part is bypassed and the output of the sinc3 is fed directly
as the output result of the AD7730’s filter (see Filter Architecture for more details on the filter
implementation).
FR8
FAST
FASTStep Mode Enable Bit. A 1 in this bit enables the FASTStep mode on the AD7730. In
this mode, if a step change on the input is detected, the FIR calculation portion of the filter is
suspended and replaced by a simple moving average on the output of the sinc3 filter. Initially,
two outputs from the sinc3 filter are used to calculate an AD7730 output. T he number of sinc3
outputs used to calculate the moving average output is increased (from 2 to 4 to 8 to 16) until
the STDY bit goes low. When the FIR filter has fully settled after a step, the STDY bit will
become active and the FIR filter is switched back into the processing loop (see Filter Architec-
ture section for more details on the FASTStep mode).
FR7–FR6
FR5
ZERO
AC
A zero m ust be written to these bits to ensure correct operation of the AD7730.
AC Excitation Bit. If the signal source to the AD7730 is ac-excited, a 1 must be placed in this
bit. For dc-excited inputs, this bit must be 0. T he ac bit has no effect if CHP is 0. With the ac
bit at 1, the AD7730 assumes that the voltage at the AIN(+)/AIN(–) and REF IN(+)/REF IN(–)
input terminals are reversed on alternate input sampling cycles (i.e. chopped). Note that when
the AD7730 is performing internal zero-scale or full-scale calibrations, the ac bit is treated as a
0, i.e., the device performs these self-calibrations with dc excitation.
FR4
CHP
Chop Enable Bit. T his bit determines if the chopping mode on the part is enabled. A 1 in this
bit location enables chopping on the part. When the chop mode is enabled, the part is effectively
chopped at its input and output to remove all offset and offset drift errors on the part. If offset
performance with time and temperature are important parameters in the design, it is recom-
mended that the user enable chopping on the part. If the input signal is dc-excited, the user has
the option of operating the part in either chop or nonchop mode. If the input signal is ac-excited,
both the ac bit and the CHP bit must be set to 1. T he chop rate on the ACX and ACX signals is
one half of the programmed output rate of the part and thus the chopping frequency varies with
the programmed output rate.
FR3–FR0
DL3–DL0
Delay Selection Bits. These four bits program the delay (in modulator cycles) to be inserted after
each chop edge when the CHP bit is 1. One modulator cycle is MCLK IN/16 and is 3.25 µs at
MCLK IN = 4.9152 MHz. A delay should only be required when in ac mode. Its purpose is to
cater for external delays between the switching signals (ACX and ACX) and when the analog
inputs are actually switched and settled. During the specified number of cycles (between 0 and
15), the modulator is held in reset and the filter does not accept any inputs. If CHP = 1, the
output rate is (MCLK IN/ 16 × (DL + 3 × SF) where DL is the value loaded to bits DL0–DL3.
T he chop rate is always one half of the output rate. T his chop period takes into account the
programmed delay and the fact that the sinc3 filter must settle every chop cycle. With CHP = 0,
the output rate is 1/SF.
–19–
REV. A
AD7730/AD7730L
D AC Register (RS2–RS0 = 1, 0, 0); P ower O n/Reset Status: 20 H ex
T he DAC Register is an 8-bit register from which data can either be read or to which data can be written. T his register provides
the code for the offset-compensation DAC on the part. T able XVI outlines the bit designations for the DAC Register. DR0
through DR7 indicate the bit location, DR denoting the bits are in the DAC Register. DR7 denotes the first bit of the data
stream. T he number in brackets indicates the power-on/reset default status of that bit. Figure 5 shows a flowchart for reading
from the registers on the AD7730 and Figure 6 shows a flowchart for writing to the registers on the part.
Table XVI. D AC Register
DR7
DR6
DR5
DR4
DR3
DR2
DR1
DR0
ZERO (0) ZERO (0) DAC5 (1) DAC4 (0) DAC3 (0) DAC2 (0) DAC1 (0) DAC0 (0)
Bit
Bit
Location
Mnem onic
D escr iption
DR7–DR6
DR5–DR0
ZERO
A zero m ust be written to these bits to ensure correct operation of the AD7730.
DAC5–DAC0 DAC Selection Bits. T hese bits program the output of the offset DAC. T he DAC is effectively
6 bits with one sign bit (DAC5) and five magnitude bits. With DAC5 at 1, the DAC output
subtracts from the analog input before it is applied to the PGA. With DAC5 at 0, the DAC
output adds to the analog input before it is applied to the PGA. T he DAC output is given by
(VREF/62.5) × (D/32) = (VREF/2000) × D where D is the decimal equivalent of bits DAC4 to
DAC0. T hus, for a 5 V reference applied across the REF IN pins, the DAC resolution is 2.5 mV
and offsets in the range –77.5 mV to +77.5 mV can be removed from the analog input signal
before it is applied to the PGA. Note, that the HIREF bit has no effect on the DAC range or
resolution, it controls the ADC range only.
O ffset Calibr ation Register (RS2–RS0 = 1, 0, 1); P ower -O n/Reset Status: 800000 H ex
T he AD7730 contains three 24-bit Offset Calibration Registers, labelled Offset Calibration Register 0 to Offset Calibration Reg-
ister 2, to which data can be written and from which data can be read. T he three registers are totally independent of each other.
T he Offset Calibration Register is used in conjunction with the associated Gain Calibration Register to form a register pair. T he
calibration register pair used to scale the output is as outlined in T able XIII. T he Offset Calibration Register is updated after an
offset calibration routine (1, 0, 0 or 1, 1, 0 loaded to the MD2, MD1, MD0 bits of the Mode Register). During subsequent
conversions, the contents of this register are subtracted from the filter output prior to gain scaling being performed on the word.
Figure 5 shows a flowchart for reading from the registers on the AD7730 and Figure 6 shows a flowchart for writing to the regis-
ters on the part.
Gain Calibr ation Register (RS2–RS0 = 1, 1, 0); P ower -O n/Reset Status: 593CEA
T he AD7730 contains three 24-bit Gain Calibration Registers, labelled Gain Calibration Register 0 to Gain Calibration Register
2, to which data can be written and from which data can be read. T he three registers are totally independent of each other. T he
Gain Calibration Register is used in conjunction with the associated Offset Calibration Register to form a register pair. T he
calibration register pair used to scale the output is as outlined in T able XIII. T he Gain Calibration Register is updated after a
gain calibration routine (1, 0, 1 or 1, 1, 1 loaded to the MD2, MD1, MD0 bits of the Mode Register). During subsequent con-
versions, the contents of this register are used to scale the number which has already been offset corrected with the Offset Cali-
bration Register contents. Figure 5 shows a flowchart for reading from the registers on the AD7730 and Figure 6 shows a
flowchart for writing to the registers on the part.
Test Register (RS2–RS0 = 1, 1, 1); P ower -O n/Reset Status: 000000H ex
T he AD7730 contains a 24-bit T est Register to which data can be written and from which data can be read. T he contents of this
Test Register are used in testing the device. The user is advised not to change the status of any of the bits in this register from the
default (Power-On or RESET) status of all 0s as the part will be placed in one of its test modes and will not operate correctly. If the
part enters one of its test modes, exercising RESET or writing 32 successive 1s to the part will exit the AD7730 from the mode and
return all register contents to their power-on/reset status. Note, if the part is placed in one of its test modes, it may not be possible to
read back the contents of the Test Register depending on the test mode in which the part has been placed.
REV. A
–20–
AD7730/AD7730L
READ ING FRO M AND WRITING TO TH E O N-CH IP REGISTERS
T he AD7730 contains a total of thirteen on-chip registers. T hese registers are all accessed over a three-wire interface. As a result,
addressing of registers is via a write operation to the topmost register on the part, the Communications Register. Figure 5 shows a
flowchart for reading from the different registers on the part summarizing the sequence and the words to be written to access each of
the registers. Figure 6 gives a flowchart for writing to the different registers on the part, again summarizing the sequence and words
to be written to the AD7730.
START
Byte W
(H ex)
Byte Y
(H ex)
Byte Z
(H ex)
Register
Status Register
Data Register
Mode Register
Filter Register
DAC Register
Offset Register
Gain Register
T est Register
10
11
12
13
14
15
16
17
20
21
22
N/A*
N/A*
N/A*
N/A*
N/A*
30
30
30
N/A*
N/A*
N/A*
N/A*
N/A*
CONTINUOUS
READS OF
REGISTER
YES
REQUIRED?
NO
WRITE BYTE W TO
COMMUNICATIONS REGISTER
(SEE ACCOMPANYING TABLE)
WRITE BYTE Y TO
COMMUNICATIONS REGISTER
(SEE ACCOMPANYING TABLE)
*N/A= Not Applicable. Continuous reads of these registers does not make sense
as the register contents would remain the same since they are only changed by a
write operation.
READ REGISTER
READ REGISTER
STOP
CONTINUOUS
NO
READ
OPERATION?
YES
WRITE BYTE Z TO
COMMUNICATIONS REGISTER
(SEE ACCOMPANYING TABLE)
Figure 5. Flowchart for Reading from the AD7730 Registers
START
Register
Byte Y (H ex)
Communications Register
Data Register
Mode Register
Filter Register
DAC Register
00
Read Only Register
02
03
04
05
06
WRITE BYTE Y TO
COMMUNICATIONS REGISTER
(SEE ACCOMPANYING TABLE)
WRITE TO REGISTER
Offset Register
Gain Register
T est Register
User is advised not to change
contents of T est Register.
END
Figure 6. Flowchart for Writing to the AD7730 Registers
–21–
REV. A
AD7730/AD7730L
CALIBRATIO N O P ERATIO N SUMMARY
T he AD7730 contains a number of calibration options as outlined previously. T able XVII summarizes the calibration types, the
operations involved and the duration of the operations. T here are two methods of determining the end of calibration. T he first is to
monitor the hardware RDY pin using either interrupt-driven or polling routines. T he second method is to do a software poll of the
RDY bit in the Status Register. T his can be achieved by setting up the part for continuous reads of the Status Register once a calibra-
tion has been initiated. T he RDY pin and RDY bit go high on initiating a calibration and return low at the end of the calibration
routine. At this time, the MD2, MD1, MD0 bits of the Mode Register have returned to 0, 0, 0. T he FAST and SKIP bits are treated
as 0 for the calibration sequence so the full filter is always used for the calibration routines. See Calibration section for full detail.
Table XVII. Calibration O perations
MD 2, MD 1,
MD 0
D uration to RDY
Low (CH P = 1)
D uration to RDY
Low (CH P = 0)
Calibration Type
Calibration Sequence
Internal Zero-Scale 1, 0, 0
22 × 1/Output Rate
24 × 1/Output Rate
Calibration on internal shorted input with PGA set for
selected input range. T he ac bit is ignored for this calibra-
tion sequence. The sequence is performed with dc excitation.
The Offset Calibration Register for the selected channel is
updated at the end of this calibration sequence. For full self-
calibration, this calibration should be preceded by an Internal
Full-Scale calibration. For applications which require an
Internal Zero-Scale and System Full-Scale calibration, this
Internal Zero-Scale calibration should be performed first.
Internal Full-Scale
1, 0, 1
44 × 1/Output Rate
48 × 1/Output Rate
Calibration on internally-generated input full-scale with
PGA set for selected input range. T he ac bit is ignored for
this calibration sequence. T he sequence is performed with
dc excitation. T he Gain Calibration Register for the
selected channel is updated at the end of this calibration
sequence. It is recommended that internal full-scale
calibrations are performed on the 80 mV range, regardless
of the subsequent operating range, to optimize the post-
calibration gain error. T his calibration should be followed
by either an Internal Zero-Scale or System Zero-Scale
calibration. T his zero-scale calibration should be
performed at the operating input range.
System Zero-Scale
1, 1, 0
22 × 1/Output Rate
24 × 1/Output Rate
Calibration on externally applied input voltage with PGA
set for selected input range. T he input applied is assumed
to be the zero scale of the system. If ac = 1, the system
continues to use ac excitation for the duration of the
calibration. For full system calibration, this System Zero-
Scale calibration should be performed first. For applications
which require a System Zero-Scale and Internal Full-Scale
calibration, this calibration should be preceded by the
Internal Full-Scale calibration. T he Offset Calibration
Register for the selected channel is updated at the end of
this calibration sequence.
System Full-Scale
1, 1, 1
22 × 1/Output Rate
24 × 1/Output Rate
Calibration on externally-applied input voltage with PGA
set for selected input range. T he input applied is assumed
to be the full-scale of the system. If ac = 1, the system
continues to use ac excitation for the duration of the
calibration. T his calibration should be preceded by a
System Zero-Scale or Internal Zero-Scale calibration. T he
Gain Calibration Register for the selected channel is
updated at the end of this calibration sequence.
REV. A
–22–
AD7730/AD7730L
CIRCUIT D ESCRIP TIO N
registers. T he output noise performance and peak-to-peak reso-
lution of the part varies with gain and with the output rate as
shown in T ables I to IV.
T he AD7730 is a sigma-delta A/D converter with on-chip digital
filtering, intended for the measurement of wide dynamic range,
low-frequency signals such as those in weigh-scale, strain-gage,
pressure transducer or temperature measurement applications.
It contains a sigma-delta (or charge-balancing) ADC, a calibra-
tion microcontroller with on-chip static RAM, a clock oscillator,
a digital filter and a bidirectional serial communications port.
The part consumes 13 mA of power supply current with a standby
mode which consumes only 25 µA. The part operates from a single
+5 V supply. The clock source for the part can be provided via an
external clock or by connecting a crystal oscillator or ceramic
resonator across the MCLK IN and MCLK OUT pins.
T he analog inputs are buffered on-chip allowing the part to
handle significant source impedances on the analog input. T his
means that external R, C filtering (for noise rejection or RFI
reduction) can be placed on the analog inputs if required. Both
analog channels are differential, with a common-mode voltage
range that comes within 1.2 V of AGND and 0.95 V of AVDD
T he reference input is also differential and the common-mode
range here is from AGND to AVDD
.
.
T he part contains a 6-bit DAC that is controlled via on-chip
registers. T his DAC can be used to remove T ARE values of up
to ±80 mV from the analog input signal range. T he resolution
on this T ARE function is 1.25 mV for a +2.5 V reference and
2.5 mV with a +5 V reference.
The part contains two programmable-gain fully differential analog
input channels. The part handles a total of eight different input
ranges which are programmed via the on-chip registers. There are
four differential unipolar ranges: 0 mV to +10 mV, 0 mV to
+20 mV, 0 mV to +40 mV and 0 mV to +80 mV and four differen-
tial bipolar ranges: ±10 mV, ±20 mV, ±40 mV and ±80 mV.
T he AD7730 can accept input signals from a dc-excited bridge.
It can also handle input signals from an ac-excited bridge by
using the ac excitation clock signals (ACX and ACX) to switch
the supplies to the bridge. ACX and ACX are nonoverlapping
clock signals used to synchronize the external ac supplies that
drive the transducer bridge. T hese ACX clocks are demodulated
on the AD7730 input.
T he AD7730 employs a sigma-delta conversion technique to
realize up to 24 bits of no missing codes performance. T he
sigma-delta modulator converts the sampled input signal into a
digital pulse train whose duty cycle contains the digital informa-
tion. A digital low-pass filter processes the output of the sigma-
delta modulator and updates the data register at a rate that can
be programmed over the serial interface. T he output data from
the part is accessed over this serial interface. The cutoff frequency
and output rate of this filter can be programmed via on-chip
The AD7730 contains a number of hardware and software
events that set or reset status flags and bits in registers. T able
XVIII summarizes which blocks and flags are affected by the
different events.
Table XVIII. Reset Events
Set Registers
to D efault
Mode
Bits
Filter
Reset
Analog
P ower-D own
Reset Serial
Interface
Set RDY
P in/Bit
Set STDY
Bit
Event
Power-On Reset
RESET Pin
STANDBY Pin
Mode 011 Write
SYNC Pin
Mode 000 Write
Conversion or
Cal Mode Write
Clock 32 1s
Yes
Yes
No
No
No
No
No
000
000
As Is
011
As Is
000
New
Value
000
Yes
Yes
Yes
Yes
Yes
Yes
Initial
Reset
Yes
No
Yes
No
Yes
Yes
No
No
No
Yes
Yes
No
No
No
No
No
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
Yes
No
No
No
Yes
No
Yes
Yes
Yes
No
Data Register Read
As Is
–23–
REV. A
AD7730/AD7730L
ANALO G INP UT
Offset DAC
Analog Input Channels
T he purpose of the Offset DAC is to either add or subtract an
offset so the input range at the input to the PGA is as close as
possible to the nominal. If the output of the 6-bit Offset DAC is
0 V, the differential voltage ranges that appear at the analog
input to the part will also appear at the input to the PGA. If,
however, the Offset DAC has an output voltage other than 0 V,
the input range to the analog inputs will differ from that applied
to the input of the PGA.
T he AD7730 contains two differential analog input channels, a
primary input channel, AIN1, and a secondary input channel,
AIN2. T he input pairs provide programmable gain, differential
channels which can handle either unipolar or bipolar input
signals. It should be noted that the bipolar input signals are
referenced to the respective AIN(–) input of the input pair. T he
secondary input channel can also be reconfigured as two digital
output port bits.
T he Offset DAC has five magnitude bits and one sign bit. T he
sign bit determines whether the value loaded to the five magni-
tude bits is added to or subtracted from the voltage at the ana-
log input pins. Control of the Offset DAC is via the DAC
Register which is discussed previously in the On-Chip Registers
section. With a 5 V reference applied between the REF IN pins,
the resolution of the Offset DAC is 2.5 mV with a range that
allows addition or subtraction of 77.5 mV. With a 2.5 V refer-
ence applied between the REF IN pins, the resolution of the
Offset DAC is 1.25 mV with a range that allows addition or
subtraction of 38.75 mV.
A two-channel differential multiplexer switches one of the two
input channels to the on-chip buffer amplifier. T his multiplexer
is controlled by the CH0 and CH1 bits of the Mode Register.
When the analog input channel is switched, the RDY output
goes high and the settling time of the part must elapse before a
valid word from the new channel is available in the Data Regis-
ter (indicated by RDY going low).
Buffer ed Inputs
T he output of the multiplexer feeds into a high impedance input
stage of the buffer amplifier. As a result, the analog inputs can
handle significant source impedances. T his buffer amplifier has
an input bias current of 50 nA (CHP = 1) and 60 nA (CHP = 0).
T his current flows in each leg of the analog input pair. T he
offset current on the part is the difference between the input
bias on the legs of the input pair. T his offset current is less than
10 nA (CHP = 1) and 30 nA (CHP = 0). Large source resis-
tances result in a dc offset voltage developed across the source
resistance on each leg, but matched impedances on the analog
input legs will reduce the offset voltage to that generated by the
input offset current.
Following is an example of how the Offset DAC works. If the
differential input voltage range the user had at the analog input
pins was +20 mV to +30 mV, the Offset DAC should be pro-
grammed to subtract 20 mV of offset so the input range to the
PGA is 0 mV to +10 mV. If the differential input voltage range
the user had at the analog input pins was –60 mV to +20 mV,
the Offset DAC should be programmed to add 20 mV of offset so
the input range to the PGA is ±40 mV.
Bipolar /Unipolar Inputs
T he analog inputs on the AD7730 can accept either unipolar or
bipolar input voltage ranges. Bipolar input ranges do not imply
that the part can handle negative voltages with respect to system
ground on its analog inputs unless the AGND of the part is also
biased below system ground. Unipolar and bipolar signals on
the AIN(+) input are referenced to the voltage on the respective
AIN(–) input. For example, if AIN(–) is +2.5 V and the AD7730 is
configured for an analog input range of 0 to +10 mV with no
DAC offset correction, the input voltage range on the AIN(+)
input is +2.5 V to +2.51 V. Similarly, if AIN(–) is +2.5 V and the
AD7730 is configured for an analog input range of ±80 mV
with no DAC offset correction, the analog input range on the
AIN(+) input is +2.42 V to +2.58 V (i.e., 2.5 V ± 80 mV).
Analog Input Ranges
T he absolute input voltage range is restricted to between
AGND + 1.2 V to AVDD – 0.95 V, which also places restrictions
on the common-mode range. Care must be taken in setting up
the common-mode voltage and input voltage range so these
limits are not exceeded, otherwise there will be a degradation in
linearity performance.
In some applications, the analog input range may be biased
either around system ground or slightly below system ground. In
such cases, the AGND of the AD7730 must be biased negative
with respect to system ground so the analog input voltage does
not go within 1.2 V of AGND. Care should taken to ensure that
the differential between either AVDD or DVDD and this biased
AGND does not exceed 5.5 V. T his is discussed in more detail
in the Applications section.
Bipolar or unipolar options are chosen by programming the B/U
bit of the Mode Register. T his programs the selected channel
for either unipolar or bipolar operation. Programming the chan-
nel for either unipolar or bipolar operation does not change any
of the input signal conditioning; it simply changes the data
output coding and the points on the transfer function where
calibrations occur. When the AD7730 is configured for unipolar
operation, the output coding is natural (straight) binary with a
zero differential voltage resulting in a code of 000 . . . 000, a
midscale voltage resulting in a code of 100 . . . 000 and a full-
scale input voltage resulting in a code of 111 . . . 111. When the
AD7730 is configured for bipolar operation, the coding is offset
binary with a negative full scale voltage resulting in a code of
000 . . . 000, a zero differential voltage resulting in a code of
100 . . . 000 and a positive full scale voltage resulting in a code
of 111 . . . 111.
Pr ogr a m m a ble Ga in Am plifier
T he output from the buffer amplifier is summed with the output
of the 6-bit Offset DAC before it is applied to the input of the
on-chip programmable gain amplifier (PGA). T he PGA can
handle four different unipolar input ranges and four bipolar
ranges. With the HIREF bit of the Mode Register at 0 and a
+2.5 V reference (or the HIREF bit at 1 and a +5 V reference),
the unipolar ranges are 0 mV to +10 mV, 0 mV to +20 mV,
0 mV to +40 mV, and 0 mV to +80 mV, while the bipolar ranges
are ±10 mV, ±20 mV, ±40 mV and ±80 mV. T hese are the
nominal ranges that should appear at the input to the on-chip
PGA.
REV. A
–24–
AD7730/AD7730L
Bur nout Cur r ents
AV
DV
DD
DD
T he AD7730 contains two 100 nA constant current generators,
one source current from AVDD to AIN(+) and one sink current
from AIN(–) to AGND. T he currents are switched to the se-
lected analog input pair. Both currents are either on or off,
depending on the BO bit of the Mode Register. T hese currents
can be used in checking that a transducer is still operational
before attempting to take measurements on that channel. If the
currents are turned on, allowed flow in the transducer, a mea-
surement of the input voltage on the analog input taken and the
voltage measured is full scale, it indicates that the transducer
has gone open-circuit. If the voltage measured is 0 V, it indicates
that the transducer has gone short circuit. For normal operation,
these burnout currents are turned off by writing a 0 to the BO
bit. T he current sources work over the normal absolute input
voltage range specifications.
REF IN(+)
REF IN(–)
IN+
IN–
EXCITATION
VOLTAGE = +5V
AD7730
AIN1(+)
AIN1(–)
OUT–
OUT+
AGND
DGND
Figure 7. Ratiom etric Generation of Reference in DC-
Excited Bridge Application
AV
DV
DD
DD
REF IN(+)
REF IN(–)
REFERENCE INP UT
T he AD7730’s reference inputs, REF IN(+) and REF IN(–),
provide a differential reference input capability. T he common-
mode range for these differential inputs is from AGND to
AVDD. T he nominal reference voltage, VREF (REF IN(+)—
REF IN(–)), for specified operation is +2.5 V with the HIREF
bit at 0 V and +5 V with the HIREF bit at 1. T he part is also
functional with VREF of +2.5 V with the HIREF bit at 1. T his
results in a halving of all input ranges. T he resolution in nV will
be unaltered but will appear halved in terms of counts.
IN+
IN–
AIN1(+)
AIN1(–)
EXCITATION
VOLTAGE = +5V
AD7730
OUT–
OUT+
ACX
AC
EXCITATION
CLOCK
ACX
Both reference inputs provide a high impedance, dynamic load.
T he typical average dc input leakage current over temperature
is 8.5 µA with HIREF = 1 and VREF = +5 V, and 2.5 µA with
HIREF = 0 and VREF = +2.5 V. Because the input impedance of
each reference input is dynamic, external resistance/capacitance
combinations on these inputs may result in gain errors on the
part.
AGND
DGND
Figure 8. Ratiom etric Generation of Reference in AC-
Excited Bridge Application
application. In this case, both the reference voltage for the part
and the excitation voltage for the transducer are chopped. Once
again, the HIREF bit should be set to 1.
T he AD7730 can be operated in either ac or dc mode. If the
bridge excitation is fixed dc, the AD7730 should be operated in
dc mode. If the analog input and the reference inputs are externally
chopped before being applied to the part the AD7730 should be
operated in ac mode and not dc mode. In ac mode, it is assumed
that both the analog inputs and reference inputs are chopped
and as a result change phase every alternate chopping cycle. If
the chopping is synchronized by the AD7730 (using the ACX
signals to control the chopping) the part then takes into account
the reversal of the analog input and reference input signals.
If the AD7730 is not used in a ratiometric application, a low
noise reference should be used. Recommended 2.5 V reference
voltage sources for the AD7730 include the AD780, REF43
and REF192. If any of these references are used as the reference
source for the AD7730, the HIREF bit should be set to 0. It is
generally recommended to decouple the output of these references
to further reduce the noise level.
Refer ence D etect
T he AD7730 includes on-chip circuitry to detect if the part
has a valid reference for conversions or calibrations. If the volt-
age between the REF IN(+) and REF IN(–) pins goes below
0.3 V or either the REF IN(+) or REF IN(–) inputs is open
circuit, the AD7730 detects that it no longer has a valid reference.
In this case, the NO REF bit of the Status Register is set to a 1.
T he output noise performance outlined in T ables I through IV
is for an analog input of 0 V and is unaffected by noise on the
reference. T o obtain the same noise performance as shown in
the noise tables over the full input range requires a low noise
reference source for the AD7730. If the reference noise in the
bandwidth of interest is excessive, it will degrade the performance
of the AD7730. In applications where the excitation voltage for
the bridge transducer on the analog input also drives the refer-
ence voltage for the part, the effect of the noise in the excita-
tion voltage will be removed as the application is ratiometric.
Figure 7 shows how the reference voltage can be connected in a
ratiometric fashion in a dc-excited bridge application. In this
case, the excitation voltage for the AD7730 and the transducer
is a dc voltage. T he HIREF bit of the Mode Register should be
set to 1. Figure 8 meanwhile shows how the reference can be
connected in a ratiometric fashion in an ac-excited bridge
If the AD7730 is performing normal conversions and the NO
REF bit becomes active, the part places all ones in the Data
Register. T herefore, it is not necessary to continuously monitor
the status of the NO REF bit when performing conversions. It is
only necessary to verify its status if the conversion result read
from the Data Register is all 1s.
–25–
REV. A
AD7730/AD7730L
If the AD7730 is performing either an offset or gain calibration
and the NOREF bit becomes active, the updating of the respec-
tive calibration register is inhibited to avoid loading incorrect
coefficients to this register. If the user is concerned about verify-
ing that a valid reference is in place every time a calibration is
performed, then the status of the NOREF bit should be checked
at the end of the calibration cycle.
Fir st Stage Filter
T he first stage filter is a low-pass, sinc3 or (sinx/x)3 filter whose
primary function is to remove the quantization noise introduced
at the modulator. T he cutoff frequency and output rate of this
filter is programmed via the SF0 to SF11 bits of the Filter Reg-
ister. T he frequency response for this first stage filter is shown in
Figure 10. T he response of this first stage filter is similar to that
of an averaging filter but with a sharper roll-off. T he output rate
for the filter corresponds with the positioning of the first notch
of the filter’s frequency response. T hus, for the plot of Figure 10,
where the output rate is 600 Hz (fCLK IN = 4.9152 MHz and
SF = 512), the first notch of the filter is at 600 Hz. T he notches
of this sinc3 filter are repeated at multiples of the first notch. The
filter provides attenuation of better than 100 dB at these notches.
Programming a different cutoff frequency via SF0 – SF11 does
not alter the profile of the filter response; it changes the fre-
quency of the notches as outlined in the Filter Registers section.
T his response is repeated at either side of the input sampling
frequency (307 kHz) and at either side of multiples of the input
sampling frequency.
SIGMA-D ELTA MO D ULATO R
A sigma-delta ADC generally consists of two main blocks, an
analog modulator and a digital filter. In the case of the AD7730,
the analog modulator consists of a difference amplifier, an inte-
grator block, a comparator and a feedback DAC as illustrated in
Figure 9. In operation, the analog signal sample is fed to the
difference amplifier along with the output of the feedback DAC.
T he difference between these two signals is integrated and fed to
the comparator. T he output of the comparator provides the
input to the feedback DAC so that the system functions as a
negative feedback loop that tries to minimize the difference
signal. T he digital data that represents the analog input voltage
is contained in the duty cycle of the pulse train appearing at the
output of the comparator. T his duty cycle data can be recovered
as a data word using the digital filter. T he sampling frequency of
the modulator loop is many times higher than the bandwidth of
the input signal. T he integrator in the modulator shapes the
quantization noise (which results from the analog-to-digital
conversion) so that the noise is pushed toward one half of the
modulator frequency. T he digital filter then bandlimits the re-
sponse to a frequency significantly lower than one half of the
modulator frequency. In this manner, the 1-bit output of the
comparator is translated into a bandlimited, low noise output
from the AD7730.
0
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ANALOG DIFFERENCE
INPUT
AMP
COMPARATOR
0
200
400
600
800 1000 1200 1400 1600 1800
FREQUENCY – Hz
DIGITAL
FILTER
INTEGRATOR
DAC
Figure 10. Frequency Response of First Stage Filter
T he first stage filter has two basic modes of operation. T he
primary mode of operation for weigh-scale applications is chop
mode, which is achieved by placing a 1 in the CHP bit of the
Filter Register. T he part should be operated in this mode when
drift and noise rejection are important criteria in the application.
T he alternative mode of operation is the nonchop mode, with
CHP at 0, which would be used when higher throughput rates
are a concern or in applications where the reduced rejection at
the chopping frequency in chop mode is an issue.
DIGITAL DATA
Figure 9. Sigm a-Delta Modulator Block Diagram
D IGITAL FILTERING
Filter Ar chitectur e
T he output of the modulator feeds directly into the digital filter.
T his digital filter consists of two portions, a first stage filter and
a second stage filter. T he first stage filter is a sinc3, low-pass
filter. T he cutoff frequency and output rate of this first stage
filter is programmable. T he second stage filter has three distinct
modes of operation. In its normal mode, it provides a low-pass
FIR filter that processes the output of the first stage filter. When
a step change is detected on the analog input, this second stage
filter enters a second mode where it performs a variable number
of averages for some time after the step change and then the
second stage filter switches back to the FIR filter. T he third
option for the second stage filter is that it is completely bypassed
so the only filtering provided on the AD7730 is the first stage.
T he various filter stages and options are discussed in the follow-
ing sections.
Nonchop Mode
With chop mode disabled on the AD7730, the first stage filter
continuously processes input data and produces a result at an
output rate determined by the SF word. Operating in nonchop
mode can result in a 20% reduction in noise for a given band-
width, but without the excellent drift and noise rejection ben-
efits which accrue from chopping the part. T he output update
and first notch of this first stage filter correspond and are deter-
mined by the relationship:
fCLK IN
1
Output Rate =
×
16
SF
where SF is the decimal equivalent of the data loaded to the SF
bits of the Filter Register and fCLK IN is the master clock frequency.
REV. A
–26–
AD7730/AD7730L
0
Chop Mode
–10
With chop mode enabled on the AD7730, the signal processing
chain is synchronously chopped at the analog input and at the
output of the first stage filter. T his means that for each output
of the first stage filter to be computed, the full settling time of
the filter has to elapse. T his results in an output rate from the
filter that is three times lower than for a given SF word than for
nonchop mode. T he output update and first notch of this first
stage filter correspond and are determined by the relationship:
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fCLK IN
1
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Output Rate =
×
16
3 ×SF
where SF is the decimal equivalent of the data loaded to the SF
bits of the Filter Register and fCLK IN is the master clock frequency.
0
10
20
30
40
50
60
70
80
90 100
FREQUENCY – Hz
Figure 11. Detailed Full Frequency Response of AD7730
(Second Stage Filter as Norm al FIR, Chop Enabled)
Second Stage Filter
As stated earlier, the second stage filter has three distinct modes
of operation which result in a different overall filter profile for
the part. T he modes of operation of the second stage filter are
discussed in the following sections along with the different filter
profiles which result.
Figure 12 shows the frequency response for the same set of
conditions as for Figure 11, but in this case the response is
shown out to 600 Hz. T his response shows that the attenuation
of input frequencies close to 200 Hz and 400 Hz is significantly
less than at other input frequencies. T hese “peaks” in the fre-
quency response are a by-product of the chopping of the input.
T he plot of Figure 12 is the amplitude for different input fre-
quencies. Note that because the output rate is 200 Hz for the
conditions under which Figure 12 is plotted, if something ex-
isted in the input frequency domain at 200 Hz, it would be
aliased and appear in the output frequency domain at dc.
Nor m a l FIR Oper a tion
T he normal mode of operation of the second stage filter is as a
22-tap low-pass FIR filter. T his second stage filter processes the
output of the first stage filter and the net frequency response of
the filter is simply a product of the filter response of both filters.
T he overall filter response of the AD7730 is guaranteed to have
no overshoot.
Figure 11 shows the full frequency response of the AD7730 when
the second stage filter is set for normal FIR operation. T his
response is for chop mode enabled with the decimal equivalent
of the word in the SF bits set to 512 and a master clock frequency
of 4.9152 MHz. T he response will scale proportionately with
master clock frequency. T he response is shown from dc to
100 Hz. T he rejection at 50 Hz ±1 Hz and 60 Hz ± 1 Hz is
better than 88 dB.
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–70
T he –3 dB frequency for the frequency response of the AD7730
with the second stage filter set for normal FIR operation and
chop mode enabled is determined by the following relationship:
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fCLK IN
1
f3 dB = 0.0395 ×
×
16
3 ×SF
0
50 100 150 200 250 300 350 400 450 500 550 600
FREQUENCY – Hz
In this case, f3 dB = 7.9 Hz and the stopband, where the attenua-
tion is greater than 64.5 dB, is determined by:
Figure 12. Expanded Full Frequency Response of AD7730
(Second Stage Filter as Norm al FIR, Chop Enabled)
f CLK IN
1
fSTOP = 0.14 ×
×
16
3 ×SF
In this case, fSTOP = 28 Hz.
–27–
REV. A
AD7730/AD7730L
Because of this effect, care should be taken in choosing an out-
put rate that is close to the line frequency in the application. If
the line frequency is 50 H z, an output update rate of 50 H z
should not be chosen as it will significantly reduce the AD7730’s
line frequency rejection (the 50 Hz will appear as a dc effect
with only 6 dB attenuation). Choosing an output rate of 55 Hz
will result in a 6 dB—attenuated aliased frequency of 5 Hz with
only a further 25 dB attenuation based on the filter profile. T his
number is based on the filter roll-off and Figure 11 can be used
as a reference by dividing the frequency scale by a factor of 4.
Choosing 57 Hz as the output rate will give better than 90 dB
attenuation of the aliased line frequency which appears as a
7 Hz signal. Similarly, multiples of the line frequency should be
avoided as the output rate because harmonics of the line fre-
quency will not be fully attenuated. T he programmability of the
AD7730’s output rate should allow the user to readily choose an
output rate that overcomes this issue. An alternative is to use
the part in nonchop mode.
T he –3 dB frequency for the frequency response of the AD7730
with the second stage filter set for normal FIR operation and
chop mode enabled, is determined by the following relationship:
fCLK IN
1
f3dB = 0.039 ×
×
16
SF
In this case, f3 dB = 7.8 Hz and the stop band, where the attentua-
tion is greater than 64.5 dB, is determined by:
f CLK IN
1
fSTOP = 0.14 ×
In this case, f3 dB = 28 Hz.
×
16
SF
Figure 14 shows the frequency response for the same set of
conditions as for Figure 13, but in this case the response is
shown out to 600 Hz. T his plot is comparable to that of Figure
12. T he most notable difference is the absence of the peaks in
the response at 200 Hz and 400 Hz. As a result, interference at
these frequencies will be effectively eliminated before being
aliased back to dc.
Figure 13 shows the frequency response for the AD7730 with
the second stage filter set for normal FIR operation, chop mode
disabled, the decimal equivalent of the word in the SF bits set to
1536 and a master clock frequency of 4.9152 MHz. The response
is analogous to that of Figure 11, with the three-times-larger SF
word producing the same 200 Hz output rate. Once again, the
response will scale proportionally with master clock frequency.
T he response is shown from dc to 100 Hz. T he rejection at
50 Hz ± 1 Hz, and 60 Hz ± 1 Hz is better than 88 dB.
0
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0
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0
50 100 150 200 250 300 350 400 450 500 550 600
FREQUENCY – Hz
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Figure 14. Expanded Full Frequency Response of AD7730
(Second Stage Filter as Norm al FIR, Chop Disabled)
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0
10
20
30
40
50
60
70
80
90 100
FREQUENCY – Hz
Figure 13. Detailed Full Frequency Response of AD7730
(Second Stage Filter as Norm al FIR, Chop Disabled)
REV. A
–28–
AD7730/AD7730L
FASTStep Mode
In FASTStep mode, the part has settled to the new value much
faster. With chopping enabled, the FASTStep mode settles to
its value in two outputs, while the normal mode settling takes
23 outputs. Between the second and 23rd output, the FASTStep
mode produces a settled result, but with additional noise com-
pared to the specified noise level for its operating conditions. It
starts at a noise level that is comparable to SKIP mode and as
the averaging increases ends up at the specified noise level. T he
complete settling time to where the part is back within the
specified noise number is the same for FASTStep mode and
normal mode. As can be seen from Figure 13, the FASTStep
mode gives a much earlier indication of where the output chan-
nel is going and its new value. T his feature is very useful in
weighing applications to give a much earlier indication of the
weight, or in an application scanning multiple channels where
the user does not have to wait the full settling time to see if a
channel has changed value.
T he second mode of operation of the second stage filter is in
FASTStep mode which enables it to respond rapidly to step
inputs. T his FASTStep mode is enabled by placing a 1 in the
FAST bit of the Filter Register. If the FAST bit is 0, the part
continues to process step inputs with the normal FIR filter as
the second stage filter. With FASTStep mode enabled, the
second stage filter will continue to process steady state inputs
with the filter in its normal FIR mode of operation. However,
the part is continuously monitoring the output of the first stage
filter and comparing it with the second previous output. If the
difference between these two outputs is greater than a predeter-
mined threshold (1% of full scale), the second stage filter switches
to a simple moving average computation. When the step change
is detected, the STDY bit of the Status Register goes to 1 and
will not return to 0 until the FIR filter is back in the processing
loop.
T he initial number of averages in the moving average computa-
tion is either 2 (chop enabled) or 1 (chop disabled). T he num-
ber of averages will be held at this value as long as the threshold
is exceeded. Once the threshold is no longer exceeded (the step
on the analog input has settled), the number of outputs used to
compute the moving average output is increased. T he first and
second outputs from the first stage filter where the threshold is
no longer exceeded is computed as an average by two, then four
outputs with an average of four, eight outputs with an average of
eight, and six outputs with an average of 16. At this time, the
second stage filter reverts back to its normal FIR mode of opera-
tion. When the second stage filter reverts back to the normal FIR,
the STDY bit of the Status Register goes to 0.
SKIP Mode
T he final method for operating the second stage filter is where it
is bypassed completely. T his is achieved by placing a 1 in the
SKIP bit of the Filter Register. When SKIP mode is enabled, it
means that the only filtering on the part is the first stage, sinc3,
filter. As a result, the complete filter profile is as described ear-
lier for the first stage filter and illustrated in Figure 10.
In SKIP mode, because there is much less processing of the data
to derive each individual output, the normal mode settling time
for the part is shorter. As a consequence of the lesser filtering,
however, the output noise from the part will be significantly
higher for a given SF word. For example with a 20 mV, an SF
word of 1536 and CHP = 0, the output rms noise increases
from 80 nV to 200 nV. With a 10 mV input range, an SF word
of 1024 and CHP = 1, the output rms noise goes from 60 nV to
200 nV.
Figure 15 shows the different responses to a step input with
FASTStep mode enabled and disabled. T he vertical axis shows
the code value returned by the AD7730 and indicates the set-
tling of the output to the input step change. The horizontal axis
shows the number of outputs it takes for that settling to occur.
With chopping disabled and SKIP mode enabled, each output
from the AD7730 is a valid result in itself. However, with chop-
ping enabled and SKIP mode enabled, the outputs from the
AD7730 must be handled in pairs as each successive output is
from reverse chopping polarities.
T he positive input step change occurs at the fifth output. In
FASTStep mode, the output has settled to the final value by the
eighth output. In normal mode, the output has not reached
close to its final value until after the 25th output.
CALIBRATIO N
20000000
15000000
10000000
5000000
0
T he AD7730 provides a number of calibration options which
can be programmed via the MD2, MD1 and MD0 bits of the
Mode Register. T he different calibration options are outlined in
the Mode Register and Calibration Operations sections. A cali-
bration cycle may be initiated at any time by writing to these
bits of the Mode Register. Calibration on the AD7730 removes
offset and gain errors from the device.
T he AD7730 gives the user access to the on-chip calibration
registers allowing the microprocessor to read the device’s cali-
bration coefficients and also to write its own calibration coeffi-
cients to the part from prestored values in E2PROM. T his gives
the microprocessor much greater control over the AD7730’s
calibration procedure. It also means that the user can verify that
the device has performed its calibration correctly by comparing
the coefficients after calibration with prestored values in
E2PROM. T he values in these calibration registers are 24 bits
wide. In addition, the span and offset for the part can be adjusted
by the user.
0
5
10
15
20
25
NUMBER OF OUTPUTS
Figure 15. Step Response for FASTStep and Norm al
Operation
–29–
REV. A
AD7730/AD7730L
Internally in the AD7730, the coefficients are normalized before
being used to scale the words coming out of the digital filter.
T he offset calibration register contains a value which, when
normalized, is subtracted from all conversion results. T he gain
calibration register contains a value which, when normalized, is
multiplied by all conversion results. T he offset calibration coeffi-
cient is subtracted from the result prior to the multiplication by
the gain coefficient.
simply performed a zero-scale calibration and updated the Off-
set Calibration Register for the selected channel. T he user must
write either 0, 0, 1 or 0, 1, 0 to the MD2, MD1, MD0 bits of the
Mode Register to initiate a conversion. If RDY is low before (or
goes low during) the calibration command write to the Mode
Register, it may take up to one modulator cycle (MCLK IN/32)
before RDY goes high to indicate that calibration is in progress.
T herefore, RDY should be ignored for up to one modulator
cycle after the last bit of the calibration command is written to
the Mode Register.
T he AD7730 offers self-calibration or system calibration facili-
ties. For full calibration to occur on the selected channel, the on-
chip microcontroller must record the modulator output for two
different input conditions. T hese are “zero-scale” and “full-
scale” points. T hese points are derived by performing a conver-
sion on the different input voltages provided to the input of the
modulator during calibration. T he result of the “zero-scale”
calibration conversion is stored in the Offset Calibration Regis-
ter for the appropriate channel. T he result of the “full-scale”
calibration conversion is stored in the Gain Calibration Register
for the appropriate channel. With these readings, the microcon-
troller can calculate the offset and the gain slope for the input to
output transfer function of the converter. Internally, the part
works with 33 bits of resolution to determine its conversion
result of either 16 bits or 24 bits.
For bipolar input ranges in the internal zero-scale calibrating
mode, the sequence is very similar to that just outlined. In this
case, the zero-scale point is exactly the same as above but since
the part is configured for bipolar operation, the output code for
zero differential input is 800000 Hex in 24-bit mode.
T he internal zero-scale calibration needs to be performed as
one part of a two part full calibration. However, once a full
calibration has been performed, additional internal zero-scale
calibrations can be performed by themselves to adjust the
part’s zero-scale point only. When performing a two step full
calibration care should be taken as to the sequence in which the
two steps are performed. If the internal zero-scale calibration is
one part of a full self-calibration, then it should take place after
an internal full-scale calibration. If it takes place in association
with a system full-scale calibration, then this internal zero-scale
calibration should be performed first.
T he sequence in which the zero-scale and full-scale calibration
occurs depends upon the type of full-scale calibration being
performed. T he internal full-scale calibration is a two-step cali-
bration that alters the value of the Offset Calibration Register.
T hus, the user must perform a zero-scale calibration (either
internal or system) after an internal full-scale calibration to correct
the Offset Calibration Register contents. When using system
full-scale calibration, it is recommended that the zero-scale
calibration (either internal or system) is performed first.
Inter nal Full-Scale Calibr ation
An internal full-scale calibration is initiated on the AD7730 by
writing the appropriate values (1, 0, 1) to the MD2, MD1 and
MD0 bits of the Mode Register. In this calibration mode, the
full-scale point used in determining the calibration coefficients is
with an internally-generated full-scale voltage. T his full-scale
voltage is derived from the reference voltage for the AD7730
and the PGA is set for the selected gain (as per the RN1, RN0
bits in the Mode Register) for this internal full-scale calibration
conversion.
Since the calibration coefficients are derived by performing a
conversion on the input voltage provided, the accuracy of the
calibration can only be as good as the noise level the part pro-
vides in normal mode. T o optimize the calibration accuracy, it
is recommended to calibrate the part at its lowest output rate
where the noise level is lowest. T he coefficients generated at any
output update rate will be valid for all selected output update
rates. T his scheme of calibrating at the lowest output update
rate does mean that the duration of calibration is longer.
In order to meet the post-calibration numbers quoted in the
specifications, it is recommended that internal full-scale calibra-
tions be performed on the 80 mV range. T his applies even if the
subsequent operating mode is on the 10 mV, 20 mV or 40 mV
input ranges.
Inter nal Zer o-Scale Calibr ation
T he internal full-scale calibration is a two-step sequence that
runs when an internal full-scale calibration command is written
to the AD7730. One part of the calibration is a zero-scale cali-
bration and as a result, the contents of the Offset Calibration
Register are altered during this Internal Full-Scale Calibration.
T he user must therefore perform a zero-scale calibration (either
internal or system) AFT ER the internal full-scale calibration.
T his zero-scale calibration should be performed at the operating input
range. T his means that internal full-scale calibrations cannot be
performed in isolation.
An internal zero-scale calibration is initiated on the AD7730 by
writing the appropriate values (1, 0, 0) to the MD2, MD1 and
MD0 bits of the Mode Register. In this calibration mode with a
unipolar input range, the zero-scale point used in determining
the calibration coefficients is with the inputs of the differential
pair internally shorted on the part (i.e., AIN(+) = AIN(–) =
Externally-Applied AIN(–) voltage). T he PGA is set for the
selected gain (as per the RN1, RN0 bits in the Mode Register)
for this internal zero-scale calibration conversion.
T he calibration is performed with dc excitation regardless of the
status of the ac bit. T he duration time of the calibration de-
pends upon the CHP bit of the Filter Register. With CHP = 1,
the duration is 22 × 1/Output Rate; with CHP = 0, the duration
is 24 × 1/Output Rate. At this time the MD2, MD1 and MD0
bits in the Mode Register return to 0, 0, 0 (Sync or Idle Mode
for the AD7730). T he RDY line goes high when calibration is
initiated and returns low when calibration is complete. Note
that the part has not performed a conversion at this time; it has
T he calibration is performed with dc excitation regardless of the
status of the ac bit. T he duration time of the calibration de-
pends upon the CHP bit of the Filter Register. With CHP = 1,
the duration is 44 × 1/Output Rate; with CHP = 0, the duration
is 48 × 1/Output Rate. At this time the MD2, MD1 and MD0
bits in the Mode Register return to 0, 0, 0 (Sync or Idle Mode
for the AD7730). T he RDY line goes high when calibration is
initiated and returns low when calibration is complete. Note
that the part has not performed a conversion at this time. T he
REV. A
–30–
AD7730/AD7730L
user must write either 0, 0, 1 or 0, 1, 0 to the MD2, MD1,
MD0 bits of the Mode Register to initiate a conversion. If
RDY is low before (or goes low during) the calibration com-
mand write to the Mode Register, it may take up to one modulator
cycle (MCLK IN/32) before RDY goes high to indicate that
calibration is in progress. T herefore, RDY should be ignored for
up to one modulator cycle after the last bit of the calibration
command is written to the Mode Register.
System Full-Scale Calibr ation
A system full-scale calibration is initiated on the AD7730 by
writing the appropriate values (1, 1, 1) to the MD2, MD1 and
MD0 bits of the Mode Register. System full-scale calibration is
performed using the system's positive full-scale voltage. T his
full-scale voltage must be set up before the calibration is initi-
ated, and it must remain stable throughout the calibration step.
T he system full-scale calibration is performed at the selected
gain (as per the RN1, RN0 bits in the Mode Register).
System Zer o-Scale Calibr ation
System calibration allows the AD7730 to compensate for system
gain and offset errors as well as its own internal errors. System
calibration performs the same slope factor calculations as self-
calibration, but uses voltage values presented by the system to
the AIN inputs for the zero- and full-scale points.
T he calibration is performed with either ac or dc excitation,
depending on the status of the ac bit. T he duration time of the
calibration depends upon the CHP bit of the Filter Register.
With CHP = 1, the duration is 22 × 1/Output Rate; with CHP =
0, the duration is 24 × 1/Output Rate. At this time the MD2,
MD1 and MD0 bits in the Mode Register return to 0, 0, 0
(Sync or Idle Mode for the AD7730). T he RDY line goes high
when calibration is initiated, and returns low when calibration is
complete. Note that the part has not performed a conversion at
this time; it has simply performed a full-scale calibration and
updated the Gain Calibration Register for the selected channel.
A system zero-scale calibration is initiated on the AD7730 by
writing the appropriate values (1, 1, 0) to the MD2, MD1 and
MD0 bits of the Mode Register. In this calibration mode, with a
unipolar input range, the zero-scale point used in determin-
ing the calibration coefficients is the bottom end of the trans-
fer function. T he system’s zero-scale point is applied to the
AD7730’s AIN input before the calibration step and this voltage
must remain stable for the duration of the system zero-scale
calibration. T he PGA is set for the selected gain (as per the
RN1, RN0 bits in the Mode Register) for this system zero-scale
calibration conversion. T he allowable range for the system zero-
scale voltage is discussed in the Span and Offsets Section.
T he user must write either 0, 0, 1 or 0, 1, 0 to the MD2, MD1,
MD0 bits of the Mode Register to initiate a conversion. If RDY
is low before (or goes low during) the calibration command
write to the Mode Register, it may take up to one modulator
cycle (MCLK IN/32) before RDY goes high to indicate that
calibration is in progress. T herefore, RDY should be ignored for
up to one modulator cycle after the last bit of the calibration
command is written to the Mode Register.
T he calibration is performed with either ac or dc excitation,
depending on the status of the AC bit. T he duration time of the
calibration depends upon the CHP bit of the Filter Register.
With CH P = 1, the duration is 22 × 1/Output Rate; with
CHP = 0, the duration is 24 × 1/Output Rate. At this time the
MD2, MD1 and MD0 bits in the Mode Register return to
0, 0, 0 (Sync or Idle Mode for the AD7730). T he RDY line
goes high when calibration is initiated and returns low when
calibration is complete. Note that the part has not performed a
conversion at this time; it has simply performed a zero-scale
calibration and updated the Offset Calibration Register for the
selected channel. T he user must write either 0, 0, 1 or 0, 1, 0 to
the MD2, MD1, MD0 bits of the Mode Register to initiate a
conversion. If RDY is low before (or goes low during) the cali-
bration command write to the Mode Register, it may take up to
one modulator cycle (MCLK IN/32) before RDY goes high to
indicate that calibration is in progress. T herefore, RDY should
be ignored for up to one modulator cycle after the last bit of the
calibration command is written to the Mode Register.
T he system full-scale calibration needs to be performed as one
part of a two part full calibration. Once a full calibration has
been performed, however, additional system full-scale calibra-
tions can be performed by themselves to adjust the part's gain
calibration point only. When performing a two-step full calibra-
tion care should be taken as to the sequence in which the two
steps are performed. A system full-scale calibration should not
be carried out unless the part contains valid zero-scale coeffi-
cients. T herefore, an internal zero-scale calibration or a system
zero-scale calibration must be performed before the system full-
scale calibration when a full two-step calibration operation is
being performed.
Span and O ffset Lim its
Whenever a system calibration mode is used, there are limits
on the amount of offset and span which can be accommodated.
T he overriding requirement in determining the amount of offset
and gain which can be accommodated by the part is the require-
For bipolar input ranges in the system zero-scale calibrating
mode, the sequence is very similar to that just outlined. In this
case, the zero-scale point is the midpoint of the AD7730’s
transfer function.
ment that the positive full-scale calibration limit is ≤1.05 × FS,
where FS is 10 mV, 20 mV, 40 mV or 80 mV depending on the
RN1, RN0 bits in the Mode Register. T his allows the input
range to go 5% above the nominal range. T he built-in head-
room in the AD7730’s analog modulator ensures that the part
will still operate correctly with a positive full-scale voltage that is
5% beyond the nominal.
T he system zero-scale calibration needs to be performed as one
part of a two part full calibration. However, once a full calibra-
tion has been performed, additional system zero-scale calibra-
tions can be performed by themselves to adjust the part’s
zero-scale point only. When performing a two-step full calibra-
tion care should be taken as to the sequence in which the two
steps are performed. If the system zero-scale calibration is one
part of a full system calibration, then it should take place before
a system full-scale calibration. If it takes place in association
with an internal full-scale calibration, then this system zero-scale
calibration should be performed after the full-scale calibration.
–31–
REV. A
AD7730/AD7730L
T he range of input span in both the unipolar and bipolar modes
has a minimum value of 0.8 × FS and a maximum value of
2.1 × FS. However, the span (which is the difference between
the bottom of the AD7730’s input range and the top of its input
range) has to take into account the limitation on the positive
full-scale voltage. T he amount of offset which can be accommo-
dated depends on whether the unipolar or bipolar mode is being
used. Once again, the offset has to take into account the limita-
tion on the positive full-scale voltage. In unipolar mode, there is
considerable flexibility in handling negative (with respect to
AIN(–)) offsets. In both unipolar and bipolar modes, the range
of positive offsets that can be handled by the part depends on
the selected span. T herefore, in determining the limits for sys-
tem zero-scale and full-scale calibrations, the user has to ensure
that the offset range plus the span range does exceed 1.05 × FS.
T his is best illustrated by looking at a few examples.
T he power dissipation and temperature drift of the AD7730 are
low and no warm-up time is required before the initial calibra-
tion is performed. If, however, an external reference is being
used, this reference must have stabilized before calibration is
initiated. Similarly, if the clock source for the part is generated
from a crystal or resonator across the MCLK pins, the start-up
time for the oscillator circuit should elapse before a calibration
is initiated on the part (see below).
D r ift Consider ations
T he AD7730 uses chopper stabilization techniques to minimize
input offset drift. Charge injection in the analog multiplexer and
dc leakage currents at the analog input are the primary sources
of offset voltage drift in the part. T he dc input leakage current is
essentially independent of the selected gain. Gain drift within
the converter depends primarily upon the temperature tracking
of the internal capacitors. It is not affected by leakage currents.
If the part is used in unipolar mode with a required span of
0.8 × FS, the offset range the system calibration can handle is
from –1.05 × FS to +0.25 × FS. If the part is used in unipolar
mode with a required span of FS, the offset range the system cali-
bration can handle is from –1.05 × FS to +0.05 × FS. Similarly, if
the part is used in unipolar mode and required to remove an
offset of 0.2 × FS, the span range the system calibration can
handle is 0.85 × FS.
When operating the part in CHOP mode (CHP = 1), the signal
chain including the first-stage filter is chopped. T his chopping
reduces the overall offset drift to 5 nV/°C. Integral and differen-
tial linearity errors are not significantly affected by temperature
changes.
Care must also be taken with external drift effects in order to
achieve optimum drift performance. T he user has to be espe-
cially careful to avoid, as much as possible, thermocouple effects
from junctions of different materials. Devices should not be
placed in sockets when evaluating temperature drift, there
should be no links in series with the analog inputs and care
must be taken as to how the input voltage is applied to the input
pins. T he true offset drift of the AD7730 itself can be evaluated
by performing temperature drift testing of the part with the
AIN(–)/AIN(–) input channel arrangement (i.e., internal shorted
input, test mode).
If the part is used in bipolar mode with a required span of
±0.4 × FS, the offset range the system calibration can handle is
from –0.65 × FS to +0.65 × FS. If the part is used in bipolar
mode with a required span of ±FS, the offset range the system
calibration can handle is from –0.05 × FS to +0.05 × FS. Simi-
larly, if the part is used in bipolar mode and required to remove
an offset of ±0.2 × FS, the span range the system calibration can
handle is ±0.85 × FS. Figure 16 summarizes the span and offset
ranges.
USING TH E AD 7730
Clocking and O scillator Cir cuit
1.05
؋
FS. UPPER LIMIT. AD7730’s INPUT
VOLTAGE CANNOT EXCEED THIS
The AD7730 requires a master clock input, which may be an
external CMOS compatible clock signal applied to the MCLK IN
pin with the MCLK OUT pin left unconnected. Alternatively, a
crystal or ceramic resonator of the correct frequency can be
connected between MCLK IN and MCLK OUT in which case
the clock circuit will function as an oscillator, providing the
clock source for the part. T he input sampling frequency, the
modulator sampling frequency, the –3 dB frequency, output
update rate and calibration time are all directly related to the
master clock frequency, fCLK IN. Reducing the master clock
frequency by a factor of two will halve the above frequencies and
update rate and double the calibration time.
GAIN CALIBRATIONS EXPAND OR
CONTRACT THE AD7730’s INPUT
RANGE
AD7730
INPUT RANGE
(0.8
؋
FS TO 2.1
؋
FS) 0V DIFFERENTIAL
NOMINAL ZERO-SCALE POINT
ZERO-SCALE CALIBRATIONS
MOVE INPUT RANGE UP OR DOWN
LOWER LIMIT. AD7730’s INPUT
VOLTAGE CANNOT EXCEED THIS
–1.05
؋
FS. Figure 16. Span and Offset Lim its
T he crystal or ceramic resonator is connected across the MCLK
IN and MCLK OUT pins, as per Figure 17. Capacitors C1 and
C2 may or may not be required and may vary in value depend-
ing on the crystal/resonator manufacturer's recommendations.
T he AD7730 has a capacitance of 5 pF on MCLK IN and 13 pF
on MCLK OUT so, in most cases, capacitors C1 and C2 will
not be required to get the crystal/resonator operating at its cor-
rect frequency.
P ower -Up and Calibr ation
On power-up, the AD7730 performs an internal reset which sets
the contents of the internal registers to a known state. T here are
default values loaded to all registers after a power-on or reset.
T he default values contain nominal calibration coefficients for
the calibration registers. T o ensure correct calibration for the
device, a calibration routine should be performed after power-up.
REV. A
–32–
AD7730/AD7730L
Reset Input
MCLK IN
T he RESET input on the AD7730 resets all the logic, the digital
filter and the analog modulator while all on-chip registers are
reset to their default state. RDY is driven high and the AD7730
ignores all communications to any of its registers while the
RESET input is low. When the RESET input returns high, the
AD7730 starts to process data and RDY will return low after
the filter has settled indicating a valid new word in the data
register. However, the AD7730 operates with its default setup
conditions after a RESET and it is generally necessary to set up all
registers and carry out a calibration after a RESET command.
C1
C2
CRYSTAL OR
CERAMIC
RESONATOR
AD7730
MCLK OUT
Figure 17. Crystal/Resonator Connections
T he on-chip oscillator circuit also has a start-up time associated
with it before it has attained its correct frequency and correct
voltage levels. T he typical start-up time for the circuit is 6 ms,
with a DVDD of +5 V and 8 ms with a DVDD of +3 V.
T he AD7730’s on-chip oscillator circuit continues to function
even when the RESET input is low. T he master clock signal
continues to be available on the MCLK OUT pin. T herefore, in
applications where the system clock is provided by the AD7730’s
clock, the AD7730 produces an uninterrupted master clock
during RESET commands.
T he AD7730’s master clock appears on the MCLK OUT pin of
the device. T he maximum recommended load on this pin is one
CMOS load. When using a crystal or ceramic resonator to gen-
erate the AD7730’s clock, it may be desirable to then use this
clock as the clock source for the system. In this case, it is recom-
mended that the MCLK OUT signal is buffered with a CMOS
buffer before being applied to the rest of the circuit.
Standby Mode
T he STANDBY input on the AD7730 allows the user to place
the part in a power-down mode when it is not required to pro-
vide conversion results. T he part can also be placed in its
standby mode by writing 0, 1, 1 to the MD2, MD1, MD0 bits
of the Mode Register. T he AD7730 retains the contents of all its
on-chip registers (including the Data Register) while in standby
mode. Data can still be read from the part in Standby Mode.
T he ST BY bit of the Status Register indicates whether the part
is in standby or normal operating mode. When the STANDBY
pin is taken high, the part returns to operating as it had been
prior to the STANDBY pin going low.
System Synchr onization
T he SYNC input allows the user to reset the modulator and
digital filter without affecting any of the setup conditions on the
part. T his allows the user to start gathering samples of the ana-
log input from a known point in time, i.e., the rising edge of
SYNC.
If multiple AD7730s are operated from a common master clock,
they can be synchronized to update their output registers simul-
taneously. A falling edge on the SYNC input resets the digital
filter and analog modulator and places the AD7730 into a con-
sistent, known state. While the SYNC input is low, the AD7730
will be maintained in this state. On the rising edge of SYNC,
the modulator and filter are taken out of this reset state and on
the next clock edge the part starts to gather input samples again.
In a system using multiple AD7730s, a common signal to their
SYNC inputs will synchronize their operation. T his would nor-
mally be done after each AD7730 has performed its own cali-
bration or has had calibration coefficients loaded to it. T he
output updates will then be synchronized with the maximum
possible difference between the output updates of the individual
AD7730s being one MCLK IN cycle.
T he STANDBY input (or 0, 1, 1 in the MD2, MD1, MD0 bits)
does not affect the digital interface. It does, however, set the
RDY bit and pin high and also sets the STDY bit high. When
STANDBY goes high again, RDY and STDY remain high until
set low by a conversion or calibration.
Placing the part in standby mode, reduces the total current to
10 µA typical when the part is operated from an external master
clock provided this master clock is stopped. If the external clock
continues to run in standby mode, the standby current increases
to 400 µA typical. If a crystal or ceramic resonator is used as the
clock source, then the total current in standby mode is 400 µA
typical. T his is because the on-chip oscillator circuit continues
to run when the part is in its standby mode. T his is important in
applications where the system clock is provided by the AD7730’s
clock, so that the AD7730 produces an uninterrupted master
clock even when it is in its standby mode.
Single-Shot Conver sions
T he SYNC input can also be used as a start convert command
allowing the AD7730 to be operated in a conventional converter
fashion. In this mode, the rising edge of SYNC starts conversion
and the falling edge of RDY indicates when conversion is com-
plete. T he disadvantage of this scheme is that the settling time
of the filter has to be taken into account for every data register
update.
D igital O utputs
T he AD7730 has two digital output pins, D0 and D1. When the
DEN bit of the Mode Register is set to 1, these digital outputs
assume the logic status of bits D0 and D1 of the Mode Register.
It gives the user access to two digital port pins which can be
programmed over the normal serial interface of the AD7730.
Writing 0, 1, 0 to the MD2, MD1, MD0 bits of the Mode regis-
ter has the same effect. T his initiates a single conversion on the
AD7730 with the part returning to idle mode at the end of
conversion. Once again, the full settling-time of the filter has to
elapse before the Data Register is updated.
T he two outputs obtain their supply voltage from AVDD
,
thus the outputs operate to 5 V levels even in cases where
DVDD = +3 V.
–33–
REV. A
AD7730/AD7730L
P O WER SUP P LIES
and analog signals. T races on opposite sides of the board should
run at right angles to each other. T his will reduce the effects of
feedthrough through the board. A microstrip technique is by far
the best but is not always possible with a double-sided board. In
this technique, the component side of the board is dedicated to
ground planes while signals are placed on the solder side.
T here is no specific power sequence required for the AD7730,
either the AVDD or the DVDD supply can come up first. While
the latch-up performance of the AD7730 is very good, it is
important that power is applied to the AD7730 before signals at
REF IN, AIN or the logic input pins in order to avoid latch-up
caused by excessive current. If this is not possible, the current
that flows in any of these pins should be limited to less than 30
mA per pin and less than 100 mA cumulative. If separate sup-
plies are used for the AD7730 and the system digital circuitry,
the AD7730 should be powered up first. If it is not possible to
guarantee this, current limiting resistors should be placed in
series with the logic inputs to again limit the current to less than
30 mA per pin and less than 100 mA total.
Good decoupling is important when using high resolution
ADCs. All analog supplies should be decoupled with 10 µF
tantalum in parallel with 0.1 µF ceramic capacitors to AGND.
T o achieve the best from these decoupling components, they
have to be placed as close as possible to the device, ideally right
up against the device. All logic chips should be decoupled with
0.1 µF disc ceramic capacitors to DGND. In systems where a
common supply voltage is used to drive both the AVDD and
DVDD of the AD7730, it is recommended that the system’s
AVDD supply is used. T his supply should have the recom-
mended analog supply decoupling capacitors between the AVDD
pin of the AD7730 and AGND and the recommended digital
supply decoupling capacitor between the DVDD pin of the
AD7730 and DGND.
Gr ounding and Layout
Since the analog inputs and reference input are differential,
most of the voltages in the analog modulator are common-mode
voltages. T he excellent common-mode rejection of the part will
remove common-mode noise on these inputs. T he analog and
digital supplies to the AD7730 are independent and separately
pinned out to minimize coupling between the analog and digital
sections of the device. T he digital filter will provide rejection of
broadband noise on the power supplies, except at integer mul-
tiples of the modulator sampling frequency or multiples of the
chop frequency in chop mode. T he digital filter also removes
noise from the analog and reference inputs provided those noise
sources do not saturate the analog modulator. As a result, the
AD7730 is more immune to noise interference than a conven-
tional high resolution converter. However, because the resolu-
tion of the AD7730 is so high and the noise levels from the
AD7730 so low, care must be taken with regard to grounding
and layout.
Evaluating the AD 7730 P er for m ance
A recommended layout for the AD7730 is outlined in the evalu-
ation board for the AD7730. T he evaluation board package
includes a fully assembled and tested evaluation board, docu-
mentation, software for controlling the board over the printer
port of a PC and software for analyzing the AD7730’s perfor-
mance on the PC . T he evaluation board order number is
EVAL-AD7730EB.
Noise levels in the signals applied to the AD7730 may also
affect performance of the part. T he AD7730 allows two tech-
niques for evaluating the true performance of the part, indepen-
dent of the analog input signal. T hese schemes should be used
after a calibration has been performed on the part.
T he printed circuit board that houses the AD7730 should be
designed so the analog and digital sections are separated and
confined to certain areas of the board. T his facilitates the use of
ground planes that can be easily separated. A minimum etch
technique is generally best for ground planes as it gives the best
shielding. Digital and analog ground planes should only be
joined in one place. If the AD7730 is the only device requiring
an AGND to DGND connection, the ground planes should
be connected at the AGND and DGND pins of the AD7730. If
the AD7730 is in a system where multiple devices require AGND
to DGND connections, the connection should still be made at
one point only, a star ground point that should be established as
closely as possible to the AD7730.
T he first method is to select the AIN1(–)/AIN1(–) input chan-
nel arrangement. In this case, the differential inputs to the
AD7730 are internally shorted together to provide a zero differ-
ential voltage for the analog modulator. External to the device,
the AIN1(–) input should be connected to a voltage which is
within the allowable common-mode range of the part.
T he second scheme is to evaluate the part with a voltage near
input full scale. T his can be achieved by again using input pair
AIN1(–), but by adding a differential voltage via the T ARE
DAC. T his allows the user to evaluate noise performance with a
near full-scale voltage.
Avoid running digital lines under the device as these will couple
noise onto the die. T he analog ground plane should be allowed
to run under the AD7730 to avoid noise coupling. T he power
supply lines to the AD7730 should use as large a trace as pos-
sible to provide low impedance paths and reduce the effects of
glitches on the power supply line. Fast switching signals such as
clocks should be shielded with digital ground to avoid radiating
noise to other sections of the board and clock signals should
never be run near the analog inputs. Avoid crossover of digital
T he software in the evaluation board package allows the user to
look at the noise performance in terms of counts, bits and nV.
Once the user has established that the noise performance of the
part is satisfactory in this mode, an external input voltage can
then be applied to the device incorporating more of the signal
chain.
REV. A
–34–
AD7730/AD7730L
SERIAL INTERFACE
data transfer when the POL input is high and should idle low
between data transfers when the POL input is low. For POL = 1,
the first falling edge of SCLK clocks data from the microcontrol-
ler onto the DIN line of the AD7730. It is then clocked into the
input shift register on the next rising edge of SCLK. For POL = 0,
the first clock edge that clocks data from the microcontroller
onto the DIN line of the AD7730 is a rising edge. It is then
clocked into the input shift register on the next falling edge of
SCLK.
T he AD7730’s programmable functions are controlled via a set
of on-chip registers. Access to these registers is via the part’s
serial interface. After power-on or RESET, the device expects a
write to its Communications Register. T he data written to this
register determines whether the next operation to the part is a
read or a write operation and also determines to which register
this read or write operation occurs. T herefore, write access to
one of the control registers on the part starts with a write opera-
tion to the Communications Register followed by a write to the
selected register. Reading from the part’s on-chip registers can
take the form of either a single or continuous read. A single read
from a register consists of a write to the Communications Regis-
ter (with RW1 = 0 and RW0 = 1) followed by the read from the
specified register. T o perform continuous reads from a register,
write to the Communications Register (with RW1 = 1 and
RW0 = 0) to place the part in continuous read mode. T he speci-
fied register can then be read from continuously until a write
operation to the Communications Register (with RW1 = 1 and
RW0 = 1) which takes the part out of continuous read mode.
When operating in continuous read mode, the part is continu-
ously monitoring its DIN line. T he DIN line should therefore
be permanently low to allow the part to stay in continuous read
mode. Figure 5 and Figure 6, shown previously, indicate the
correct flow diagrams when reading and writing from the
AD7730’s registers.
In other microcontroller applications which require a decoding
of the AD7730, CS can be generated from a port line. In this
case, CS would go low well in advance of the first falling edge of
SCLK (POL = 1) or the first rising edge of SCLK (POL = 0).
Clocking of each bit of data is as just described.
In DSP applications, the SCLK is generally a continuous clock.
In these applications, the CS input for the AD7730 is generated
from a frame synchronization signal from the DSP. For proces-
sors with the rising edge of SCLK as the active edge, the POL
input should be tied high. For processors with the falling edge of
SCLK as the active edge, the POL input should be tied low. In
these applications, the first edge after CS goes low is the active
edge. T he MSB of the data to be shifted into the AD7730 must
be set up prior to this first active edge.
Read O per ation
T he reading of data from the part is from an output shift regis-
ter. On initiation of a read operation, data is transferred from
the specified register to the output shift register. T his is a paral-
lel shift and is transparent to the user. Figure 19 shows a timing
diagram for a read operation from the output shift register of the
AD7730. With the POL input at a logic high, the data is clocked
out of the output shift register on the falling edge of SCLK.
With the POL input at a logic low, the data is clocked out of the
output shift register on the rising edge of SCLK.
T he AD7730’s serial interface consists of five signals, CS,
SCLK, DIN, DOUT and RDY. T he DIN line is used for
transferring data into the on-chip registers while the DOUT line
is used for accessing data from the on-chip registers. SCLK is
the serial clock input for the device and all data transfers (either
on DIN or DOUT ) take place with respect to this SCLK signal.
Wr ite O per ation
T he transfer of data into the part is to an input shift register. On
completion of a write operation, data is transferred to the speci-
fied register. T his internal transfer will not take place until the
correct number of bits for the specified register have been
loaded to the input shift register. For example, the transfer of
data from the input shift register takes place after eight serial
clock cycles for a DAC Register write, while the transfer of data
from the input shift register takes place after 24 serial clock
cycles when writing to the Filter Register. Figure 18 shows a
timing diagram for a write operation to the input shift register of
the AD7730. With the POL input at a logic high, the data is
latched into the input shift register on the rising edge of SCLK.
With the POL input at a logic low, the data is latched into the
input shift register on the falling edge of SCLK.
Figure 19 also shows the CS input being used to decode the
read operation to the AD7730. However, this CS input can be
used in a number of different ways. It is possible to operate the
part in three-wire mode where the CS input is permanently tied
low. In this case, the SCLK line should idle high between data
transfer when the POL input is high, and should idle low be-
tween data transfers when the POL input is low. For POL = 1,
the first falling edge of SCLK clocks data from the output shift
register onto the DOUT line of the AD7730. It is then clocked
into the microcontroller on the next rising edge of SCLK. For
POL = 0, the first clock edge that clocks data from the AD7730
onto the DOUT line is a rising edge. It is then clocked into the
microcontroller on the next falling edge of SCLK.
In other microcontroller applications which require a decoding
of the AD7730, CS can be generated from a port line. In this
case, CS would go low well in advance of the first falling edge of
SCLK (POL = 1) or the first rising edge of SCLK (POL = 0).
Clocking of each bit of data is as just described.
Figure 18 also shows the CS input being used to decode the
write operation to the AD7730. However, this CS input can be
used in a number of different ways. It is possible to operate the
part in three-wire mode where the CS input is tied low perma-
nently. In this case, the SCLK line should idle high between
–35–
REV. A
AD7730/AD7730L
In DSP applications, the SCLK is generally a continuous clock.
In these applications, the CS input for the AD7730 is generated
from a frame synchronization signal from the DSP. In these
applications, the first edge after CS goes low is the active edge.
T he MSB of the data to be shifted into the DSP must be set up
prior to this first active edge. Unlike microcontroller applica-
tions, the DSP does not provide a clock edge to clock the MSB
from the AD7730. In this case, the CS of the AD7730 places
the MSB on the DOUT line. For processors with the rising edge
of SCLK as the active edge, the POL input should be tied high.
In this case, the DSP takes data on the rising edge. If CS goes
low while SCLK is low, the MSB is clocked out on the DOUT
line from the CS. Subsequent data bits are clocked from the
falling edge of SCLK. For processors with the falling edge of
SCLK as the active edge, the POL input should be tied low. In
this case, the DSP takes data on the falling edge. If CS goes low
while SCLK is high, the MSB is clocked out on the DOUT line
from the CS. Subsequent data bits are clocked from the rising
edge of SCLK.
T he RDY line is used as a status signal to indicate when data is
ready to be read from the AD7730’s data register. RDY goes
low when a new data word is available in the data register. It is
reset high when a read operation from the data register is com-
plete. It also goes high prior to the updating of the data register
to indicate when a read from the data register should not be
initiated. T his is to ensure that the transfer of data from the data
register to the output shift register does not occur while the data
register is being updated. It is possible to read the same data
twice from the output register even though the RDY line returns
high after the first read operation. Care must be taken, however,
to ensure that the read operations are not initiated as the next
output update is about to take place.
For systems with a single data line, the DIN and DOUT lines
on the AD7730 can be connected together, but care must be
taken in this case not to place the part in continuous read mode
as the part monitors DIN while supplying data on DOUT and
as a result, it may not be possible to take the part out of its
continuous read mode.
RDY
t10
t3
CS
t8
t4
t6
SCLK
(POL = 1)
t7
t6
SCLK
(POL = 0)
t7
t5
t9
t5A
DOUT
MSB
LSB
Figure 18. Read Cycle Tim ing Diagram
CS
t16
t11
t14
SCLK
(POL = 1)
t15
t14
SCLK
(POL = 0)
t15
t12
t13
MSB
LSB
DIN
Figure 19. Write Cycle Tim ing Diagram
REV. A
–36–
AD7730/AD7730L
CO NFIGURING TH E AD 7730
T he AD7730 contains twelve on-chip registers that can be accessed via the serial interface. Figure 5 and Figure 6 have outlined a
flowchart for the reading and writing of these registers. T able XIX and T able XX outline sample pseudo-code for some commonly
used routines. T he required operating conditions will dictate the values loaded to the Mode, Filter and DAC Registers. T he values
given here are for example purposes only.
Table XIX. P seudo-Code for Initiating a Self-Calibration after P ower-O n/Reset
Write 03 Hex to Serial Port1
/* Writes to Communications Register Setting Next Operation as Write to
Filter Register*/
Write 800010 Hex to Serial Port1
Write 04 Hex to Serial Port1
/* Writes to Filter Register Setting a 50 Hz Output Rate in CHOP Mode*/
/* Writes to Communications Register Setting Next Operation as Write to
DAC Register*/
Write 23 Hex to Serial Port1
Write 02 Hex to Serial Port
Write B180 Hex to Serial Port
/* Writes to DAC Register Setting a Subtraction Value of 7.5 mV (5 V Refer-
ence) on the T ARE DAC*/
/* Writes to Communications Register Setting Next Operation as Write to
Mode Register*/
/* Writes to Mode Register Initiating Internal Full-Scale Calibration for 0 mV
to +10 mV Input Range*/
Wait for RDY Low
/* Wait for RDY pin to go low to indicate end of calibration cycle*/
Write 02 Hex to Serial Port
/* Writes to Communications Register Setting Next Operation as Write to
Mode Register*/
Write 9180 Hex to Serial Port
Wait for RDY Low
/* Writes to Mode Register Initiating Internal Zero-Scale Calibration for
0 mV to +10 mV Input Range*/
/* Wait for RDY pin to go low to indicate end of calibration cycle*/
/* T he part has now completed self-calibration and is in idle mode*/
1T his operation is not necessary if the default values of the Filter Register or the DAC Register are the values used in the application.
Table XX. P seudo-Code for Setting Up AD 7730 for Continuous Conversion and Continuous Read O peration
Write 02 Hex to Serial Port
Write 2180 Hex to Serial Port
Write 21 Hex to Serial Port
/* Writes to Communications Register Setting Next Operation as Write to
Mode Register*/
/* Writes to Mode Register Starting Continuous Conversions for 0 mV to
+10 mV Input Range*/
/* Writes to Communications Register Setting Next Operation as Continuous
Read From Data Register*/
Set DIN Line of AD7730 Low
/* Ensures Part is not Reset While in Continuous Read Mode*/
/* Wait for RDY pin to go low to Indicate Output Update*/
/* Read Conversion Result from AD7730's Data Register*/
READ_DAT A: Wait for RDY Low
Read 24-Bit Data From Serial Port
Loop to READ_DAT A Until All Data Gathered
Write 30 Hex to Serial Port
/* Ends Continuous Read Operation and Places Part in Mode Where It
Expects Write to Communications Register*/
–37–
REV. A
AD7730/AD7730L
MICRO CO MP UTER/MICRO P RO CESSO R INTERFACING
T he AD7730’s flexible serial interface allows for easy interface
to most microcomputers and microprocessors. T he pseudo-code
of T able XIX and T able XX outline typical sequences for inter-
facing a microcontroller or microprocessor to the AD7730.
Figures 20, 21 and 22 show some typical interface circuits.
T he 68HC11 is configured in the master mode with its CPOL
bit set to a logic zero and its CPHA bit set to a logic one. When
the 68H C11 is configured like this, its SCLK line idles low
between data transfers. Therefore, the POL input of the AD7730
should be hardwired low. For systems where it is preferable that
the SCLK idle high, the CPOL bit of the 68HC11 should be set
to a Logic 1 and the POL input of the AD7730 should be hard-
wired to a logic high.
T he serial interface on the AD7730 has the capability of operat-
ing from just three wires and is compatible with SPI interface
protocols. T he three-wire operation makes the part ideal for
isolated systems where minimizing the number of interface lines
minimizes the number of opto-isolators required in the system.
T he AD7730 is not capable of full duplex operation. If the
AD7730 is configured for a write operation, no data appears on
the DAT A OUT lines even when the SCLK input is active.
When the AD7730 is configured for continuous read operation,
data presented to the part on the DAT A IN line is monitored to
determine when to exit the continuous read mode.
Register lengths on the AD7730 vary from 8 to 16 to 24 bits.
T he 8-bit serial ports of most microcontrollers can handle
communication with these registers as either one, two or three
8-bit transfers. DSP processors and microprocessors generally
transfer 16 bits of data in a serial data operation. Some of these
processors, such as the ADSP-2105, have the facility to program
the amount of cycles in a serial transfer. T his allows the user to
tailor the number of bits in any transfer to match the register
length of the required register in the AD7730. In any case,
writing 32 bits of data to a 24-bit register is not an issue provided
the final eight bits of the word are all 1s. T his is because the
part returns to the Communications Register following a write
operation.
DV
DV
DD
DD
SS
SYNC
RESET
68HC11
AD7730
SCLK
SCK
MISO
MOSI
DATA OUT
DATA IN
CS
Even though some of the registers on the AD7730 are only eight
bits in length, communicating with two of these registers in
successive write operations can be handled as a single 16-bit
data transfer if required. For example, if the DAC Register is to
be updated, the processor must first write to the Communica-
tions Register (saying that the next operation is a write to the
Mode Register) and then write eight bits to the DAC Register.
T his can all be done in a single 16-bit transfer, if required, be-
cause once the eight serial clocks of the write operation to the
Communications Register have been completed, the part imme-
diately sets itself up for a write operation to the DAC Register.
POL
Figure 20. AD7730 to 68HC11 Interface
AD 7730 to 8051 Inter face
An interface circuit between the AD7730 and the 8XC51 mi-
crocontroller is shown in Figure 21. T he diagram shows the
minimum number of interface connections with CS on the
AD7730 hardwired low. In the case of the 8XC51 interface, the
minimum number of interconnects is just two. In this scheme,
the RDY bit of the Status Register is monitored to determine
when the Data Register is updated. T he alternative scheme,
which increases the number of interface lines to three, is to
monitor the RDY output line from the AD7730. T he monitor-
ing of the RDY line can be done in two ways. First, RDY can be
connected to one of the 8XC51’s port bits (such as P1.0), which
is configured as an input. T his port bit is then polled to deter-
mine the status of RDY. T he second scheme is to use an inter-
rupt driven system, in which case the RDY output is connected
to the INT1 input of the 8XC51. For interfaces that require
control of the CS input on the AD7730, one of the port bits of
the 8XC51 (such as P1.1), which is configured as an output,
can be used to drive the CS input.
AD 7730 to 68H C11 Inter face
Figure 20 shows an interface between the AD7730 and the
68HC11 microcontroller. T he diagram shows the minimum
(three-wire) interface with CS on the AD7730 hardwired low.
In this scheme, the RDY bit of the Status Register is monitored
to determine when the Data Register is updated. An alternative
scheme, which increases the number of interface lines to four, is
to monitor the RDY output line from the AD7730. T he moni-
toring of the RDY line can be done in two ways. First, RDY can
be connected to one of the 68HC11’s port bits (such as PC0),
which is configured as an input. T his port bit is then polled to
determine the status of RDY. T he second scheme is to use an
interrupt driven system, in which case the RDY output is con-
nected to the IRQ input of the 68HC11. For interfaces which
require control of the CS input on the AD7730, one of the port
bits of the 68HC11 (such as PC1), which is configured as an
output, can be used to drive the CS input.
T he 8XC51 is configured in its Mode 0 serial interface mode.
Its serial interface contains a single data line. As a result, the
DAT A OUT and DAT A IN pins of the AD7730 should be
connected together. T his means that the AD7730 must not be
REV. A
–38–
AD7730/AD7730L
configured for continuous read operation when interfacing to
the 8XC51. The serial clock on the 8XC51 idles high between data
transfers and therefore the POL input of the AD7730 should be
hardwired to a logic high. T he 8XC51 outputs the LSB first in a
write operation while the AD7730 expects the MSB first so the
data to be transmitted has to be rearranged before being written
to the output serial register. Similarly, the AD7730 outputs the
MSB first during a read operation while the 8XC51 expects the
LSB first. Therefore, the data read into the serial buffer needs to
be rearranged before the correct data word from the AD7730 is
available in the accumulator.
AD 7730 to AD SP -2103/AD SP -2105 Inter face
Figure 22 shows an interface between the AD7730 and the
ADSP-2105 DSP processor. In the interface shown, the RDY
bit of the Status Register is again monitored to determine when
the Data Register is updated. T he alternative scheme is to use
an interrupt driven system, in which case the RDY output is
connected to the IRQ2 input of the ADSP-2105. T he RFS and
TFS pins of the ADSP-2105 are configured as active low out-
puts and the ADSP-2105 serial clock line, SCLK, is also config-
ured as an output. T he POL pin of the AD7730 is hardwired
low. Because the SCLK from the ADSP-2105 is a continuous
clock, the CS of the AD7730 must be used to gate off the clock
once the transfer is complete. T he CS for the AD7730 is active
when either the RFS or TFS outputs from the ADSP-2105 are
active. T he serial clock rate on the ADSP-2105 should be lim-
ited to 3 MHz to ensure correct operation with the AD7730.
DV
DD
SYNC
RESET
POL
8XC51
AD7730
DV
DD
P3.0
P3.1
DATA OUT
DATA IN
SYNC
RESET
SCLK
ADSP-2105
AD7730
RFS
TFS
CS
CS
DR
DT
DATA OUT
DATA IN
SCLK
Figure 21. AD7730 to 8XC51 Interface
SCLK
POL
Figure 22. AD7730 to ADSP-2105 Interface
–39–
REV. A
AD7730/AD7730L
AP P LICATIO NS
can be used with the excitation voltage and analog ground con-
nected local to the AD7730’s REF IN(+) and REF IN(–) termi-
nals. Illustrating a major advantage of the AD7730, the 5 V
excitation voltage for the bridge can be used directly as the refer-
ence voltage for the AD7730, eliminating the need for precision
matched resistors in generating a scaled-down reference.
T he on-chip PGA allows the AD7730 to handle analog input
voltage ranges as low as 10 mV full scale. T his allows the user to
connect a transducer directly to the input of the AD7730. T he
AD7730 is primarily targeted for weigh-scale and load-cell
applications. T he majority of the applications have a strain-
gage transducer whose resistance changes when subjected to
mechanical stress. Normally, the gages are configured in a
Wheatstone bridge arrangement. T he strain gage is a passive
device and requires an excitation voltage (or in some cases a
current) to derive a voltage output. T wo types of voltage excita-
tion can be provided for the bridge: dc excitation or ac excita-
tion. T hese are discussed in the following sections. While the
desire in most applications is to provide a single supply solution
(something that is aided by the AD7730’s single supply capabil-
ity), some applications provide a bipolar excitation voltage in
order to increase the output voltage from the bridge. In such
cases, the input voltage applied to the AD7730 can be slightly
negative with respect to ground. Figure 23 shows how to config-
ure the AD7730 to handle this type of input signal.
T he application is a ratiometric one with variations in the exci-
tation voltage being reflected in variations in the analog input
voltage and reference voltage of the AD7730. Because the
AD7730 is a truly ratiometric part, with the reference voltage
and excitation voltages equal, it is possible to evaluate its total
excitation voltage rejection. T his is unlike other converters
which give a separate indication of the rejection of reference,
analog inputs and power supply. T he combined (total) rejection
for the AD7730 when moving the excitation voltage (which was
also the power supply voltage) was better than 115 dB when
evaluated with a load cell simulator.
Drift considerations are a primary concern for load cell applica-
tions. It is recommended for these applications that the AD7730
is operated in CHOP mode to accrue the benefits of the excel-
lent drift performance of the part in CHOP mode. A common
source of unwanted drift effects are parasitic thermocouples.
T hermocouple effects are generated every time there is a junc-
tion of two dissimilar metals. All components in the signal path
should be chosen to minimize thermocouple effects. IC sockets
and link options should be avoided as much as possible. While
it is impossible to remove all thermocouple effects, attempts should
be made to equalize the thermocouples on each leg of the differen-
tial input to minimize the differential voltage generated.
D C Excitation of Br idge
In dc-excitation applications, the excitation voltage provided for
the bridge is a fixed dc voltage. Connections between the AD7730
and the bridge are very straightforward in this type of applica-
tion as illustrated in Figure 23. T he bridge configuration shown
is a six-lead configuration with separate return leads for the
reference lines. T his allows a force/sense effect on the load cell
excitation voltage, eliminating voltage drops caused by the exci-
tation current flowing through the lead resistances. In applica-
tions where the lead lengths are short, a four-wire configuration
EXCITATION VOLTAGE = +5V
DV
AV
DD
DD
REF IN(+)
REF IN(–)
AD7730
IN+
IN–
SIGMA-DELTA A/D CONVERTER
AIN1(+)
AIN1(–)
STANDBY
BUFFER
SIGMA-
DELTA
MODULATOR
PROGRAMMABLE
DIGITAL
SYNC
OUT–
OUT+
+
FILTER
MUX
PGA
+/–
MCLK IN
AIN2(+)/D1
AIN2(–)/D0
CLOCK
6-BIT
DAC
GENERATION
SERIAL INTERFACE
AND CONTROL LOGIC
MCLK OUT
REGISTER BANK
SCLK
CS
CALIBRATION
MICROCONTROLLER
DIN
ACX
AC
DOUT
EXCITATION
CLOCK
ACX
AGND
DGND
POL
RDY
RESET
Figure 23. Typical Connections for DC-Excited Bridge Application
REV. A
–40–
AD7730/AD7730L
Long lead lengths from the bridge to the AD7730 facilitate the
pickup of mains frequency on the analog input, the reference
input and the power supply. T he analog inputs to the AD7730
are buffered, which allows the user to connect whatever noise
reduction capacitors are necessary in the application. The AD7730
boasts excellent common-mode and normal- mode rejection of
mains frequency on both the analog and reference inputs. In
CHOP mode, care must be taken in choosing the output update
rate so it does not result in reducing line frequency rejection
(see DIGIT AL FILT ERING section). T he input offset current
on the AD7730 is 10 nA maximum which results in a maxi-
mum, dc offset voltage of 1.75 mV in a 350 Ω bridge applica-
tion. Care should taken with inserting large source impedances
on the reference input pins as these inputs are not buffered and
the source impedances can result in gain errors.
discrete matched bipolar or MOS transistors, or a dedicated
bridge driver chip such as the 4427 from Micrel can be used to
perform the task.
Since the analog input voltage and the reference voltage are
reversed on alternate cycles, the AD7730 must be synchronized
with this reversing of the excitation voltage. T o allow the
AD7730 to synchronize itself with this switching, it provides the
logic control signals for the switching of the excitation voltage.
T hese signals are the nonoverlapping CMOS outputs ACX
and ACX.
One of the problems encountered with ac-excitation is the set-
tling time associated with the analog input signals after the
excitation voltage is switched. T his is particularly true in appli-
cations where there are long lead lengths from the bridge to the
AD7730. It means that the converter could encounter errors
because it is processing signals which are not fully settled. T he
AD7730 addresses this problem by allowing the user to program
a delay of up to 48.75 µs between the switching of the ACX
signals and the processing of data at the analog inputs. T his is
achieved using the DL bits of the Filter Register.
In many load-cell applications, a portion of the dynamic range
of the bridge output is consumed by a pan weight or tare weight.
In such applications, the 6-bit T ARE DAC of the AD7730 can
be used to adjust out this tare weight as outlined previously.
AC Excitation of Br idge
AC excitation of the bridge addresses many of the concerns with
thermocouple, offset and drift effects encountered in dc-excited
applications. In ac-excitation, the polarity of the excitation volt-
age to the bridge is reversed on alternate cycles. T he result is the
elimination of dc errors at the expense of a more complex sys-
tem design. Figure 24 outlines the connections for an ac-excited
bridge application based on the AD7730.
T he AD7730 also scales the ACX switching frequency in accor-
dance with the output update rate. T his avoids situations where
the bridge is switched at an unnecessarily faster rate than the
system requires.
T he fact that the AD7730 can handle reference voltages which
are the same as the excitation voltages is particularly useful in
ac-excitation where resistor divider arrangements on the
reference input add to the settling time associated with the
switching.
The excitation voltage to the bridge must be switched on
alternate cycles. T ransistors T 1 to T 4 in Figure 24 perform
the switching of the excitation voltage. T hese transistors can be
EXCITATION VOLTAGE = +5V
DV
DD
AV
DD
T1
T2
REF IN(+)
REF IN(–)
AD7730
IN+
IN–
SIGMA-DELTA A/D CONVERTER
AIN1(+)
AIN1(–)
STANDBY
BUFFER
SIGMA-
DELTA
MODULATOR
PROGRAMMABLE
DIGITAL
SYNC
OUT–
OUT+
+
FILTER
MUX
PGA
+/–
MCLK IN
AIN2(+)/D1
AIN2(–)/D0
CLOCK
6-BIT
DAC
GENERATION
SERIAL INTERFACE
AND CONTROL LOGIC
MCLK OUT
T3
T4
REGISTER BANK
SCLK
CS
CALIBRATION
MICROCONTROLLER
DIN
ACX
AC
DOUT
EXCITATION
CLOCK
ACX
AGND
DGND
POL
RDY
RESET
Figure 24. Typical Connections for AC-Excited Bridge Application
–41–
REV. A
AD7730/AD7730L
Bipolar Excitation of the Br idge
a minimum of 1.2 V. T he 10 V excitation voltage must be re-
duced to 5 V before being applied as the reference voltage for
the AD7730.
As mentioned previously, some applications will require that the
AD7730 handle inputs from a bridge that is excited by a bipolar
voltage. T he number of applications requiring this are limited,
but with the addition of some external components the AD7730
is capable of handling such signals. Figure 25 outlines one ap-
proach to the problem.
T he resistor string R1, R2 and R3, takes the 10 V excitation
voltage and generates differential voltage of nominally 5 V.
Amplifiers A1 and A2 buffer the resistor string voltages and
provide the AVDD and AGND voltages as well as the REF IN(+)
and REF IN(–) voltages for the AD7730. T he differential
reference voltage for the part is +5 V. T he AD7730 retains its
ratiometric operation with this reference voltage varying in sym-
pathy with the analog input voltage.
T he example shown is a dc-excited bridge that is driven from
±5 V supplies. In such a circuit, two issues must be addressed.
T he first is how to get the AD7730 to handle input voltages
near or below ground and the second is how to take the 10 V
excitation voltage which appears across the bridge and generate
a suitable reference voltage for the AD7730. T he circuit of Figure
25 attempts to address these two issues simultaneously.
T he values of the resistors in the resistor string can be changed
to allow a larger DVDD voltage. For example, if R1 = 3 kΩ,
R2 = 10 kΩ and R3 = 7 kΩ, the AVDD and AGND voltages
become +3.5 V and –1.5 V respectively. This allows the AD7730
to be used with a +3.6 V DVDD voltage while still allowing the
analog input range to be within the specified common-mode
range.
T he AD7730’s analog and digital supplies can be split such that
AVDD and DVDD can be at separate potentials and AGND and
DGND can also be at separate potentials. T he only stipulation
is that AVDD or DVDD must not exceed the AGND by 5.5 V.
In Figure 25, the DVDD is operated at +3 V, which allows the
AGND to go down to –2.5 V with respect to system ground.
T his means that all logic signals to the part must not exceed 3 V
with respect to system ground. T he AVDD is operated at +2.5 V
with respect to system ground.
An alternate scheme to this is to generate the AVDD and AGND
voltages from regulators or Zener diodes driven from the +5 V
and –5 V supplies respectively. T he reference voltage for the
part would be generated in the same manner as just outlined but
amplifiers A1 and A2 would not be required to buffer the volt-
ages as they are now only driving the reference pins of the
AD7730. However, care must be taken in this scheme to ensure
that the REF IN(+) voltage does not exceed AVDD and that the
REF IN(–) voltage does not go below AGND.
T he bridge is excited with 10 V across its inputs. T he output of
the bridge is biased around the midpoint of the excitation volt-
ages which in this case is system ground or 0 V. In order for the
common-mode voltage of the analog inputs to sit correctly, the
AGND of the AD7730 must be biased below system ground by
+3V
DV
DD
1/2 OP284
OR 1/2 OP213
AV
+5V
DD
+5V
AD7730
R1
5k
REF IN(+)
A1
SIGMA-DELTA A/D CONVERTER
–5V
IN+
IN–
STANDBY
SYNC
AIN1(+)
AIN1(–)
BUFFER
SIGMA-
DELTA
MODULATOR
PROGRAMMABLE
DIGITAL
R2
10k
OUT–
+
FILTER
OUT+
MUX
PGA
+/–
+5V
–5V
MCLK IN
REF IN(–)
AGND
CLOCK
A2
6-BIT
DAC
R3
5k
GENERATION
SERIAL INTERFACE
AND CONTROL LOGIC
MCLK OUT
REGISTER BANK
1/2 OP284
OR 1/2 OP213
–5V
SCLK
CS
CALIBRATION
MICROCONTROLLER
DIN
DOUT
DGND
POL
RDY
RESET
ALL VOLTAGE VALUES ARE WITH
RESPECT TO SYSTEM GROUND.
SYSTEM
GROUND
Figure 25. AD7730 with Bipolar Excitation of the Bridge
REV. A
–42–
AP P END IX A
AD 7730L SP ECIFICATIO NS
–43–
a
APPENDIX–AD7730L*
GENERAL D ESCRIP TIO N
LOW POWER
T he AD7730L is a complete low power analog front-end for
weigh-scale and pressure measurement applications. T he device
accepts low level signals directly from a transducer and outputs
a serial digital word. T he input signal is applied to a proprietary
programmable gain front end based around an analog modula-
tor. T he modulator output is processed by a low pass program-
mable digital filter, allowing adjustment of filter cutoff, output
rate and settling-time.
BRIDGE TRANSDUCER ADC
KEY FEATURES
Resolution of 110,000 Counts (Peak-to-Peak)
Pow er Consum ption: 15 m W typ
Offset Drift: < 1 ppm / ؇C
Gain Drift: 3 ppm / ؇C
Line Frequency Rejection: >150 dB
Buffered Differential Inputs
Program m able Filter Cutoffs
Specified for Drift Over Tim e
Operates w ith Reference Voltages of 1 V to 5 V
T he part features two buffered differential programmable gain
analog inputs as well as a differential reference input. T he part
operates from a single +5 V supply and typically consumes less
than 3 mA. It accepts four unipolar analog input ranges: 0 mV
to +10 mV, +20 mV, +40 mV and +80 mV and four bipolar
ranges ±10 mV, ±20 mV, ±40 mV and ±80 mV. T he peak-to-
peak resolution achievable directly from the part is 1 in 110,000
counts. An on-chip 6-bit DAC allows the removal of T ARE
voltages. Clock signals for synchronizing ac excitation of the
bridge are also provided.
ADDITIONAL FEATURES
Tw o-Channel Program m able Gain Front End
On-Chip DAC for Offset/ TARE Rem oval
FASTStep Mode
AC or DC Excitation
Single Supply Operation
T he serial interface on the part can be configured for three-wire
operation and is compatible with microcontrollers and digital
signal processors. T he AD7730L contains self-calibration and
system calibration options and features an offset drift of less
than 5 nV/°C and a gain drift of less than 3 ppm/°C.
APPLICATIONS
Portable Weigh Scales
T he part is available in a 24-lead SOIC and 24-lead T SSOP
package.
FUNCTIO NAL BLO CK D IAGRAM
REF IN(–)
DV
REF IN(+)
AV
DD
DD
AD7730L
REFERENCE DETECT
VBIAS
AV
DD
STANDBY
SYNC
AIN1(+)
AIN1(–)
SIGMA-DELTA A/D CONVERTER
BUFFER
100nA
SIGMA-
DELTA
PROGRAMMABLE
DIGITAL
+
–
MODULATOR
FILTER
MUX
PGA
100nA
AGND
AIN2(+)/D1
AIN2(–)/D0
MCLK IN
CLOCK
GENERATION
6-BIT
DAC
SERIAL INTERFACE
AND CONTROL LOGIC
MCLK OUT
REGISTER BANK
SCLK
CS
DIN
ACX
AC
DOUT
EXCITATION
CLOCK
ACX
AGND
DGND
POL
RDY
RESET
*Protected by U.S. Patent No: 5, 134, 401. Other Patent Applications Filed.
REV. A
–44–
AD7730/AD7730L
(AV = +5 V, DV = +3 V or +5 V; REF IN(+) =
AV ; REF IN(–) = AGND = DGND = 0 V; fCLK IN = 2.4576 MHz. All specifications TMIN to TMAX unless otherwise noted.)
AD7730L–SPECIFICATIONS
DD
DD
DD
P aram eter
B Version1
Units
Conditions/Com m ents
ST AT IC PERFORMANCE (CHP = 1)
No Missing Codes2
24
Bits min
Output Noise and Update Rates2
Integral Nonlinearity
See Tables XXI & XXII
22
See Note 3
5
ppm of FSR max
Offset Error2
Offset Error and Offset Drift Refer to Both
Unipolar Offset and Bipolar Zero Errors
Offset Drift vs. T emperature2
Offset Drift vs. T ime4
nV/°C typ
nV/1000 Hours typ
25
Positive Full-Scale Error2, 5
Positive Full-Scale Drift vs T emp2, 6, 7
Positive Full-Scale Drift vs T ime4
Gain Error2, 8
See Note 3
3
10
See Note 3
3
10
ppm of FS/°C max
ppm of FS/1000 Hours typ
Gain Drift vs. T emperature2, 6, 9
Gain Drift vs. T ime4
ppm/°C max
ppm/1000 Hours typ
Bipolar Negative Full-Scale Error2
Negative Full-Scale Drift vs. T emp2, 6
Power Supply Rejection
See Note 3
3
ppm of FS/°C max
dB typ
dB min
nA max
pA/°C typ
nA max
120
118
40
100
10
Measured with Zero Differential Voltage
At DC. Measured with Zero Differential Voltage
Common-Mode Rejection (CMR)
Analog Input DC Bias Current2
Analog Input DC Bias Current Drift2
Analog Input DC Offset Current2
Analog Input DC Offset Current Drift2
50
pA/°C typ
ST AT IC PERFORMANCE (CHP = 0)2
No Missing Codes
24
Bits min
SKIP = 010
Output Noise and Update Rates
Integral Nonlinearity
See Tables XXIII & XXIV
22
ppm of FSR max
Offset Error
See Note 3
0.5
2.5
See Note 3
0.6
3
See Note 3
2
Offset Error and Offset Drift Refer to Both
Unipolar Offset and Bipolar Zero Errors
Offset Drift vs. T emperature6
Offset Drift vs. T ime4
µV/°C typ
µV/1000 Hours typ
Positive Full-Scale Error5
Positive Full-Scale Drift vs. T emp6, 7
Positive Full-Scale Drift vs. T ime4
Gain Error8
µV/°C typ
µV/1000 Hours typ
Gain Drift vs. T emperature6, 9
ppm/°C typ
ppm/1000 Hours typ
Gain Drift vs. T ime4
10
Bipolar Negative Full-Scale Error
Negative Full-Scale Drift vs. T emp
Power Supply Rejection
See Note 3
0.6
90
µV/°C typ
dB typ
Measured with Zero Differential Voltage
Common-Mode Rejection (CMR) on AIN 105
dB typ
dB typ
nA max
pA/°C typ
nA max
pA/°C typ
At DC. Measured with Zero Differential Voltage
At DC. Measured with Zero Differential Voltage
CMR on REF IN
100
50
Analog Input DC Bias Current
Analog Input DC Bias Current Drift
Analog Input DC Offset Current
Analog Input DC Offset Current Drift
150
25
75
ANALOG INPUTS/REFERENCE INPUTS
Normal-Mode 50 Hz Rejection2
Normal-Mode 60 Hz Rejection2
Common-Mode 50 Hz Rejection2
Common-Mode 60 Hz Rejection2
Analog Inputs
88
88
120
120
dB min
dB min
dB min
dB min
From 49 Hz to 51 Hz
From 59 Hz to 61 Hz
From 49 Hz to 51 Hz
From 59 Hz to 61 Hz
Differential Input Voltage Ranges11
Assuming 2.5 V or 5 V Reference with
HIREF Bit Set Appropriately
Gain = 250
Gain = 125
Gain = 62.5
0 to +10 or ±10
0 to +20 or ±20
0 to +40 or ±40
0 to +80 or ±80
AGND + 1.2 V
AVDD – 0.95 V
mV nom
mV nom
mV nom
mV nom
V min
Gain = 31.25
Absolute/Common-Mode Voltage12
V max
Reference Input
REF IN(+) – REF IN(–) Voltage
REF IN(+) – REF IN(–) Voltage
Absolute/Common-Mode Voltage13
+2.5
+5
AGND – 30 mV
AVDD + 30 mV
0.3
V nom
V nom
V min
V max
V min
V max
HIREF Bit of Mode Register = 0
HIREF Bit of Mode Register = 1
NO REF T rigger Voltage
NO REF Bit Active If VREF Below T his Voltage
NO REF Bit Inactive If VREF Above T his Voltage
0.65
REV. A
–45–
AD7730/AD7730L
P aram eter
B Version1
Units
Conditions/Com m ents
LOGIC INPUT S
Input Current
±10
µA max
All Inputs Except SCLK and MCLK IN
V
INL, Input Low Voltage
VINL, Input Low Voltage
INH, Input High Voltage
0.8
0.4
2.0
V max
V max
V min
DVDD = +5 V
DVDD = +3 V
V
SCLK Only (Schmitt T rigerred Input)
VT +
VT +
VT –
VT –
1.4/3
1/2.5
0.8/1.4
0.4/1.1
0.4/0.8
0.4/0.8
V min to V max
V min to V max
V min to V max
V min to V max
V min to V max
V min to V max
DVDD = +5 V
DVDD = +3 V
DVDD = +5 V
DVDD = +3 V
DVDD = +5 V
DVDD = +3 V
VT + – VT –
VT + – VT –
MCLK IN Only
V
INL, Input Low Voltage
0.8
0.4
3.5
2.5
V max
V max
V min
V min
DVDD = +5 V
DVDD = +3 V
DVDD = +5 V
DVDD = +3 V
VINL, Input Low Voltage
VINH, Input High Voltage
VINH, Input High Voltage
LOGIC OUT PUT S (Including MCLK OUT )
VOL, Output Low Voltage
ISINK = 800 µA Except for MCLK OUT 14
;
;
15
0.4
0.4
4.0
V max
V max
V min
VDD = +5 V
VOL, Output Low Voltage
ISINK = 100 µA Except for MCLK OUT 14
15
VDD = +3 V
VOH, Output High Voltage
VOH, Output High Voltage
ISOURCE = 200 µA Except for MCLK OUT 14
;
;
15
VDD = +5 V
ISOURCE = 100 µA Except for MCLK OUT 14
15
VDD – 0.6 V
V min
VDD = +3 V
Floating State Leakage Current
±10
9
µA max
pF typ
Floating State Output Capacitance2
T RANSDUCER BURNOUT
AIN1(+) Current
AIN1(–) Current
–100
100
±10
0.1
nA nom
nA nom
% typ
Initial T olerance @ 25°C
Drift2
%/°C typ
OFFSET (T ARE) DAC
Resolution
6
Bit
LSB Size
2.3/2.6
3.5
25
mV min/mV max
ppm/°C max
ppm/1000 Hours typ
LSB max
2.5 mV Nominal with 5 V Reference (REF IN/2000)
Guaranteed Monotonic
DAC Drift16
DAC Drift vs. T ime4, 16
Differential Linearity
–0.25/+0.75
SYST EM CALIBRAT ION
Positive Full-Scale Calibration Limit17
1.05 × FS
V max
FS Is the Nominal Full-Scale Voltage
(10 mV, 20 mV, 40 mV or 80 mV)
Negative Full-Scale Calibration Limit17
Offset Calibration Limit18
Input Span17
–1.05 × FS
–1.05 × FS
0.8 × FS
V max
V max
V min
V max
2.1 × FS
POWER REQUIREMENT S
Power Supply Voltages
AVDD – AGND Voltage
DVDD Voltage
+4.75 to +5.25 V min to V max
+2.7 to +5.25
V min to V max
With AGND = 0 V
Power Supply Currents
AVDD Current (Normal Mode)
External MCLK. Digital I/Ps = 0 V or DVDD
All Input Ranges Except 0 mV to +10 mV and ±10 mV,
T ypically 2.7 mA
Input Ranges of 0 mV to +10 mV and ±10 mV Only,
T ypically 4 mA
3.7
5.5
mA max
AVDD Current (Normal Mode)
mA max
DVDD Current (Normal Mode)
DVDD Current (Normal Mode)
AVDD + DVDD Current (Standby Mode)
Power Dissipation
0.45
1
21
mA max
mA max
µA max
DVDD of 2.7 V to 3.3 V, T ypically 0.3 mA
DVDD of 4.75 V to 5.25 V, T ypically 0.75 mA
T ypically 13 µA. External MCLK IN = 0 V or DVDD
AVDD = DVDD = +5 V. Digital I/Ps = 0 V or DVDD
All Input Ranges Except 0 mV to +10 mV and ±10 mV,
T ypically 15 mW
Input Ranges of 0 mV to +10 mV and ±10 mV Only,
T ypically 23.75 mW
T ypically 65 µW. External MCLK IN = 0 V or DVDD
Normal Mode
23.5
32.5
105
mW max
mW max
Standby Mode
µW max
REV. A
–46–
AD7730/AD7730L
NOT ES
11T emperature range: –40°C to +85°C.
12Sample tested during initial release.
13The offset (or zero) numbers with CHP = 1 are typically 3 µV precalibration. Internal zero-scale calibration reduces this by about 1 µV. Offset numbers with CHP = 0 can be up to
1 mV precalibration. Internal zero-scale calibration reduces this to 2 µV typical. System zero-scale calibration reduces offset numbers with CHP = 1 and CHP = 0 to the order of the
noise. Gain errors can be up to 3000 ppm precalibration with CHP = 0 and CHP = 1. Performing internal full-scale calibrations on the 80 mV range reduces the gain error to less than
100 ppm for the 80 mV and 40 mV ranges, to about 250 ppm for the 20 mV range and to about 500 ppm on the 10 mV range. System full-scale calibration reduces this to the order of
the noise. Positive and negative full-scale errors can be calculated from the offset and gain errors.
14T hese numbers are generated during life testing of the part.
15Positive Full-Scale Error includes Offset Errors (Unipolar Offset Error or Bipolar Zero Error) and applies to both unipolar and bipolar input ranges.
16Recalibration at any temperature will remove these errors.
17Full-Scale Drift includes Offset Drift (Unipolar Offset Drift or Bipolar Zero Drift) and applies to both unipolar and bipolar input ranges.
18Gain Error is a measure of the difference between the measured and the ideal span between any two points in the transfer function. T he two points used to calculate the gain
error are positive full scale and negative full scale. See T erminology.
19Gain Error Drift is a span drift and is effectively the drift of the part if zero-scale calibrations only were performed.
10No Missing Codes performance with CHP = 0 and SKIP = 1 is reduced below 24 bits for SF words lower than 180 decimal.
11T he analog input voltage range on the AIN1(+) and AIN2(+) inputs is given here with respect to the voltage on the AIN1(–) and AIN2(–) inputs respectively.
12T he common-mode voltage range on the input pairs applies provided the absolute input voltage specification is obeyed.
13T he common-mode voltage range on the reference input pair (REF IN(+) and REF IN(–)) applies provided the absolute input voltage specification is obeyed.
14T hese logic output levels apply to the MCLK OUT output only when it is loaded with a single CMOS load.
15
V
DD
refers to DVDD for all logic outputs expect D0, D1, ACX and ACX where it refers to AVDD. In other words, the output logic high for these four outputs is determined by AVDD.
16T his number represents the total drift of the channel with a zero input and the DAC output near full scale.
17After calibration, if the input voltage exceeds positive full scale, the converter will output all 1s. If the input is less than negative full scale, the device outputs all 0s.
18T hese calibration and span limits apply provided the absolute input voltage specification is obeyed. T he offset calibration limit applies to both the unipolar zero point and the
bipolar zero point.
Specifications subject to change without notice.
(AV = +4.75 V to +5.25 V; DV = +3 V to +5.25 V; AGND = DGND = 0 V; fCLK IN = 2.4576 MHz;
DD
DD
1, 2
TIMING CHARACTERISTICS Input Logic 0 = 0 V, Logic 1 = DV unless otherwise noted).
DD
Lim it at TMIN to TMAX
P aram eter
(B Version)
Units
Conditions/Com m ents
Master Clock Range
1
5
50
MHz min
MHz max
ns min
For Specified Performance
t1
t2
SYNC Pulsewidth
RESET Pulsewidth
50
ns min
Read O per ation
t3
0
0
0
ns min
ns min
ns min
ns max
ns max
ns min
ns max
ns max
ns min
ns min
ns min
ns min
ns max
ns max
RDY to CS Setup T ime
t44
CS Falling Edge to SCLK Active Edge Setup T ime3
SCLK Active Edge to Data Valid Delay3
DVDD = +4.75 V to +5.25 V
t5
60
80
0
60
80
100
100
0
DVDD = +2.75 V to +3.3 V
CS Falling Edge to Data Valid Delay
DVDD = +4.75 V to +5.25 V
DVDD = +2.7 V to +3.3 V
SCLK High Pulsewidth
4, 5
t5A
t6
t7
SCLK Low Pulsewidth
t86
CS Rising Edge to SCLK Inactive Edge Hold T ime3
Bus Relinquish T ime after SCLK Inactive Edge3
t9
10
80
100
t10
SCLK Active Edge to RDY High3, 7
Wr ite O per ation
t11
t12
t13
t14
t15
t16
0
ns min
ns min
ns min
ns min
ns min
ns min
CS Falling Edge to SCLK Active Edge Setup T ime3
Data Valid to SCLK Edge Setup T ime
Data Valid to SCLK Edge Hold T ime
SCLK High Pulsewidth
SCLK Low Pulsewidth
CS Rising Edge to SCLK Edge Hold T ime
30
25
100
100
0
NOT ES
1Sample tested during initial release to ensure compliance. All input signals are specified with tr = tf = 5 ns (10% to 90% of DV DD) and timed from a voltage level of 1.6 V.
2See Figures 18 and 19.
3SCLK active edge is falling edge of SCLK with POL = 1; SCLK active edge is rising edge of SCLK with POL = 0.
4T hese numbers are measured with the load circuit of Figure 1 and defined as the time required for the output to cross the VOL or VOH limits.
5T his specification only comes into play if CS goes low while SCLK is low (POL = 1) or if CS goes low while SCLK is high (POL = 0). It is primarily required for
interfacing to DSP machines.
6T hese numbers are derived from the measured time taken by the data output to change 0.5 V when loaded with the circuit of Figure 1. T he measured number is then
extrapolated back to remove effects of charging or discharging the 50 pF capacitor. T his means that the times quoted in the timing characteristics are the true bus
relinquish times of the part and as such are independent of external bus loading capacitances.
7RDY returns high after the first read from the device after an output update. T he same data can be read again, if required, while RDY is high, although care should
be taken that subsequent reads do not occur close to the next output update.
REV. A
–47–
AD7730/AD7730L
O UTP UT NO ISE AND RESO LUTIO N SP ECIFICATIO N
T he AD7730L can be programmed to operate in either chop mode or nonchop mode. T he chop mode can be enabled in ac-excited
or dc-excited applications; it is optional in dc-excited applications, but chop mode must be enabled in ac-excited applications. T hese
options are discussed in more detail in earlier sections. T he chop mode has the advantage of lower drift numbers and better noise
immunity, but the noise is approximately 20% higher for a given –3 dB frequency and output data rate. It is envisaged that the major-
ity of weigh-scale users of the AD7730L will operate the part in chop mode to avail themselves of the excellent drift performance and
noise immunity when chopping is enabled. T he following tables outline the noise performance of the part in both chop and nonchop
modes over all input ranges for a selection of output rates.
O utput Noise (CH P = 1)
T his mode is the primary mode of operation of the device. T able XXI shows the output rms noise for some typical output update
rates and –3 dB frequencies for the AD7730 when used in chopping mode (CHP of Filter Register = 1) with a master clock
frequency of 2.4576 MHz. T hese numbers are typical and are generated at a differential analog input voltage of 0 V. T he output
update rate is selected via the SF0 to SF11 bits of the Filter Register. T able XXII, meanwhile, shows the output peak-to-peak resolu-
tion in counts for the same output update rates. T he numbers in brackets are the effective peak-to-peak resolution in bits (rounded to
the nearest 0.5 LSB). It is important to note that the numbers in T able XXII represent the resolution for which there will be no code
flicker within a six-sigma limit. T hey are not calculated based on rms noise, but on peak-to-peak noise.
T he numbers are generated for the bipolar input ranges. When the part is operated in unipolar mode, the output noise will be the
same as the equivalent bipolar input range. As a result, the numbers in T able XXI will remain the same for unipolar ranges while the
numbers in T able II will change. T o calculate the numbers for T able XXII for unipolar input ranges simply divide the peak-to-peak
resolution number in counts by two or subtract one from the peak-to-peak resolution number in bits.
Table XXI. O utput Noise vs. Input Range and Update Rate (CH P = 1)
Typical O utput RMS Noise in nV
O utput
D ata Rate Frequency
–3 dB
SF
Word
Settling Tim e
Norm al Mode
Settling Tim e
Fast Mode
Input Range
= ؎80 m V
Input Range Input Range Input Range
= ؎40 m V
= ؎20 m V
= ؎10 m V
25 Hz
50 Hz
75 Hz
100 Hz*
0.98 Hz
1.97 Hz
2.96 Hz
3.95 Hz
7.9 Hz
2048
1024
683
512
256
920 ms
460 ms
306 ms
230 ms
115 ms
120 ms
60 ms
40 ms
30 ms
15 ms
245
340
420
500
650
140
220
270
290
490
105
160
170
180
280
70
100
110
130
165
200 Hz
*Power-On Default
Table XXII. P eak-to-P eak Resolution vs. Input Range and Update Rate (CH P = 1)
P eak-to-P eak Resolution in Counts (Bits)
O utput
D ata Rate Frequency
–3 dB
SF
Word
Settling Tim e
Norm al Mode
Settling Tim e
Fast Mode
Input Range
= ؎80 m V
Input Range Input Range Input Range
= ؎40 m V
= ؎20 m V
= ؎10 m V
25 Hz
50 Hz
75 Hz
100 Hz*
200 Hz
0.98 Hz
1.97 Hz
2.96 Hz
3.95 Hz
7.9 Hz
2048
1024
683
512
256
920 ms
460 ms
306 ms
230 ms
115 ms
120 ms
60 ms
40 ms
30 ms
15 ms
110k (17)
80k (16.5)
62k (16)
53k (15.5)
44k (15.5)
94k (16.5)
60k (16)
50k (15.5)
46k (15.5)
27k (15)
64k (16)
42k (15.5)
39k (15)
36k (15)
24k (14.5)
46k (15.5)
33k (15)
31k (15)
25k (14.5)
20k (14.5)
*Power-On Default
O utput Noise (CH P = 0)
T able XXIII shows the output rms noise for some typical output update rates and –3 dB frequencies for the AD7730L when used in
nonchopping mode (CHP of Filter Register = 0) with a master clock frequency of 2.4576 MHz. T hese numbers are typical and are
generated at a differential analog input voltage of 0 V. T he output update rate is selected via the SF0 to SF11 bits of the Filter Regis-
ter. T able XXIV, meanwhile, shows the output peak-to-peak resolution in counts for the same output update rates. T he numbers in
brackets are the effective peak-to-peak resolution in bits (rounded to the nearest 0.5 LSB). It is important to note that the numbers in
T able XXIV represent the resolution for which there will be no code flicker within a six-sigma limit. T hey are not calculated based on
rms noise, but on peak-to-peak noise.
T he numbers are generated for the bipolar input ranges. When the part is operated in unipolar mode, the output noise will be the
same as the equivalent bipolar input range. As a result, the numbers in T able XXIII will remain the same for unipolar ranges while
the numbers in T able XXIV will change. T o calculate the number for T able XXIV for unipolar input ranges simply divide the peak-
to-peak resolution number in counts by two or subtract one from the peak-to-peak resolution number in bits.
REV. A
–48–
AD7730/AD7730L
Table XXIII. O utput Noise vs. Input Range and Update Rate (CH P = 0)
Typical O utput RMS Noise in nV
O utput
D ata Rate Frequency
–3 dB
SF
Word
Settling Tim e
Norm al Mode
Settling Tim e
Fast Mode
Input Range
= ؎80 m V
Input Range Input Range Input Range
= ؎40 m V
= ؎20 m V
= ؎10 m V
75 Hz
2.9 Hz
3.9 Hz
5.85 Hz
11.7 Hz
23.4 Hz
2048
1536
1024
512
332 ms
250 ms
166 ms
83 ms
53.2 ms
40 ms
26.6 ms
13.3 ms
6.6 ms
320
325
410
590
910
215
245
275
370
580
135
160
180
265
350
100
110
130
180
220
100 Hz
150 Hz
300 Hz
600 Hz
256
41.6 ms
Table XXIV. P eak-to-P eak Resolution vs. Input Range and Update Rate (CH P = 0)
P eak-to-P eak Resolution in Counts (Bits)
O utput
D ata Rate Frequency
–3 dB
SF
Word
Settling Tim e
Norm al Mode
Settling Tim e
Fast Mode
Input Range
= ؎80 m V
Input Range Input Range Input Range
= ؎40 m V
= ؎20 m V
= ؎10 m V
75 Hz
2.9 Hz
3.9 Hz
5.85 Hz
11.7 Hz
23.4 Hz
2048
1536
1024
512
332 ms
250 ms
166 ms
83 ms
53.2 ms
40 ms
26.6 ms
13.3 ms
6.63 ms
85k (16.5)
82k (16.5)
65k (16)
45k (15.5)
30k (15)
62k (16)
49k (15.5)
42k (15.5)
36k (15)
25k (14.5)
19k (14)
33k (15)
30k (15)
25k (14.5)
18k (14)
15k (14)
100 Hz
150 Hz
300 Hz
600 Hz
55k (15.5)
48k (15.5)
36k (15)
256
41.6 ms
23k (14.5)
–49–
REV. A
AD7730/AD7730L
P AGE IND EX
SERIAL INT ERFACE . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Write Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Read Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
CONFIGURING T HE AD7730 . . . . . . . . . . . . . . . . . . . . . 37
MICROCOMPUT ER/MICROPROCESSOR
INT ERFACING . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
AD7730 to 68HC11 Interface . . . . . . . . . . . . . . . . . . . . . 38
AD7730 to 8051 Interface . . . . . . . . . . . . . . . . . . . . . . . . 38
AD7730 to ADSP-2105 Interface . . . . . . . . . . . . . . . . . . 39
APPLICAT IONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
DC Excitation of Bridge . . . . . . . . . . . . . . . . . . . . . . . . . . 40
AC Excitation of Bridge . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Bipolar Excitation of Bridge . . . . . . . . . . . . . . . . . . . . . . . 42
APPENDIX A–AD7730L SPECIFICAT IONS . . . . . . . . . . 43
SPECIFICAT IONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
INDEX . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50
OUT LINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . 51
Topic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . P age
FEAT URES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
GENERAL DESCRIPT ION . . . . . . . . . . . . . . . . . . . . . . . . . 1
AD7730 SPECIFICAT IONS . . . . . . . . . . . . . . . . . . . . . . . . 2
T IMING CHARACT ERIST ICS . . . . . . . . . . . . . . . . . . . . . 4
ABSOLUT E MAXIMUM RAT INGS . . . . . . . . . . . . . . . . . 5
ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
DET AILED FUNCT IONAL BLOCK DIAGRAM . . . . . . . 6
SIGNAL PROCESSING CHAIN . . . . . . . . . . . . . . . . . . . . . 7
PIN CONFIGURAT ION . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
PIN FUNCT ION DESCRIPT ION . . . . . . . . . . . . . . . . . . . 7
T ERMINOLOGY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
OUT PUT NOISE AND RESOLUT ION
SPECIFICAT ION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
ON-CHIP REGIST ERS . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Summary Of On-Chip Registers . . . . . . . . . . . . . . . . . . . . 12
Communications Register . . . . . . . . . . . . . . . . . . . . . . . . 13
Status Register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Data Register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Mode Register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Filter Register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
DAC Register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Offset Calibration Register . . . . . . . . . . . . . . . . . . . . . . . . 20
Gain Calibration Register . . . . . . . . . . . . . . . . . . . . . . . . . 20
T est Register . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
READING FROM AND WRIT ING T O T HE
ON-CHIP REGIST ERS. . . . . . . . . . . . . . . . . . . . . . . . . . 21
CALIBRAT ION OPERAT ION SUMMARY . . . . . . . . . . . 22
CIRCUIT DESCRIPT ION . . . . . . . . . . . . . . . . . . . . . . . . 23
ANALOG INPUT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Analog Input Channels . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Analog Input Ranges . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Bipolar/Unipolar Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Burnout Currents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
REFERENCE INPUT . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Reference Detect . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
SIGMA-DELT A MODULAT OR . . . . . . . . . . . . . . . . . . . . 26
DIGIT AL FILT ERING . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Filter Architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
First Stage Filter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Second Stage Filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
CALIBRAT ION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Internal Zero-Scale Calibration . . . . . . . . . . . . . . . . . . . . 30
Internal Full-Scale Calibration . . . . . . . . . . . . . . . . . . . . . 30
System Zero-Scale Calibration . . . . . . . . . . . . . . . . . . . . . 31
System Full-Scale Calibration . . . . . . . . . . . . . . . . . . . . . 31
Span and Offset Limits . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Power-Up and Calibration . . . . . . . . . . . . . . . . . . . . . . . . 32
Drift Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
USING T HE AD7730 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
Clocking and Oscillator Circuit . . . . . . . . . . . . . . . . . . . . 32
System Synchronization . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Single-Shot Conversions . . . . . . . . . . . . . . . . . . . . . . . . . 33
Reset Input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Standby Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Digital Outputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
POWER SUPPLIES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Grounding and Layout . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Evaluating the AD7730 Performance . . . . . . . . . . . . . . . . 34
TABLE IND EX
Table
T able I.
Title
P age
10
Output Noise vs. Input Range and
Update Rate (CHP = 1)
Peak-to-Peak Resolution vs. Input Range
and Update Rate (CHP = 1)
Output Noise vs. Input Range and
Update Rate (CHP = 0)
Peak-to-Peak Resolution vs. Input Range
and Update Rate (CHP = 0)
Summary of On-Chip Registers
Communications Register
Read/Write Mode
Register Selection
Status Register
Mode Register
Operating Modes
Input Range Selection
Channel Selection
Filter Register
SF Ranges
DAC Register
T able II.
T able III.
T able IV.
10
11
11
12
13
13
14
14
15
15
17
18
18
19
20
22
23
T able V.
T able VI.
T able VII.
T able VIII.
T able IX.
T able X.
T able XI.
T able XII.
T able XIII.
T able XIV.
T able XV.
T able XVI.
T able XVII.
Calibration Operations
T able XVIII. Reset Events
T able XIX.
T able XX.
Pseudo-Code for Initiating a
Self-Calibration after Power-On/Reset
Pseudo-Code for Setting Up AD7730 for
Continuous Conversion and Continuous
Read Operation
Output Noise vs. Input Range and
Update Rate (CHP = 1)
37
37
48
48
49
49
T able XXI.
T able XXII.
Peak-to-Peak Resolution vs. Input Range
and Update Rate (CHP = 1)
T able XXIII. Output Noise vs. Input Range and
Update Rate (CHP = 0)
T able XXIV. Peak-to-Peak Resolution vs. Input Range
and Update Rate (CHP = 0)
REV. A
–50–
AD7730/AD7730L
O UTLINE D IMENSIO NS
D imensions shown in inches and (mm).
P lastic D IP
(N-24)
1.275 (32.30)
1.125 (28.60)
13
24
1
0.280 (7.11)
0.240 (6.10)
12
0.325 (8.25)
0.300 (7.62)
0.195 (4.95)
0.115 (2.93)
PIN 1
0.060 (1.52)
0.015 (0.38)
0.210
(5.33)
MAX
0.150
(3.81)
MIN
0.200 (5.05)
0.125 (3.18)
0.015 (0.381)
0.008 (0.204)
0.100 (2.54)
BSC
0.022 (0.558)
0.014 (0.356)
0.070 (1.77)
0.045 (1.15)
SEATING
PLANE
Sm all O utline
(R-24)
0.6141 (15.60)
0.5985 (15.20)
24
13
12
1
PIN 1
0.1043 (2.65)
0.0926 (2.35)
0.0291 (0.74)
x 45°
0.0098 (0.25)
0.0500 (1.27)
0.0157 (0.40)
8°
0°
0.0500
(1.27)
BSC
0.0118 (0.30)
0.0040 (0.10)
0.0192 (0.49)
SEATING
PLANE
0.0125 (0.32)
0.0138 (0.35)
0.0091 (0.23)
Thin Shr ink Sm all O utline
(RU-24)
0.311 (7.90)
0.303 (7.70)
24
13
12
1
0.006 (0.15)
0.002 (0.05)
PIN 1
0.0433
(1.10)
MAX
0.028 (0.70)
0.020 (0.50)
8°
0°
0.0256 (0.65) 0.0118 (0.30)
SEATING
PLANE
0.0079 (0.20)
0.0035 (0.090)
BSC
0.0075 (0.19)
–51–
REV. A
–52–
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